Circuit arrangement for actuating a transistor

[0001] Embodiments of the invention relate to a circuit arrangement for actuating a transistor comprising a driver circuit for actuating a high-side transistor and a power supply circuit for providing a supply voltage to the driver circuit.

[0002] Loads are frequently switched using what are known as high-side switches which are connected between the load and a circuit node which is at operating voltage. The load is therefore connected between the high-side switch and a circuit node which is at reference potential (usually ground). High-side switches are often used in preference to low-side switches connected between reference potential and load because, when high-side switches are used, the operating voltage is not applied to the load when they are in the off state. Often, transistor half-bridges comprising a low-side switching transistor and a high-side switching transistor are also used for switching loads, such as in power converters.

[0003] Suitable switches are, in principle, all kinds of transistors and thyristors. In the field of the power electronics, power MOSFETs and IGBTs are used as switching transistors. In contrast to logic circuits, where transistor half-bridges are frequently manufactured in CMOS technologies, NMOS transistors are preferred for switching loads in power due to their lower turn-on resistance.

[0004] However, actuating the high-side transistor in pure NMOS (and also in NPN bipolar) half-bridges requires a voltage which is higher than the operating voltage of the half-bridge and of the load, respectively. For this reason, driver circuits require a dedicated power supply for actuating high-side semiconductor switches. This can be provided, by way of example, by a capacitor which is charged via a diode when the high-side switch is off. Such an arrangement comprising a capacitor and a diode for supplying power to the floating (free of ground) driver circuit is also called a bootstrap circuit.

[0005] However, one problem is that such circuit arrangements comprising a high-side switch, a driver circuit for actuating the high-side switch and a bootstrap circuit for providing supply voltage to the driver are not capable of keeping the high-side switch in an on-state for as long as desired, since the capacitor is discharged by a current consumption, albeit a low one, in the driver circuit. Particularly in very-large-scale-integrated circuits, it is a complex matter to manufacture capacitors of appropriate size, and therefore the possible turned-on time of a high-side switch is limited by the size of the capacitor and the current consumption of the driver circuit.

[0006] However, there are applications in which the limited turned-on time of high-side switches is troublesome, and there is a need for high-side switches which can be kept in an on-state for an arbitrarily long time. Circuits are known which ensure that the driver circuit of the high-side switch is supplied with power permanently, e.g., with the aid of DC/DC converters. Such circuit arrangements are usually very complex, however. There is therefore also a need for a simple and inexpensive circuit for permanently supplying power to driver circuits for high-side switches.

[0007] One example of the invention relates to a circuit arrangement for actuating a high-side transistor which comprises a control terminal and a load terminal; the circuit arrangement comprises: a driver circuit that is designed to generate, in response to a control signal, a driver signal for the control terminal of the high-side transistor; a radio-frequency signal source; a supply circuit which is capacitively coupled to the radio-frequency signal source and which is designed to provide a supply voltage to the driver circuit, the supply voltage being referenced to a floating reference potential.

[0015] As described above, driver circuits for actuating high-side transistors require a power supply, where the supply voltage for the driver circuit referenced to a fixed reference potential V_SS (e.g., ground) is higher than the operating voltage of the load to be switched when the high-side transistor is in the on-state. However, the supply voltage is not referenced to the fixed reference potential V_SS, but rather to the load connection of the high-side switching transistor which faces the load. In the case of an N-channel MOSFET as the high-side switch, this is the source connection, and in the case of an NPN bipolar transistor or an N-channel IGBT it is the emitter connection. The potential of the load connection (i.e., source potential or emitter potential) is thus the ground-free reference potential for the driver circuit “from the point of view” of the high-side transistor or its driver circuit. This ground-free reference potential V_REFHS is approximately the same as the fixed reference potential V_SS when the high-side switch is in the off-state, and it is approximately the same as the operating voltage V_DD of the load when the high-side switch is in the on-state. This approximation applies when the voltage drop when the transistor is in the on-state is negligibly small in comparison with the operating voltage of the load.

[0016] A good example of what has been described above is shown in FIG. 1. FIG. 1 is a diagram of a circuit arrangement comprising a half-bridge with a high-side switch T_HS and a low-side switch T_LS, two driver circuits (high-side driver 11 and low-side driver 12) for actuating the transistor switches T_HS and T_LS, a voltage source which provides a DC supply voltage V₁ for supplying power to the driver circuit 12, and a bootstrap circuit 10 for supplying power to the driver circuit 11. The driver circuits 11 and 12 are connected to the respective control connections of the associated transistors T_HS, T_LS and are designed to take a control signal S_HS or S_LS as a basis for turning the respective transistor T_HS or T_LS on or off. The supply voltage for the high-side driver 11 is denoted by V₂, and that for the low-side driver 12 is denoted by V₁.

[0017] In the case illustrated, the high-side transistor T_HS is an NMOS transistor with a gate connection as control connection G and with a drain connection D and a source connection S as load connections. To actuate the high-side MOS transistor T_HS, the driver circuit 11 needs to produce a gate voltage of several volts referenced to the potential of the source connection S (which is equal to the ground-free reference potential V_REFHS). When the high-side transistor T_HS is on, the gate voltage V_GHS, referenced to the fixed reference potential V_SS (e.g., ground), is then higher than the operating voltage V_DD of the load. Since the level of the gate voltage referenced to the source potential V_REFHS is relevant to the high-side transistor T_HS, i.e., the gate/source voltage V_GHS−V_REFHS, the supply voltage V₂ for the high-side driver 11 is also referenced to the ground-free reference potential V_REFHS. In contrast to this, the supply voltage V₁ for the low-side driver 12 is referenced to the fixed reference potential V_SS. The supply voltage V₁ is provided by an external voltage source, and the supply voltage V₂ is provided by a capacitor C_HS which is charged by means of the bootstrap circuit 10. To this end, the capacitor C_HS is connected firstly to the high-side driver 11 at a circuit node N₁ and secondly to the source connection S of the high-side transistor T_HS, that is to say to the ground-free reference potential V_REFHS. In order to charge the capacitor C_HS, the bootstrap circuit 10 comprises a diode D_BS which is connected between the capacitor C_HS (at the circuit node N₁) and a connection for the supply voltage V₁ (at a circuit node N₂) for the low-side driver 12 such that the capacitor C_HS is charged via the diode D_BS when the high-side switch T_HS is in an off-state (and the low-side switch T_LS is on) and therefore the ground-free reference potential V_REFHS is approximately the same as the fixed reference potential V_SS (e.g., ground). To limit the charging current for the capacitor C_HS, a resistor R_b may also be connected in series with the diode D_BS. In addition, a Zener diode (not shown) may be connected in parallel with the capacitor C_HS in order to limit the capacitor voltage on the capacitor C_HS.

[0018] When the high-side transistor T_HS is in the on-state, the capacitor C_HS is prevented from being discharged by the diode D_BS, so that the capacitor C_HS can provide the supply voltage V₂ for the high-side driver 11.

[0019] However, the capacitor C_HS is nevertheless discharged on account of the current consumption of the high-side driver 11 and on account of leakage currents. It is obvious that when the circuit arrangement shown in FIG. 1 is used, the turned-on time of the high-side switch T_HS is likewise limited by the limited capacitance of the capacitor C_HS.

[0020] A circuit arrangement which is improved over the circuit arrangement described in FIG. 1 is shown in FIG. 2.

[0021] The circuit arrangement from FIG. 2 essentially corresponds to the circuit arrangement from FIG. 1, but in line with a first example of the invention the bootstrap circuit 10 is replaced by a high-side power supply circuit 20 which is capacitively coupled to a radio-frequency signal source Q_RF. The aforementioned capacitive coupling is provided by means of a capacitor C_E which is connected between the power supply circuit 20 and the radio-frequency signal source Q_RF. The radio-frequency signal source Q_RF is operated at the fixed reference potential V_SS (that is to say at the “low signal” of the circuit arrangement) and produces a radio-frequency current signal which is supplied to the power supply circuit 20 (which is situated on the “high side” of the circuit arrangement) via the capacitor C_E. The capacitor C_E decouples the DC supply voltage V₁ on the “low side” from the ground-free DC supply voltage V₂ on the “high side”, but allows a radio-frequency flow of current (and hence a flow of power) between the two circuit parts.

[0022] The power supply circuit 20 is designed to rectify the radio-frequency signal which is produced by the radio-frequency signal source 20 and capacitively coupled into the circuit 20 and to use the rectified radio-frequency signal to charge the capacitor C_HS. This prevents the capacitor C_HS from being discharged and ensures that the high-side driver 11 is permanently supplied with power. This is made possible because the capacitor C_HS allows a (radio-frequency) flow of current and hence power from the “low side” to the “high side” of the circuit arrangement regardless of the absolute value of the ground-free (“floating”) reference potential V_REFHS.

[0023] FIG. 3 shows a diagram of a further example of a circuit arrangement according to an embodiment of the invention. The circuit arrangement from FIG. 3 essentially corresponds to the circuit arrangement from FIG. 2. In comparison with the circuit arrangement from FIG. 2, the circuit arrangement from FIG. 3 comprises an additional bootstrap circuit 10′. The bootstrap circuit 10′ comprises not only the diode D_BS and the optional series resistor R_b (see FIG. 1: bootstrap circuit 10) but also, additionally, inductive components L_EL and L_EH which are connected to the power supply circuit 20, to the high-side driver 11 and to the connection of the supply voltage V₁ for the low-side driver (i.e., to the circuit node N₂) such that the radio-frequency signal is decoupled from the DC supply voltages V₁ and V₂.

[0024] In contrast to the circuit arrangement from FIG. 2, the coupling capacitor C_E is not connected to the power supply circuit 20 directly, but rather the diode D_BS is also connected between the capacitor C_E and the power supply circuit 20. The capacitive coupling between the radio-frequency signal source Q_RF and the power supply circuit 20 is thus provided by means of the coupling capacitor C_E and by means of the junction capacitance of the diode D_BS. In the present arrangement, the diode D_BS performs a dual function. Firstly, the diode D_BS, as a component of the bootstrap circuit 10′, serves to charge the capacitor C_HS when the high-side transistor T_HS is off (and the low-side transistor is on). Secondly, the diode D_BS also serves to capacitively couple the radio-frequency signal to the power supply circuit 20 by means of its junction capacitance.

[0025] The coupling capacitor C_E and the junction capacitance of the diode D_BS can be used to supply the high-side driver 11 permanently with power and to allow the high-side transistor T_HS to remain permanently on, even if the diode D_BS is reverse-biased in this case. To achieve the capacitive coupling, it would suffice to connect the coupling capacitor C_E directly to the power supply circuit 20. In this case, the coupling capacitor C_E would need to be high-voltage resistant, however, since the total operating voltage V_DD−V_SS of the half-bridge in this case is present across the capacitor C_E. However, a high-voltage capacitor is too expensive and has excessive geometric dimensions for many applications. Furthermore, the output of the radio-frequency signal source Q_RF would at least briefly (when the half-bridge is switched) need to be able to accept the full operating voltage V_DD−V_SS without being damaged. Such a radio-frequency signal source Q_RF can also be implemented in an integrated circuit only with very great complexity, however.

[0026] The series circuit comprising the coupling capacitor C_E and the diode D_BS ensures sufficient capacitive coupling of the radio-frequency signal to the power supply circuit 20 if the junction capacitance of the diode D_BS is high enough. Furthermore, the high voltage (operating voltage V_DD−V_SS) can readily be accepted by the diode, and the coupling capacitor C_E can be in the form of a low-voltage capacitor which can also be integrated on a chip. In addition, it is possible to use a simple radio-frequency signal source Q_RF which is not resistant to high voltage.

[0027] Diode types which are frequently used for such applications are MUR160 or UF1007, for example, which may have junction capacitances of about 4 pF to about 10 pF, in some cases even up to about 20 pF, which is sufficient for the inventive purpose when the radio-frequency signal source Q_RF has frequencies of between approximately 20 MHz and 200 MHz. To decouple the radio-frequency signal from the DC supply voltage V₁, the coupling capacitor C_E and the circuit node N₂ to which the supply voltage V₁ is applied have a coil L_EL connected between them. This may be integrated in or on a chip as a flat coil, for example, as described in U.S. Pat. No. 6,927,662. To decouple the radio-frequency signal and the DC supply voltage V₂, the input of the power supply circuit 20 and the high-side driver 11 (circuit node N₁) have a further coil L_EL connected between them which can likewise be integrated on a chip in microtransformer technology. The coils are intended to have inductance values of approximately 1 μH to 10 μH and may also be implemented on a PCB board as planar coils.

[0028] The circuit is closed to the RF signal by means of the drain/source (or collector/emitter) capacitances of the low-side transistor T_LS. To increase this capacitance, a further capacitor C_h may be connected in parallel with the load path of the low-side transistor T_LS.

[0029] FIG. 4 and FIG. 5 show examples of the power supply circuit 20 as may be used in the circuit arrangements from FIG. 2 or FIG. 3, for example. FIG. 4 illustrates a simple voltage multiplier circuit, also called Villard cascades. The voltage multiplier circuit comprises two cascades which each double the voltage. The first cascade comprises the capacitors C₁ and C₂ and the diodes D₁ and D₂, and the second cascade comprises the capacitors C₃ and C₄ and the diodes D₃ and D₄. In the present case, the voltage is thus multiplied by four, and any number of cascades can be strung together to achieve different voltage multipliers. Voltage multipliers are described briefly in Horowitz, Hill: The Art of Electronics, 2nd Edition, Cambridge University Press, pp. 47f, for example. To achieve the best possible impedance matching of the power supply circuit 20 to the radio-frequency signal source Q_RF, the input of the power supply circuit 20 has a coil L_S connected in series with the capacitor C₁ of the first Villard cascade. The capacitor C₁ and the coil L_S therefore form a series resonant circuit whose resonant frequency is in tune with the frequency of the radio-frequency signal.

[0030] The power supply circuit 20 from FIG. 5 essentially corresponds to the voltage multiplier circuit from FIG. 4, with a parallel resonant circuit comprising a capacitor C_P and a coil L_P being connected to the first Villard cascade instead of the series resonant circuit at the input of the power supply circuit 20. The coils can be easily integrated on a silicon chip using the aforementioned microtransformer technology.

[0031] An alternative to the Villard cascades shown in FIGS. 4 and 5 is shown in FIGS. 6a and 6b, collectively FIG. 6. The power supply circuit 20 shown in FIG. 6 comprises not only a diode D₁ as rectifier element but also a charge pump, which in the configuration shown is also referred to as a Cockcroft-Walton voltage multiplier. An overview of charge pump circuits can be found in Louie Pylarinos, Charge Pumps: An Overview, Edward S. Rogers Sr. Department of Electrical and Computer Engineering, University of Toronto, for example.

[0032] The Cockcroft-Walton circuit shown in FIG. 6 comprises three series-connected capacitors C_A, C_B and C_C with the same respective capacitance C. The capacitor C_A is connected to the input connection IN of the power supply circuit 20 via the rectifier diode D₁ and to the ground-free reference potential V_REFHS. The entire series circuit comprising the three capacitors is connected between the output and the ground-free reference potential. Switches can be used to transfer the charge stored in the capacitor C_A to a further capacitor C₁ and therefrom in turn to the capacitor C_B. In the same way, the switches are used to “pump” the charge from the capacitor C_B via a capacitor C₂ to the capacitor C_C.

[0033] The Boolean variables φ and φ (i.e., NOT φ) show the switching state of the individual switches. FIG. 6a shows a first switching state (φ=1) in which charge is transferred from the capacitor C_A to the capacitor C₁ and from the capacitor C_B to the capacitor C₂. FIG. 6b shows a second switching state (φ=0) in which charge is transferred from the capacitor C₁ to the capacitor C_B and from the capacitor C₂ to the capacitor C_C. It can easily be appreciated that after a few switching cycles the voltage, referenced to the ground-free reference potential V_REFHS, at the output OUT of the power supply circuit 20 is three times the voltage at the input IN.

[0034] Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a variety of alternate and/or equivalent implementations may be substituted for the specific embodiments shown and described without departing from the scope of the present invention. This application is intended to cover any adaptations or variations of the specific embodiments discussed herein. Therefore, it is intended that this invention be limited only by the claims and the equivalents thereof.