Power-mode controlled power converter

[0001] 1. Field of the Invention

[0002] The present invention relates to a power converter and more specifically relates to a primary-side controlled power converter.

[0003] 2. Description of Related Art

[0004] Various modes of control for power converters have been widely used to regulate an output voltage and an output current. Two common modes for such regulation are voltage-mode and current-mode.

[0005] FIG. 1 shows a traditional power converter. An error amplifier 12 and an error amplifier 13 are utilized to produce a feedback signal from the output current and the output voltage. The feedback signal will be provided to a PWM controller 11 in the primary-side circuit via an optical-coupler 14. The PWM controller 11 generates a PWM signal to switch a transformer 51 by switching a transistor 70. A switching current flowing through the transistor 70 is converted to a detection voltage signal via a resistor 20. The PWM controller 11 compares the feedback signal with the detection voltage signal to produce a PWM signal. The PWM signal will regulate the output voltage and/or the output current. However, the optical-coupler and other secondary-side control circuitry add significantly to the size and device count of the power converter. This is a main drawback of this power converter.

[0006] FIG. 2 shows the schematic circuit of a primary-side controlled power converter. The primary-side controlled power converter includes a PWM controller 12. A reflected-voltage from an auxiliary winding N_A of a transformer 50 is supplied to the PWM controller 12 via a resistor 61. This is used to provide the feedback signal for output voltage control. Since the reflected-voltage produced by the auxiliary winding N_A is strongly correlated with an output voltage provided by a secondary winding N_S, the PWM controller 12 can regulate the output voltage from the primary-side of the transformer 50.

[0007] To regulate the output current from the primary-side of the transformer 50, it is necessary to control the power delivered from the primary-side to the secondary-side of the transformer 50. The relationship between an output power P_O and a switching current I_IN of the power converter can be expressed as:

[0008] $\begin{array}{cc}{P}_O=V_O\times I_O=\eta \times P_{\mathrm{IN}}=\eta \times I_{\mathrm{IN}}\times V_{\mathrm{IN}}& \left(1\right)\\ I_{\mathrm{IN}}=\left[\left(\frac{V_{\mathrm{IN}}}{L_P}\times \frac{T_{\mathrm{ON}}}{2T}\right)+\left(I_A\times \frac{T_{\mathrm{ON}}}{T}\right)\right]& \left(2\right)\end{array}$

[0009] Where V_O is the output voltage; I_O is the output current; η is the efficiency; I_A is the reflected load current; L_P is the primary magnetized inductance; T is the switching period; and T_ON is the on-time of the switching signal.

[0010] In order to control the output voltage V_O, the on-time T_ON of the switching signal is adjusted in response to the feedback signal. When the output current I_O increases, a switching current of the transistor will also increase. The switching current of the transistor 70 will be converted to a switching-voltage. When the switching-voltage exceeds a maximum threshold voltage V_LIMIT, the on-time T_ON of the switching signal will be restricted to regulate the output current I_O The maximum input power P_IN—MAX can be expressed as:

[0011] $\begin{array}{cc}{P}_{\mathrm{IN\_MAX}}=\left[\frac{1}{2\times T}\times L_P\times {\left(\frac{V_{\mathrm{LIMIT}}}{R_S}\right)}^2\right]& \left(3\right)\end{array}$

[0012] The equations (1) and (3) can then be rewritten as:

[0013] $\begin{array}{cc}{I}_O=\frac{P_O}{V_O}=\frac{\eta \times P_{\mathrm{IN\_MAX}}}{V_O}=\left\{\frac{\eta \times \left[\frac{1}{2\times T}\times L_P\times {\left(\frac{V_{\mathrm{LIMIT}}}{R_S}\right)}^2\right]}{V_O}\right\}& \left(4\right)\end{array}$

[0014] Because the maximum input power P_IN—MAX is limited, the output voltage V_O will decrease whenever the output current I_O increases. The output voltage deviation and the output current variation (the slope of the V/I curve) determine the constant current output performance. FIG. 3 provides an illustration of this.

[0015] Referring to equation (4), when the output voltage V_O decreases, a constant output current I_O can be achieved by increasing the switching period T and/or reducing the voltage signal V_LIMIT. However, any of several factors can skew the accuracy of this method. Deviation in the primary magnetized inductance L_P and a drifting switching frequency (1/T) can cause the maximum input power limit P_IN—MAX and the output current I_O to fluctuate.

[0016] Furthermore, the propagation delay time T_D affects the precision of the output current I_O. Referring to FIG. 2, a resistor 20 is used to convert the switching current I_IN to a switching-current voltage V_IDET. As shown in FIG. 4, a voltage V_IA can be expressed as:
V_IA=I_A×Rs
Where I_A is the reflected load current; R_S is the resistance of the resistor 20.

[0017] The switching-current voltage V_IDET exceeds the threshold voltage V_LIMIT at the time T_ONX. However, V_PWM is not turned off until after a propagation delay time T_D. During this propagation delay time T_D, the switching current I_IN continues to increase. This will generate an extra input current I_IN-ex.

[0018] The amplitude of the current I_IN-ex can be calculated according to equation (5). Referring to equation (6), the extra input current I_IN-ex will cause the maximum input power P_IN-MAX and the output current I_O to increase in response to an increasing input voltage V_IN.

[0019] $\begin{array}{cc}{I}_{\mathrm{IN}-\theta x}=\left\{\left(\frac{V_{\mathrm{IN}}}{L_P}\times \frac{T_D}{2T}\right)+\left[\left(\frac{V_{\mathrm{IN}}}{L_P}\times T_{\mathrm{ONX}}\right)+I_A\right]\times \frac{T_D}{T}\right\}& \left(5\right)\\ P_{\mathrm{IN\_MAX}}=\left[\frac{1}{2\times T}\times L_P\times {\left(\frac{V_{\mathrm{LIMIT}}}{R_S}+I_{\mathrm{IN}-\mathrm{ex}}\right)}^2\right]& \left(6\right)\end{array}$

[0020] According to foregoing description, we can find that the primary-side control of the prior art is not accurate and is not fit for the mass production of the power converter.

[0021] The object of the present invention is to provide a primary-side controlled power converter that can precisely regulate the output voltage and generate a constant output current.

[0022] A power-mode controlled power converter according to the present invention comprises a transformer, a power transistor, a line resistor, an auxiliary rectifier, a decouple capacitor, a detection resistor, a sense resistor, an output rectifier, and an output capacitor, an integrate-capacitor, and a PWM controller.

[0023] The PWM controller of the power-mode controlled power converter has a supply terminal, a voltage-detection terminal, an output terminal, a current-detection terminal, a filter terminal, a line-voltage input terminal and a ground terminal.

[0024] The line resistor is connected from an input of the power-mode controlled power converter to the line-voltage input terminal to start up the PWM controller and provide input-voltage detection after the PWM controller is started up. The auxiliary rectifier connected from an auxiliary winding of the transformer to the decouple capacitor further provides DC power to the PWM controller when the PWM controller starts to operate.

[0025] The sense resistor is connected to the power transistor and the current-detection terminal of the PWM controller for providing a switching-current voltage. The integrate-capacitor connected to the filter terminal of the PWM controller forms a low-pass filter and integrates the switching-current voltage to an average-current signal. The PWM controller uses the average-current signal and a line-voltage input signal for output current regulation.

[0026] The PWM controller of the power-mode controlled power converter further includes a sample-hold circuit for sampling the voltage at the voltage-detection terminal and producing a sample-voltage. A first error amplifier compares the sample-voltage with a reference voltage to generate a feedback voltage. A first comparator compares the feedback voltage with the switching-current voltage to produce a voltage-reset signal. A programmable current-sink connected to the voltage-detection terminal generates a programmable current in response to the feedback voltage. The programmable current and the detection resistor produce an offset-voltage to compensate for the voltage drop across the output rectifier. The PWM controller also includes an attenuator to attenuate the line-voltage input signal to an input-voltage signal. A multiplier multiplies the average-current signal with the input-voltage signal to generate a power-control signal. A second error amplifier compares the power-control signal with a power-reference voltage, which is generated by a reference-voltage generator, to generate a limit voltage. A second comparator compares the limit voltage with the switching-current voltage to produce a current-reset signal.

[0027] The PWM controller further comprises an oscillator and a SR flip-flop to generate a PWM signal. The SR flip-flop is periodically set by an output of the oscillator and reset by both the current-reset signal and the voltage-reset signal. A blanking circuit produces a blanking time to ensure that the minimum on-time of the PWM signal, which creates a sufficient delay to precisely generate the sample-voltage. A filter resistor connected in between the current-detection terminal and the filter terminal provides high impedance for the low-pass filter. The PWM controller further includes a first switch and a second switch. The first switch is connected from the line-voltage input terminal to the supply terminal. The second switch is connected from the line-voltage input terminal to the attenuator. The first switch is turned off and the second switch is turned on once the PWM controller starts to operate.

[0028] It is to be understood that both the foregoing general descriptions and the following detailed descriptions are exemplary, and are intended to provide further explanation of the invention as claimed. Still further objects and advantages will become apparent from a consideration of the ensuing description and drawings.

[0042] FIG. 5 shows a power-mode controlled power converter according to the present invention. The power-mode controlled power converter according to the present invention includes a PWM controller 10 having a supply terminal VCC, a voltage-detection terminal VDET, a line-voltage input terminal VI, a filter terminal CINT, a current-detection terminal IDET, an output terminal OUT, and a ground terminal GND. The power-mode controlled power converter according to the present invention also includes a line resistor 65, a decouple capacitor 35, an auxiliary rectifier 40, a detection resistor 60, an integrate-capacitor 30, a sense resistor 20, a power transistor 70, an output rectifier 80, and an output capacitor 90. The power-mode controlled power converter of the present invention further includes a transformer 50 having an auxiliary winding N_A, a primary winding N_P, and a secondary winding N_S.

[0043] A first terminal of the line resistor 65 is supplied with an input voltage V_IN of the power converter. A second terminal of the line resistor 65 is connected to the line-voltage input terminal VI of the PWM controller 10. A positive end of the auxiliary winding N_A is connected to an anode of the auxiliary rectifier 40. A cathode of the auxiliary rectifier 40 is connected to the decouple capacitor 35. The supply terminal VCC of the PWM controller 10 is connected to the cathode of the auxiliary rectifier 40. The detection resistor 60 is connected between the anode of the auxiliary rectifier 40 and the voltage-detection terminal VDET of the PWM controller 10. The output terminal OUT of the PWM controller 10 drives a gate of the power transistor 70. The sense resistor 20 is connected between a source of the power transistor 70 and a ground reference. The current-detection terminal IDET of the PWM controller 10 is connected to the source of the power transistor 70. The integrate-capacitor 30 is connected from the filter terminal CINT of the PWM controller 10 to the ground reference. The ground terminal GND of the PWM controller 10 is connected to the ground reference. An anode of the output rectifier 80 is connected to a positive end of the secondary winding N_S. The output capacitor 90 is connected between a cathode of the output rectifier 80 and a negative terminal of the secondary winding N_S. An output voltage V_O of the power-mode controlled power converter is supplied from the cathode of the output rectifier 80.

[0044] FIG. 6 shows the PWM controller 10 according to a preferred embodiment of the present invention. The PWM controller 10 includes a filter resistor 185, two switches 140 and 145, an attenuator 180, a multiplier 190, a sample-hold circuit 155, a reference-voltage generator 160, two error amplifiers 125 and 126, a current source 115, a resistor 170, two comparators 120 and 121, two NAND gates 108 and 109, an oscillator 100, a SR flip-flop 107, a blanking circuit 150, a programmable current-sink 110, and a buffer 105.

[0045] The switch 140 is connected between the supply terminal VCC and the line-voltage input terminal VI. The switch 145 is connected between the line-voltage input terminal VI and an input of the attenuator 180. Referring to FIG. 5 and FIG. 6, the switch 140 is turned on via the line resistor 65 to start up the PWM controller 10. After the PWM controller 10 starts up, the switch 140 will be turned off and the switch 145 will be turned on. To detect the line-voltage, a line-voltage input signal is then sent to the attenuator 180 via the switch 145.

[0046] The filter resistor 185 is connected between the current-detection terminal IDET and the filter terminal CINT. The filter resistor 185 is coupled with the integrate-capacitor 30 to supply an average-current signal to a first input of the multiplier 190. The multiplier 190 has a second input supplied with an input-voltage signal. The input-voltage signal is generated from an output of the attenuator 180. The multiplier 190 also has an output for supplying a power-control signal to a negative input of the error amplifier 126.

[0047] The sample-hold circuit 155 has a first input connected to the voltage-detection terminal VDET. The sample-hold circuit 155 further has an output for supplying a sample-voltage V_SH to a negative input of the error amplifier 125. A positive input of the error amplifier 125 is supplied with a reference voltage V_R1. An input of the reference-voltage generator 160 is supplied with the sample-voltage V_SH. An output of the reference-voltage generator 160 supplies a power-reference voltage V_R2 to a positive input of the error amplifier 126. An output of the error amplifier 126 supplies a limit voltage V_LIMIT to a positive input of the comparator 121.

[0048] An input terminal of the current source 115 is supplied with a supply voltage V_CC. An output terminal of the current source 115 is connected to the positive input of the comparator 121. The resistor 170 is connected from the output terminal of the current source 115 to the ground reference. Since the error amplifier 126 of the present invention is an open-drain device, the maximum value of the limit voltage V_LIMIT is determined by the current source 115 and the resistance of the resistor 170.

[0049] A negative input of the comparator 120 and a negative input of the comparator 121 are connected to the current-detection terminal IDET to receive a switching-current voltage V_IDET. The switching-current voltage V_IDET is converted from a switching current I_IN of the power transistor 70 by the sense resistor 20. The error amplifier 125 compares the sample-voltage V_SH and the reference voltage V_R1 to generate a feedback voltage V_FB via an output of the error amplifier 125. The feedback voltage V_FB is supplied to a positive input of the comparator 120 and a control terminal of the programmable current-sink 110.

[0050] The programmable current-sink 110 is connected between the first input of the sample-hold circuit 155 and the ground reference. The comparator 120 compares the feedback voltage V_FB with the switching-current voltage V_IDET to generate a voltage-reset signal. The voltage-reset signal is supplied to a first input of the NAND gate 108. The comparator 121 compares the limit voltage V_LIMIT with the switching-current voltage V_IDET to generate a current-reset signal. The current-reset signal is supplied to a second input of the NAND gate 108. An output of the NAND gate 108 is connected to a first input of the NAND gate 109. An output of the oscillator 100 sets the SR flip-flop 107. The SR flip-flop 107 is reset by an output of the NAND gate 109. An output of the SR flip-flop 107 supplies a PWM signal V_PWM to a second input of the sample-hold circuit 155, an input of the buffer 105, and an input of the blanking circuit 150. An output of the blanking circuit 150 supplies a blanking signal V_BLK to a second input of the NAND gate 109. The buffer 105 connected between the output of the SR flip-flop 107 and the output terminal OUT of the PWM controller 10 provides sufficient driving capability for the power transistor 70.

[0051] FIG. 7 shows the blanking circuit 150 according to a preferred embodiment of the present invention. The blanking circuit 150 includes an inverter 203, a transistor 205, a current source 201, a capacitor 207, and a buffer 209. An input of the inverter 203 is supplied with the PWM signal V_PWM. An output of the inverter 203 is connected to a gate of the transistor 205. A source of the transistor 205 is connected to the ground reference. An input terminal of the current source 201 is supplied with the supply voltage V_CC. An output terminal of the current source 201 is connected to a drain of the transistor 205 and an input of the buffer 209. The capacitor 207 is connected between the drain of the transistor 205 and the ground reference. The blanking signal V_BLK is generated from an output of the buffer 209.

[0052] As the PWM signal V_PWM goes logic-high, the transistor 205 will be turned off. Therefore, the current source 201 will start to charge up the capacitor 207. Once the voltage across the capacitor 207 exceeds a logic-high threshold, the buffer 209 will output a logic-high blanking signal V_BLK. The current source 201 and the capacitor 207 are connected such that a blanking delay time is inserted after the PWM signal V_PWM becomes logic-high, and before the blanking signal V_BLK becomes logic-high.

[0053] FIG. 8 shows a preferred embodiment of the reference-voltage generator 160. The reference-voltage generator 160 includes an operation amplifier 138, three transistors 137, 132 and 131, a current source 133, and two resistors 136 and 135. A positive input of the operation amplifier 138 is supplied with the sample-voltage V_SH. An output terminal of the operation amplifier 138 is connected to a gate of the transistor 137. A negative input of the operation amplifier 138 is connected to a source of the transistor 137. The resistor 136 is connected between the source of the transistor 137 and the ground reference. A gate of the transistor 132, a gate of the transistor 131, a drain of the transistor 132, and a drain of the transistor 137 are tied together. An input terminal of the current source 133 is supplied with the supply voltage V_CC. An output terminal of the current source 133 is connected to a source of the transistor 132 and to a source of the transistor 131. The resistor 135 is connected from a drain of the transistor 131 to the ground reference.

[0054] The sample-voltage V_SH and the resistor 136 determine a current I₁₃₇ flowing through the transistor 137. Since the transistor 132 and the transistor 131 build a current mirror, a current I₁₃₁ flowing through the transistor 131 will be proportional to the current I₁₃₇. The amplitude of the current I₁₃₇ can be expressed as:

[0055] $I_{137}=\frac{V_{\mathrm{SH}}}{R_{136}}$
Where R₁₃₆ is the resistance of the resistor 136.

[0056] The resistor 135 will convert the current I₁₃₁ to the power-reference voltage V_R2. Therefore, as the following equation shows, the power-reference voltage V_R2 will vary in proportion to the sample-voltage V_SH:

[0057] $V_{\mathrm{R2}}=I_{131}\times R_{135}=N\times I_{137}\times R_{135}=N\times \frac{V_{\mathrm{SH}}}{R_{136}}\times R_{135}\Rightarrow V_{\mathrm{R2}}\propto V_{\mathrm{SH}}$
Where N is the mirror ratio of the current mirror built by transistors 131 and 132.

[0058] FIG. 9 shows the sample-hold circuit 155 according to a preferred embodiment of the present invention. The sample-hold circuit 155 includes a current source 231, a transistor 235, a capacitor 240, an inverter 250, an inverter 252, a current source 232, a transistor 236, a capacitor 241, an inverter 251, an AND gate 260, a switch 220, and a capacitor 225.

[0059] A gate of the transistor 235 and an input of the inverter 252 are both supplied with the PWM signal V_PWM. An input terminal of the current source 231 is supplied with the supply voltage V_CC. An output terminal of the current source 231 is connected to a drain of the transistor 235. A source of the transistor 235 is connected to the ground reference. The drain of the transistor 235 is connected to an input of the inverter 250. The capacitor 240 is connected from the drain of the transistor 235 to the ground reference. An output of the inverter 250 is connected to a gate of the transistor 236. An input terminal of the current source 232 is supplied with the supply voltage V_CC. An output terminal of the current source 232 is connected to a drain of the transistor 236. A source of the transistor 236 is connected to the ground reference. The drain of the transistor 236 is connected to an input of the inverter 251. The capacitor 241 is connected from the drain of the transistor 236 to the ground reference. A first input of the AND gate 260 is connected to an output of the inverter 251. A second input of the AND gate 260 is connected to the drain of the transistor 235. A third input of the AND gate 260 is connected to an output of the inverter 252. An output of the AND gate 260 supplies a sample-control signal SMP to a control terminal of the switch 220. An input terminal of the switch 220 is connected to the voltage-detection terminal VDET of the PWM controller 10 for receiving a voltage-detection signal V_DT. The capacitor 225 is connected from an output terminal of the switch 220 to the ground reference. The sample-voltage V_SH is supplied from the output terminal of the switch 220 whenever the control terminal of the switch 220 is enabled by the output of the AND gate 260.

[0060] FIG. 10 shows the internal signal waveforms of the sample-hold circuit 155. Once the PWM signal V_PWM drops to logic-low, as FIG. 9 shows, the transistor 235 will be turned off. The current source 231 will start to charge up the capacitor 240. Before the voltage across the capacitor 240 exceeds the logic-high threshold, the inverter 250 will output a logic-high signal to turn on the transistor 236. This will discharge the capacitor 241 and produce a logic-high signal at the first input of the AND gate 260. The sample-control signal SMP generated from the output of the AND gate 260 will be logic-low. Therefore, the switch 220 will remain open.

[0061] Once the voltage across the capacitor 240 exceeds the logic-high threshold, the transistor 236 will be turned off. The sample-control signal SMP generated from the output of the AND gate 260 will become logic-high. This will turn on the switch 220 and enable the voltage-detection signal V_DT to charge up the capacitor 225 via the switch 220. Meanwhile, the current source 232 will start to charge up the capacitor 241. Once the voltage across the capacitor 241 exceeds the logic-high threshold, the inverter 251 will supply a logic-low signal to the first input of the AND gate 260. The sample-control signal SMP will then drop to logic-low and turn off the switch 220. The voltage across the capacitor 225 will then be equivalent to the sample-voltage V_SH.

[0062] As FIG. 10 shows, the duration of a delay time t_D1 is determined by the current source 231 and the capacitor 240. The duration of a delay time t_D2 is determined by the current source 232 and the capacitor 241.

[0063] FIG. 11 shows the multiplier 190 according to a preferred embodiment of the present invention. The multiplier 190 includes a constant current source 300, an operation amplifier 310, five transistors 350, 351, 358, 357 and 356, an operation amplifier 320, and a resistor 325. The multiplier 190 further includes a first input terminal supplied with a voltage V₁, a second input terminal supplied with a voltage V₂, and an output terminal for generating a voltage V₃.

[0064] A negative input of the operation amplifier 310 is connected to the first input terminal of the multiplier 190. An input terminal of the constant current source 300 is supplied with the supply voltage V_CC. An output terminal of the constant current source 300 is connected to a positive input of the operation amplifier 310 and a drain of the transistor 350. A gate of the transistor 350, a gate of the transistor 351, and an output terminal of the operation amplifier 310 are tied together. A source of the transistor 350 and a source of the transistor 351 are both connected to the ground reference. A drain of the transistor 351 and a drain of the transistor 358 are connected to the output terminal of the multiplier 190. A source of the transistor 358 and a source of the transistor 357 are supplied with the supply voltage V_CC. A gate of the transistor 358, a gate of the transistor 357, a drain of the transistor 357, and a drain of the transistor 356 are tied together. A positive input of the operation amplifier 320 is connected to the second input terminal of the multiplier 190. A negative input of the operation amplifier 320, a source of the transistor 356, and a first terminal of the resistor 325 are tied together. A second terminal of the resistor 325 is connected to the ground reference. An output terminal of the operation amplifier 320 is connected to a gate of the transistor 356.

[0065] The transistors 350 and 351 both operate in the linear region. When operating in the linear region, MOS transistors behave as resistors. Modulating the operating voltage and the operating current bias can control the equivalent resistance of the transistor. The operation amplifier 310 controls the resistance of the transistor 350 in response to its operating bias, which is determined by the voltage V₁ and a constant current I₃₀₀ that is provided by the constant current source 300. Since the transistor 351 is mirrored from the transistor 350, the resistance R₃₅₁ of the transistor 351 will be the same as the resistance of the transistor 350.

[0066] $\begin{array}{cc}{R}_{351}=\frac{V_1}{I_{300}}& \left(7\right)\end{array}$

[0067] The voltage V₂ and the resistor 325 determine the amplitude of the current I₃₅₆ flowing through the transistor 356. Since the transistor 358 is mirrored from the transistor 357, a current I₃₅₈ flowing through the transistor 358 will be proportional to the amplitude of the current I₃₅₆ that flows through the transistors 357 and 356.

[0068] $I_{358}\propto I_{356}\Rightarrow I_{358}\propto \frac{V_2}{R_{325}}$
Where R₃₂₅ is the resistance of the resistor 325.

[0069] The current I₃₅₈ and the resistance of the transistor 351 will produce a voltage V₃ at the drain of the transistor 351. As the following equation shows, the voltage V₃ will be proportional to the product of the voltage V₁ and the voltage V₂.

[0070] $\begin{array}{cc}{V}_3=I_{358}\times R_{351}=\frac{V_2}{R_{325}}\times \frac{V_1}{I_{300}}=K_0\times V_1\times V_2& \left(8\right)\end{array}$

[0071] FIG. 12 shows the transfer function block diagram of the power-mode controlled power converter. The block H(s) represents the low-pass filter. Av represents the gain of the error amplifier 126. The I_IN-err means the switching current errors caused by the variations of components, the propagation delay time T_D and etc. As shown in FIG. 12, the switching current I_IN can be expressed as:

[0072] $\begin{array}{cc}{I}_{\mathrm{IN}}=I_{\mathrm{IN}-\mathrm{err}}+\frac{V_{\mathrm{LIMIT}}}{R_S}& \left(9\right)\\ I_{\mathrm{IN}}=I_{\mathrm{IN}-\mathrm{err}}+\frac{1}{R_S}\times \left\{\mathrm{Av}\times V_{\mathrm{R2}}-\mathrm{Av}\times \left[\alpha \times V_{\mathrm{IN}}\times H\left(s\right)\times I_{\mathrm{IN}}\right]\right\}& \left(10\right)\end{array}$
Where α is the attenuation factor of the attenuator 180.

[0073] $\begin{array}{cc}{I}_{\mathrm{IN}}=\frac{I_{\mathrm{IN}-\mathrm{err}}+\left(\mathrm{Av}\times \frac{V_{\mathrm{R2}}}{R_s}\right)}{1+\frac{\mathrm{AV}\times \alpha \times H\left(s\right)\times V_{\mathrm{IN}}}{R_s}}& \left(11\right)\end{array}$
When Av is much greater than 1, and V_R2/R_S is greater than I_IN-err, the switching current I_IN can be rewritten as:

[0074] $\begin{array}{cc}{I}_{\mathrm{IN}}=\frac{V_{\mathrm{R2}}}{\alpha \times H\left(s\right)\times V_{\mathrm{IN}}}& \left(12\right)\end{array}$
Therefore, programming the power-reference voltage V_R2 and the attenuation factor α can precisely control the power delivered from the primary-side circuit to the secondary-side circuit of the power converter, as following equation shows:

[0075] $\begin{array}{cc}{P}_{\mathrm{IN}}=V_{\mathrm{IN}}\times I_{\mathrm{IN}}=\frac{V_{\mathrm{R2}}}{\alpha \times H\left(s\right)}& \left(13\right)\end{array}$

[0076] It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the present invention without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the present invention cover modifications and variations of this invention provided they fall within the scope of the following claims and their equivalents.