Primary-side controlled flyback power converter

[00002] 1. Field of Invention

[00003] The present invention relates to a switching mode power converter and more particularly relates to a flyback power converter.

[00004] 2. Description of Related Art

[00005] A flyback power converter typically includes a PWM controller, a power MOSFET (Metal Oxide Semiconductor Field Effect Transistor), a transformer, and a feedback-control circuit. The purpose of the feedback-control circuit is to sense the output voltage and/or the output current of the secondary-side of the power supply, and to supply a feedback signal to the PWM controller through an isolated device such as an optical-coupler.

[00006] FIG. 1 shows a traditional flyback power converter. Although this circuit is able to regulate the output voltage and output current, it has several drawbacks. One drawback of this circuit is its large size, due to the need for an optical-coupler and a secondary feedback-control circuit. Another drawback of this flyback power converter is high power consumption. To maintain a constant secondary-side output current, this circuit includes a current-sense resistor. The current-sense resistor significantly increases the power consumption of the power converter.

[00007] In recent years, several primary-side control schemes for flyback power converters have been proposed. These prior-art primary-side control schemes have attempted in various ways to reduce the size and the cost of flyback power converters. One prior-art primary-side control scheme is “Switching Power Supply Packages” by Arthur J. Collmeyer, Mark D. Telefus, Dickson T. Wong, and David B. Manner (U.S. Pat. No. 6,434,021). Although this circuit is able to regulate the output voltage and the output current, it has several drawbacks. One drawback of this prior-art invention is that the feedback control voltage is sensed from a high voltage source. This method results in a loss of accuracy, and it increases the cost of the controller. Another drawback is that the voltage drop of the output rectifier is not compensated for. Generally, the voltage drop of the output rectifier decreases in response to temperature increases, typically at a rate of 2 mV/° C. Thus, the output voltage of this prior-art invention will deviate significantly from a constant DC level.

[00008] Another prior-art control scheme is “Method and Apparatus Providing a Multi-Function Terminal for a Power Supply Controller” by Balu Balakrishnan, Alex B. Djenguerian, and Leif O. Lund (U.S. Pat. No. 6,538,908). The drawback of this prior-art is that the optical-coupler and the secondary feedback circuit are still required for loop control. Otherwise, the output voltage and the output current will fluctuate significantly.

[00009] Reflected voltage control has also been proposed as a means for primary-side control. Two prior-art patents teaching this method include “Switched Mode Power Supply Responsive to Voltage across Energy Transfer Element” by Balu Balakrishnan, David Michael, and Hugh Matthews (U.S. Pat. No. 6,233,161) and “Switched Mode Power Supply Responsive to Current Derived from Voltage across Energy Transfer Element Input” by Balu Balakrishnan, David Michael, and Hugh Matthews (U.S. Pat. No. 6,480,399).

[00010] One principal drawback of these two prior-arts is inaccurate feedback control. In order to generate a feedback control signal, the reflected voltage of the transformer is filtered and turned into a DC voltage (or current) through a resistor-capacitor circuit. However, this reflected voltage signal is not directly proportional to the output voltage, because of the spike voltage generated by the leakage inductance of the transformer. Thus, the output voltage of this prior-art invention will deviate significantly from a constant DC level. Furthermore, the voltage drop of the output rectifier is not compensated for in the feedback loop. When load changes occur, this problem will introduce additional distortion into the output voltage.

[00011] Another drawback of these two prior-art inventions is high power consumption. The reflected voltage is filtered to supply power for PWM control. However, the resistor in the filter burns the majority of the reflected power, even if the power consumption required for PWM control is low. Therefore, the power consumption of the power supply is high.

[00012] Thus, a need still remains for an efficient primary-side flyback power converter with a well regulated, constant, output voltage and output current.

[00013] A principal object of the present invention is to provide a flyback power converter under primary-side PWM control. The flyback power converter according to the present invention supplies a well-regulated constant voltage and constant current output. However, unlike prior-art PWM controllers, it does not require a secondary-side feedback circuit or an optical-coupler. This enables the device count, the size, and the cost of the power supply to be reduced.

[00014] A further object of the present invention is to solve the drawbacks of the foregoing prior-art inventions.

[00015] A further object of the present invention is to reduce power consumption. To achieve this, the present invention provides a PWM controller with a power supply that has a low-voltage source.

[00016] A further object of the present invention is to improve the DC output voltage accuracy and reduce the cost of the PWM controller. To achieve this, the present invention uses a low voltage input to detect the output voltage.

[00017] Another object of the present invention is to further improve DC output voltage accuracy. The present invention includes a double sample amplifier that precisely acquires a sampled voltage from a flyback voltage of the transformer, in a manner such that the sampled voltage is compensated and tightly-correlated to the output voltage.

[00018] The primary-side PWM controller according to the present invention can provide a well-regulated output voltage and output current. This allows the device count, the size, and the cost of the power converter to be greatly reduced.

[00019] The flyback power converter according to the present invention includes the PWM controller that generates a PWM signal to drive a switching transistor. The PWM signal is generated in response to the sampled voltage.

[00020] The flyback energy from the primary winding of the transformer is recycled to reduce power consumption. After the falling-edge of the PWM signal, the flyback energy of the primary winding is rectified and filtered to supply DC power to the PWM controller. This flyback energy includes the flyback voltage reflected from the secondary winding and an induced voltage caused by the leakage inductance.

[00021] Another feature of the flyback power converter according to the present invention is a pulse generator. The pulse generator of the PWM controller generates sampling pulses after each delay time interval. The sampling pulses are used to accurately detect the flyback voltage of the transformer. The delay time is inserted to avoid interference from the induced voltage created by the leakage inductance of the transformer. The double sample amplifier samples the flyback voltage just before the amplitude of the transformer current falls to zero. This method of sampling compensates for load-related voltage variations across the output rectifier.

[00022] Another feature of the flyback power converter according to the present invention is an offset current source. The offset current source is used to compensate for the temperature dependence of the output rectifier. This allows the PWM controller to be supplied with a more accurate feedback signal. The offset current source of the PWM controller pulls an offset current from an input of the double sample amplifier in response to the operating temperature of the flyback power converter. The offset current will produce a voltage drop across a detection resistor to compensate for voltage variations across the output rectifier.

[00023] Another feature of the flyback power converter according to the present invention is a blanking circuit. The blanking circuit produces a blanking time to ensure that the on-time of the PWM signal will create a sufficient delay to precisely sample the flyback voltage.

[00024] Another feature of the flyback power converter according to the present invention is the use of an oscillator in the PWM controller to produce a progranunable switching frequency. The programmable switching frequency is produced in response to the voltage sampled from the flyback voltage. The programmable switching frequency is used by the flyback power converter to control the amount of power delivered from the primary-side of the transformer to the output of the power converter.

[00025] It is to be understood that both the foregoing general descriptions and the following detailed descriptions are exemplary, and are intended to provide further explanation of the invention as claimed. Still further objects and advantages will become apparent from a consideration of the ensuing description and drawings.

[00037] FIG. 1 shows a traditional flyback power converter. A capacitor 34 is connected to a PWM controller 90 and is charged via a resistor 22. The PWM controller 90 will be started up once its supply voltage V_CC is higher than a start-threshold voltage. When the PWM controller 90 starts to operate, it will output a PWM signal to drive a switching transistor 80 and a transformer 40. Meanwhile, an auxiliary winding N_A of the transformer 40 supplies the supply voltage V_CC via a rectifier 14. A current-sense resistor 25 converts a switching current of the transformer 40 into a voltage signal for PWM control and over-power protection. An output of an optical-coupler 92 supplies a feedback voltage V_FB.

[00038] The output voltage V_O and the Zener voltage of a Zener diode 96 drive an input of the optical-coupler 92 via a resistor 28 to form the feedback loop. The magnitude of the feedback voltage V_FB of the PWM controller 90 determines the on-time (T_ON) of the PWM signal and regulates the output power. A transistor 84 coupled with a current-limit resistor 86 control the maximum amplitude of the output current I_O. As the output current I_O increases, the voltage across the current-limit resistor 86 increases as well. When this voltage exceeds the junction voltage of the transistor 84 (e.g., 0.7V), the transistor 84 will be turned on. This reduces the on-time of the PWM signal by decreasing the feedback voltage V_FB. In this manner, the output current I_O of the power supply is kept constant.

[00039] FIG. 2 shows a flyback power converter according to the present invention. The flyback power converter supplies a constant output voltage and a constant current output under primary-side PWM control. An input VIN of the power converter is connected to a drain of a switching transistor 80. A first primary winding N_P1 and a second primary winding N_P2 are connected in series to construct a primary winding of a transformer 50. A first terminal of the primary winding is a first terminal of the first primary winding N_P1, and a second terminal of the primary winding is a second terminal of the second primary winding N_P2. A second terminal of the first primary winding N_P1 is connected to a first terminal of the second primary winding N_P2. The transformer 50 further comprises a secondary winding N_S.

[00040] The secondary winding N_S of the transformer 50 is connected to an output of the power converter via an output rectifier 19. A source of the switching transistor 80 is connected via a current-sense resistor 25 to the first terminal of the primary winding of the transformer 50. The second terminal of the primary winding is connected to the ground reference. When power is applied to the input V_IN of the power converter, a capacitor 35 is charged up via a start-up resistor 20. The capacitor 35 is connected to a supply-voltage input VCC of a PWM controller 100. The capacitor 35 stores energy that is used by the PWM controller 100.

[00041] Once the voltage at the supply-voltage input VCC of the PWM controller 100 exceeds the start-threshold voltage, the PWM controller 100 will start to operate and generate a PWM signal V_PWM. The PWM signal V_PWM will drive a gate of the switching transistor 80 to deliver energy to the output of the power converter. At the instant the PWM signal turns off, a flyback voltage will be reflected from the secondary winding N_S to the first primary winding N_P1 and the second primary winding N_P2. The voltage across the secondary winding N_S is equal to the sum of the voltage drop across the output rectifier 19 and the output voltage V_O of the power converter.

[00042] The PWM controller 100 has a detection input VS for sampling the flyback voltage V_P1 from the first primary winding N_P1. This is used to regulate the output voltage V_O. The PWM controller 100 regulates the output voltage of the power converter by modulating the PWM signal V_PWM in response to the voltage sampled at the detection input VS. While the PWM signal V_PWM is on, a primary current of the transformer 50 will produce a current-sense voltage V_IS across the current-sense resistor 25. A capacitor 37 is connected from the COM input of the PWM controller 100 to the ground reference. The capacitor 37 is used for frequency compensation to stabilize the control loop of the power converter.

[00043] A sense-junction connects the source of the switching transistor 80 and the current-sense resistor 25. The sense junction is connected to a current-sense input IS of the PWM controller 100. The current-sense input IS detects the current-sense voltage V_IS, so that the PWM controller 100 can successfully limit the peak value of the primary current flowing through the transformer 50.

[00044] An anode of a diode 15 is connected to the second terminal of the first primary winding N_P1. A cathode of the diode 15 is connected to the supply-voltage input VCC of the PWM controller 100. After the PWM signal V_PWM turns off, the rectified flyback voltage of the first primary winding N_P1 is supplied to the supply-voltage input VCC of the PWM controller 100 via the diode 15. The capacitor 35 filters this rectified flyback voltage to supply the PWM controller 100 with a DC power source.

[00045] One component of the voltage supplied to the supply-voltage input VCC of the PWM controller 100 is from the flyback voltage reflected from the secondary winding N_S. However, the voltage supplied to the supply-voltage input VCC of the PWM controller 100 also includes an induced voltage from the leakage inductance of the transformer 50. This flyback energy of the first primary winding N_P1 of the transformer 50 is recycled to power the PWM controller 100, thus reducing power consumption.

[00046] The flyback energy of the second primary winding N_P2 of the transformer 50 is not utilized. Consequently, to eliminate the induced voltage caused by the leakage inductance of the second primary winding N_P2 of the transformer 50, a snubber circuit is connected in parallel with the second primary winding N_P2. The snubber circuit comprises a diode 17 connected in series with a voltage-clamping device 47. The voltage-clamping device can either be a Zener diode or a TVS (Transient Voltage Suppressor).

[00047] FIG. 3 shows a preferred embodiment of the PWM controller 100 including a double sample amplifier (DSA) 200, an oscillator 300, a PWM circuit 500, a pulse generator 700, a zero current detection (ZCD) comparator 900, and a threshold voltage 910. When the PWM signal V_PWM is off, the pulse generator 700 generates a sampling pulse V_SP1 and a sampling pulse V_SP2 alternately. The sampling pulse V_SP1 and the sampling pulse V_SP1 are generated following a delay time T_d. The delay time T_d is chosen such that the flyback voltage of the first primary winding N_P1 of the transformer 50 can be sampled. The delay time T_d is needed to avoid sampling the induced voltage from the leakage inductance of the transformer 50.

[00048] Once the ZCD comparator 900 detects a zero current status from the transformer 50, the pulse generator 700 will be inhibited to generate the sampling pulse V_SP1 and the sampling pulse V_SP2. In the mean time, the feedback signals sampled during the sampling pulses V_SP1 and V_SP2 are stored in a signal buffer of the double sample amplifier 200.

[00049] Before initiating the next switching cycle, the pulse generator 700 will produce a sampling pulse V_SP3 for acquiring a sampled voltage V_SH from the signal buffer. In response to the sampled voltage V_SH, the double sample amplifier 200 will generate a clamped-sample voltage V_HSH for supplying to the oscillator 300. In order to eliminate the possibility of sampling a falling edge signal, only the higher voltage in the signal buffer is taken as the sampled voltage V_SH. The available sampled voltage V_SH is acquired just before the transformer current drops to zero. More specifically, the available sampled voltage V_SH is acquired when the current of the secondary winding N_S of the transformer 50 drops to a minimum value. Sampling the flyback voltage V_P1 during a minimum secondary current reduces the error margin from the voltage drop of the output rectifier 19. The ZCD comparator 900 outputs a ZCD signal V_ZCD to indicate a zero current status once the voltage at the detection input VS drops below a lower-limit voltage. The lower-limit voltage is the threshold voltage 910 plus an output voltage V_H of the signal buffer. The flyback voltage V_P1 is sampled through a detection resistor 23 shown in FIG. 2.

[00050] The sampled voltage V_SH is used for voltage regulation. One problem is that the voltage drop across the output rectifier 19 varies with respect to temperature. In order to compensate for this, an offset current source of the PWM controller 100 pulls an offset current I_M from the detection input VS. The offset current I_M is decreased in proportion to the increase of the temperature. The offset current I_M produces a voltage drop across the detection resistor 23 so that the voltage drop is proportional to the voltage drop across the output rectifier 19. By properly selecting the resistance of the detection resistor 23, it is possible to accurately offset the variation of the voltage drop across the output rectifier 19. When the variation of the voltage drop across the detection resistor 23 is correlated to the variation of the voltage drop across the output rectifier 19, the detection resistor 23 can adequately compensate for the temperature coefficient of the output rectifier 19. In this manner, the flyback power converter according to the present invention can supply a well-regulated output voltage over a wide range of operating temperatures.

[00051] As shown in FIG. 3, the oscillator 300 of the PWM controller 100 produces the programmable switching frequency in response to the clamped-sample voltage V_HSH, and thus controls the power delivered from the primary-side of the transformer to the output of the power converter. The clamped-sample voltage V_HSH is correlated to the output voltage V_O. Thus, the switching frequency will vary in proportion to the output voltage V_O. With a programmable switching frequency, the flyback power converter according to the present invention can maintain a constant output current I_O.

[00052] The PWM controller 100 generates the PWM signal V_PWM from a feedback voltage V_COM, a limit voltage V_LIMIT, and the current-sense voltage V_IS. The PWM signal V_PWM is used for PWM control. The PWM signal is generated such that the output voltage and the output current are both well regulated.

[00053] FIG. 4 shows a preferred embodiment of the double sample amplifier 200 of the PWM controller 100 according to the present invention. In this embodiment, the detection input VS is connected to an input terminal of a switch 220 and to an input terminal of a switch 221. The switch 220 is turned on/off by the sampling pulse V_SP1. An output terminal of the switch 220 is connected to a capacitor 230 to produce a hold voltage V_H1. The capacitor 230 is further connected to a positive terminal of an operational amplifier 246. The switch 221 is turned on/off by the sampling pulse V_SP2. An output terminal of the switch 221 is connected to a capacitor 231 to produce a hold voltage V_H2. The capacitor 231 is connected in parallel with a switch 225. The PWM signal V_PWM will control the switch 225 to discharge the capacitor 231 during each switching cycle. The capacitor 231 is further connected between a positive terminal of an operational amplifier 245 and the ground reference. The operational amplifier 245, the operational amplifier 246, the capacitor 230, the capacitor 231, a diode 266, and a diode 265 develop the signal buffer of the double sample amplifier 200. The signal buffer outputs the voltage V_H. Via the diodes 266 and 265, an output of the operational amplifier 246 and an output of the operational amplifier 245 are connected to the output of the signal buffer. A negative terminal of the operational amplifier 246 and a negative terminal of the operational amplifier 245 are also connected to the output of the signal buffer. Therefore, the magnitude of the signal buffer output voltage V_H is equal to the maximum of the voltages V_H1 and V_H2.

[00054] The double sample amplifier 200 of the PWM controller 100 further includes a current source 285 connected from the output of the signal buffer to the ground reference. This is done in order to pull the signal buffer signal low. To acquire the sampled voltage V_SH from the output of the signal buffer, a switch 222 is turned on/off by the sampling pulse V_SP3. An input terminal of the switch 222 is connected to the output of the signal buffer. A capacitor 232 for holding the sampled voltage V_SH is connected to an output terminal of the switch 222. The capacitor 232 is further connected to a positive terminal of an operational amplifier 241. The output of the operational amplifier 241 is connected to a first terminal of a resistor 252. A second terminal of the resistor 252 outputs the clamped-sample voltage V_HSH. A negative terminal of the operational amplifier 241 is connected to the second terminal of the resistor 252. The operational amplifier 241 keeps the clamped-sample voltage V_HSH equal to the sampled voltage V_SH, as long as the clamped-sample voltage V_HSH is higher than a minimum voltage value V_MIN. A current source 281 is connected to the second terminal of the resistor 252. The current source 281 and the resistor 252 are used to produce the minimum voltage value V_MIN for the clamped-sample voltage V_HSH.

[00055] The double sample amplifier 200 of the PWM controller 100 further includes an operational amplifier 240 having a positive terminal connected to a reference voltage terminal V_RV. An output of the operational amplifier 240 drives a gate of a transistor 213. A negative terminal of the operation amplifier 240 is connected to a source of the transistor 213. The source of the transistor 213 is further connected to the second terminal of the resistor 252 via a resistor 250. A drain of the transistor 213 is connected to a current mirror. The current mirror comprises a transistor 215 and a transistor 217. The transistor 215 is an input of the current mirror. A drain of the transistor 217 is connected to a resistor 251, to produce the feedback voltage V_COM at the COM input of the PWM controller 100. In this manner as described above, an error amplifier is provided for the voltage feedback loop of the PWM controller 100. The resistor 250 determines the gain of the error amplifier, and the capacitor 37 shown in the FIG. 2 determines the bandwidth of the error amplifier.

[00056] The double sample amplifier 200 of the PWM controller 100 further includes a current source 280 and a programmable current generator 290 connected in series. A junction of the current source 280 and the programmable current generator 290 is connected to an input of the offset current source. The offset current source comprises a diode 260, a transistor 210 and a transistor 211. A drain of the transistor 210 is the input of the offset current source. To sink the offset current I_M, the detection input VS is connected to a drain of the transistor 211. The programmable current generator 290 generates a programmable current I_T that is inversely proportional to temperature variation. Therefore, the amplitude of the offset current I_M is inversely proportional to temperature variation, such that the offset current I_M decrease in response to temperature increases.

[00057] FIG. 5 shows a preferred embodiment of the programmable current generator 290 that generates the programmable current I_T in response to temperature variation. The programmable current generator 290 comprises bipolar transistors 291 and 292, p-mirror transistors 294, 295 and 296, n-mirror transistors 297 and 298 and a resistor 293. The programmable current I_T is given by, $\begin{array}{cc}{I}_T=N_M\times \frac{k\times T_{\mathrm{emp}}}{q}\times \frac{\ln \left(r\right)}{R_T}& \left(1\right)\end{array}$
where R_T is the resistance of the resistor 293; N_M=M₁×M₂; M₁ is the geometrical ratio of the transistor 295 and 296; M₂ is the geometrical ratio of the transistor 297 and 298; k is the Boltzmann's constant; q is the charge on an electron; r is the emitter area ratio of the bipolar transistor 291 and 292; and T_temp is the absolute temperature.

[00059] FIG. 6 shows a preferred embodiment of the oscillator 300 of the PWM controller 100 according to the present invention. In this embodiment, the clamped-sample voltage V_HSH is supplied to a positive input of an operational amplifier 340. The operational amplifier 340 is coupled to a transistor 310 and a resistor 360, to generate a variable charge current I₃₁₀. A transistor 311 paired with a transistor 312 produce a first discharge current I₃₁₂ by mirroring the variable charge current I₃₁₀. The pair of transistors 311 and 314 produces a charge current I₃₁₄ by mirroring the variable charge current I₃₁₀. A transistor 315 paired with a transistor 316 produces a second discharge current I₃₁₆ by mirroring the first discharge current I₃₁₂. A switch 351 is used to enable and disable the charge current I₃₁₄. A switch 352 is used to enable and disable the second discharge current I₃₁₆. The charge current I₃₁₄ and the second discharge current I₃₁₆ are supplied to a capacitor 365, to generate a saw-tooth signal. A comparator 345, a comparator 346, a reference voltage V₂, a reference voltage V₁, a NAND-gate 381, and a NAND-gate 382, are coupled to generate a clock signal V_CLK to control the switch 352. The clock signal V_CLK also controls the switch 351 via an inverter 380.

[00060] FIG. 7 shows a preferred embodiment of the PWM circuit 500 of the PWM controller 100 according to the present invention. The PWM circuit 500 comprises a comparator 545, a comparator 546, a NAND-gate 510, a NAND-gate 511, a flip-flop 515, an inverter 512, an AND-gate 519, and a blanking circuit 520. A negative input of the comparator 545 and a negative input of the comparator 546 are connected to the current-sense input IS of the PWM controller 100. The comparator 545 is used to compare the feedback voltage V_COM with the current-sense voltage V_IS. The comparator 546 is used to compare a limit voltage V_LIMIT with the current-sense voltage V_IS. The limit voltage V_LIMIT is used to limit the peak primary current of the transformer 50. An output of the comparator 545 is connected to a first input of the NAND-gate 510. An output of the comparator 546 is also connected to a second input of the NAND-gate 510. A first input of the NAND-gate 511 is connected to an output of the NAND-gate 510. A second input of the NAND-gate 511 is connected to an output of the blanking circuit 520. The clock signal V_CLX is supplied to a clock input of the flip-flop 515 via the inverter 512, which is further connected to a first input of the AND-gate 519. The flip-flop 515 is reset by an output of the NAND-gate 511. An input of the flip-flop 515 is supplied from the supply voltage V_CC. An output of the flip-flop 515 connects to a second input of the AND-gate 519. An input of the flip-flop 515 is supplied from the supply voltage V_CC. An output of the AND-gate 519 supplies the PWM signal V_PWM to an input of the blanking circuit 520. The blanking circuit 520 generates a blanking signal V_BLK to ensure a minimum on-time for the PWM signal V_PWM.

[00061] FIG. 8 shows a preferred embodiment of the blanking circuit 520 of the PWM circuit 500 according to the present invention. The blanking circuit 520 comprises an inverter 521, an inverter 522, a NAND-gate 523, a transistor 526, a capacitor 527, and a current source 525. The purpose of the blanking circuit 520 is to generate the blanking signal V_BLK. The PWM signal V_PWM is supplied to an input of the inverter 521 and a first input of the NAND-gate 523. The transistor 526 is coupled with the current source 525, the capacitor 527, and the inverter 522, to produce a blanking time T_BLK. An output of the inverter 521 drives a gate of the transistor 526 to start the blanking time T_BLK, once the PWM signal V_PWM is on. An output of the inverter 522 is connected to a second input of the NAND-gate 523. An output of the NAND-gate 523, which is an output of the blanking circuit 520, generates the blanking signal V_BLK. Its waveform is shown in FIG. 10.

[00062] FIG. 9 shows a preferred embodiment of the pulse generator 700 of the PWM controller 100 according to the present invention. To produce a sampling clock signal the pulse generator 700 includes a capacitor 751, an inverter 736, a switch 737, a current source 710, a current source 711, and a hysteresis buffer 735. The capacitor 751 is charged by the current source 710 via the switch 737, and is discharged by the current source 711. The capacitor 751 is connected to an input of the hysteresis buffer 735. The hysteresis buffer 735 outputs a sampling clock signal to control the switch 737 via the inverter 736. The current source 710 and the current source 711 respectively determine an on-time and an off-time of the sampling clock signal. The on-time of the sampling clock signal further determines the sample time T_S1 for both of the sampling pulse signals V_SP1 and V_SP2. To produce a delay time T_d after the PWM signal V_PWM goes off, the pulse generator 700 includes a transistor 741, a current source 712, a capacitor 752, an inverter 721, and an inverter 722. The PWM signal V_PWM is also supplied to an input of the inverter 721. The output of the inverter 721 drives the inverter 722. The inverter 722 further drives a gate of the transistor 741. When the PWM signal V_PWM goes off, the transistor 741 is also turned-off. The current source 712 will then charges the capacitor 752 to produce the delay time T_d.

[00063] The pulse generator 700 of the PWM controller 100 further comprises an AND-gate 726 having three inputs. A first input is supplied by the sampling clock signal, a second input is connected to the capacitor 752, and a third input is connected to the output of the inverter 721. An output of the AND-gate 726 produces the sampling pulses V_SP1 and V_SP2 via an AND-gate 733 and an AND-gate 732 respectively. The AND-gates 732 and 733 are alternately enabled by a flip-flop 731. The sampling clock signal is supplied to a clock input of the flip-flop 731. An output and an inverse-output of the flip-flop 731 are connected to the AND-gates 732 and 733 respectively, thus the sampling pulses V_SP1 and V_SP2 can be alternately produced. Furthermore, in order to sample the flyback voltage just before the transformer current drops to zero, the ZCD signal V_ZCD controls the AND-gates 732 and 733 via an OR-gate 729.

[00064] However, to ensure that the double sample amplifier 200 will produce the sampled voltage V_SH during every switching cycle, the ZCD signal V_ZCD disables the sampling pulses V_SP1 and V_SP2 after they are produced.

[00065] The inverse-output of a flip-flop 730 connects to a first input of the OR-gate 729. A second input of the OR-gate 729 is supplied by the ZCD signal V_ZCD. The flip-flop 730 is reset by the output of the inverter 721. A clock input of the flip-flop 730 is connected to the output of the AND-gate 726 via an inverter 725. Thus, the ZCD signal V_ZCD will disable the AND-gates 732 and 733 following either of the sampling pulses, V_SP1 or V_SP2.

[00066] To produce a sampling pulse V_SP3 with a pulse width equal to a sample time T_S2, the pulse generator 700 includes a transistor 742, a current source 714, a capacitor 753, an inverter 723, an inverter 724, and an AND-gate 727. The clock signal V_CLK is supplied to an input of the inverter 723 and a first input of the AND-gate 727. An output of the inverter 723 is connected to a gate of the transistor 742 to control the start of the sample time T_S2. The current source 714 and the capacitor 753 are coupled to produce the sample time T_S2. The capacitor 753 drives a second input of the AND-gate 727 via the inverter 724. The AND-gate 727 outputs the sampling pulse V_SP3 with a pulse width equal to the sample time T_S2.

[00067] FIG. 10 shows the timing diagram of the PWM circuit 500 and the pulse generator 700. After the PWM signal goes low, the sampling pulses V_SP1 and V_SP1 are alternately generated, following the delay time T_d. The purpose of the delay time T_d is to eliminate as much as possible the influence of the leakage inductance of the transformer 50. To accomplish this, the delay time T_d is inserted during each cycle between the falling-edge of the PWM signal V_PWM and the beginning of the flyback voltage sampling process. The blanking circuit 520 shown in FIG. 7 produces the blanking time T_BLK. This determines the minimum on-time of the PWM signal, once the PWM signal is on. The delay time T_d is determined by the minimum on-time of the PWM signal. The blanking time T_BLK ensures that the delay time T_d and the sample time T_S1 and T_S2 are sufficient to precisely sample the flyback voltage. When a zero current status is detected at the detection input VS, both the sampling pulses V_SP1 and V_SP2 will be disabled. The flyback voltage will be sampled and held by the capacitors 230 and 231, depicted in FIG. 4. The sampling pulse V_SP3 is then generated in response to the clock signal V_CLK. When the sampling pulse V_SP3 is generated, the capacitor 232 will further sample and hold the higher voltage stored by the capacitors 230 or 231. Referring now to FIG. 2, the output voltage V_O of the power converter can be expressed as, $\begin{array}{cc}{V}_O=V_{\mathrm{NS}}-V_D=\frac{N_S}{N_{\mathrm{P1}}}\times V_{\mathrm{P1}}-V_D=\frac{N_S}{N_{\mathrm{P1}}}\times \left(V_{\mathrm{SH}}+I_M\times R_{23}\right)-V_D& \left(2\right)\end{array}$
where V_NS is the voltage across the secondary winding N_S, V_P1 is the voltage of the first primary winding, V_D is the voltage drop across the output rectifier 19, V_SH is the voltage sampled at the detection input VS, and R₂₃ is the resistance of the detection resistor 23. The output voltage V_O can also be expressed in terms of the PWM feedback-control circuit:
V_O=G_M×(V_RV−V_SH)  (3)
where G_M is the loop gain of the PWM feedback-control circuit and V_RV is the reference voltage of the double sample amplifier 200. Based on equations (2) and (3), the output voltage can be rewritten as, $V_O=\left\{\frac{N_S}{N_{\mathrm{P1}}}\times V_{\mathrm{RV}}+\left[\left(\frac{N_S}{N_{\mathrm{P1}}}\times I_M\times R_{23}\right)-V_D\right]\right\}÷\left\{1+\left(\frac{N_S}{N_{\mathrm{P1}}}\times \frac{1}{G_M}\right)\right\}$
Because G_M>>1, V_O can be expressed in a simplified form as, $\begin{array}{cc}{V}_O=\frac{N_S}{N_{\mathrm{P1}}}\times V_{\mathrm{RV}}+\left[\left(\frac{N_S}{N_{\mathrm{P1}}}\times I_M\times R_{23}\right)-V_D\right]& \left(4\right)\end{array}$

[00072] The flyback voltage is sampled just before the transformer current drops to zero. The problem is that the voltage drop across the output rectifier 19 varies with respect to temperature.

[00073] To compensate for this, the present invention introduces the offset current I_M. The offset current I_M is modulated in inverse proportion to temperature variation. By properly selecting the resistance of the detection resistor 23, it is possible to offset the adverse temperature effect of the voltage drop across the output rectifier 19. Thus, the flyback power converter according to the present invention can supply a well-regulated output voltage V_O.

[00074] The oscillator 300 produces the clock signal V_CLK with a programmable switching frequency. The switching frequency is determined in response to the clamped-sample voltage V_HSH. This controls the power delivered from the primary-side of the transformer to the output of the power converter. Because the clamped-sample voltage V_HSH is correlated to the output voltage V_O, the switching frequency will be proportional to output voltage V_O. Using the programmable switching frequency, the flyback power converter according to the present invention can maintain a constant output current I_O.

[00075] Since the output power is a function of the output voltage V_O, a constant current output can be achieved when the output current of the power converter is always less than a maximum value. The output power P_O is given by, $\begin{array}{cc}{P}_O=\mathrm{Vo}\times \mathrm{Io}=\eta \times P_{\mathrm{IN}}=\eta \times \frac{1}{2\times T}\times L_P\times {\mathrm{Ip}}^2& \left(5\right)\end{array}$
where PIN is the power input to the primary-side, η is the power conversion efficiency, T is the switching period, L_P is the primary inductance of the transformer 50, and I_P is the primary current of the transformer 50.

[00077] The primary current I_P produces the current-sense voltage V_IS across the current-sense resistor 25. The current-sense resistor 25 is connected to the current-sense input IS of the PWM controller 100. Once the current-sense voltage V_IS exceeds the limit voltage V_LIMIT, the logic circuit of the PWM controller 100 will turn off the PWM signal to keep the primary current I_P constant. Referring to equation (5), the output current I_O of the power converter can be shown as, $\begin{array}{cc}{I}_O=\frac{1}{\mathrm{Vo}}\times \frac{\eta }{2\times T}\times L_P\times {\left(I_P\right)}^2& \left(6\right)\end{array}$

[00078] In order to produce a constant output current I_O, when V_O2=0.5×V_O1 (P_O2=0.5×P_O1), T should be increased as follows:
T₂=2×T₁
where the first switching period T₁ refers to the first output voltage V_O1, and the second period of the switching frequency T₂ refers to the second output voltage V_O2.

[00081] The oscillator 300 generates the clock signal V_CLK with a programmable switching frequency. The switching frequency is determined in response to the clamped-sample voltage V_HSH. When the clamped-sample voltage V_HSH exceeds the minimum voltage V_MIN, the clamped-sample voltage V_HSH is equal to the sampled voltage V_SH. If the clamped-sample voltage V_HSH goes below the minimum voltage value V_MIN, the switching frequency of the oscillator 300 will be insufficient to operate the PWM controller 100. To prevent this from happening, the clamped-sample voltage V_HSH will be set to the minimum voltage value V_MIN, whenever it starts to dip below V_MIN.

[00082] Referring to equation (2), the sampled voltage V_SH is per se a function of the output voltage V_O. Therefore a constant output current can be easily achieved by reducing the switching period to (2×T), whenever the output voltage is decreased to (0.5×V_O).

[00083] As described above, the flyback power converter includes the PWM controller 100, to generate the offset current I_M and the sampled voltage V_SH. The sampled voltage V_SH is generated during every PWM cycle in response to the flyback voltage sampled across the first primary winding. In this manner, the flyback power converter according to the present invention can keep the output voltage constant. To limit the power transferred through the primary winding, the switching frequency is generated in response to the output voltage V_O. In this manner, the flyback power converter according to the present invention keeps the output current constant.

[00084] It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the present invention without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the present invention cover modifications and variations of this invention provided they fall within the scope of the following claims and their equivalents.