Apparatus for comparing an input voltage with a threshold voltage

[0001] The present invention is directed to apparatuses for comparing an input voltage with a reference or threshold voltage. Such apparatuses are sometimes referred to as comparators. The present invention is preferably embodied in a bandgap comparator apparatus that compares an input voltage with an inherent threshold voltage presented, or established by the apparatus.

[0002] A bandgap comparator generates a digital output that transitions high when an input voltage exceeds an internally generated reference or threshold voltage. One problem with typical prior art bandgap comparator apparatuses is that they draw excessive current when the input voltage exceeds the threshold voltage.

[0003] There is a need for an apparatus for comparing an input voltage with a threshold voltage, such as a bandgap comparator, that reduces current drain without compromising accuracy of comparator thresholds.

[0004] A preferred embodiment of an apparatus for comparing an input voltage with a threshold voltage includes: (a) a first current mirror device that includes a first bipolar transistor with a first base and a first collector; the first base and the first collector establish a diode-connected first collector; the input voltage is received at the first current mirror device; (b) a second current mirror device that includes a second bipolar transistor with a second base and a second collector; the second base and the second collector establish a diode-connected second collector; (c) a first impedance coupled in series with the diode-connected first collector and the diode-connected second collector; and (d) a second impedance coupled between ground and the second current mirror device. The first current mirror device and the second current mirror device are further coupled with an output locus. Output signals appearing at the output locus indicate relative voltage levels of the input voltage and the threshold voltage.

[0005] It is, therefore, an object of the present invention to provide an apparatus for comparing an input voltage with a threshold voltage, such as a bandgap comparator, that reduces current drain without compromising accuracy of comparator thresholds.

[0006] Further objects and features of the present invention will be apparent from the following specification and claims when considered in connection with the accompanying drawings, in which like elements are labeled using like reference numerals in the various figures, illustrating the preferred embodiments of the invention.

[0013] FIG. 1 is an electrical schematic diagram of a representative prior art comparator apparatus. In FIG. 1, a prior art comparator apparatus 10 is based upon a Brokaw bandgap cell, as generally described in “A Simple Three-Terminal IC Bandgap Reference,” by A. Paul Brokaw; IEEE Journal of Solid-State Circuits; December 1974; pp. 388-393. Comparator apparatus 10 includes a first current mirror 12, a second current mirror 14 and a third current mirror 16. First current mirror 12 includes metal oxide semiconductor (MOS) transistors M₁, M₂; second current mirror 14 includes MOS transistors M₃, M₄; and third current mirror 16 includes MOS transistors M₅, M₆. MOS transistor M₁ has a source 20, a gate 22 and a drain 24. MOS transistor M₂ has a source 26, a gate 28 and a drain 30. MOS transistor M₂ has gate 28 coupled with drain 30 to establish the MOS transistor pair M₁, M₂ as current mirror 12.

[0014] MOS transistor M₃ has a source 32, a gate 34 and a drain 36. MOS transistor M₄ has a source 38, a gate 40 and a drain 42. MOS transistor M₃ has gate 34 coupled with drain 36 to establish the MOS transistor pair M₃, M₄ as current mirror 14.

[0015] MOS transistor M₅ has a source 44, a gate 46 and a drain 48. MOS transistor M₆ has a source 50, a gate 52 and a drain 54. MOS transistor M₅ has gate 46 coupled with drain 48 to establish the MOS transistor pair M₅, M₆ as current mirror 16. Drain 24 is coupled with drain 48. Drain 42 is coupled with drain 54. A supply voltage V_CC is provided at a supply locus 56 that is coupled with sources 20, 26, 32, 38. Source 44 is coupled with ground 82. Source 50 is coupled with ground 84.

[0016] Comparator apparatus 10 also includes a sensing unit 60 and a scaling unit 62. Sensing unit 60 includes a pair of NPN bipolar transistors Q₁, Q₂. Transistor Q₂ has a collector 64, a base 66 and an emitter 68. Transistor Q₁ has a collector 70, a base 72 and an emitter 74. Emitter area A_e2 of emitter 68 of transistor Q₂ is larger than emitter area A_e1 of emitter 74 of transistor Q₁ by a factor of N. That is:

 A_e2=N·A_e1  [1]

[0017] Bases 66, 72 are coupled together. An input locus 76 is coupled with bases 66, 72. Resistors R₁, R₂ are coupled with input locus 76 and with an input node 78. Resistors R₁, R₂ scale input signals V_IN applied to input node 78 for presentation at input locus 76.

[0018] Scaling unit 62 includes a resistor R₃ coupled with emitter 68 and with emitter 74, and a resistor R₄ coupled between emitter 74 and ground 80.

[0019] An output locus 86 is coupled with drains 42, 54; an output voltage V_OUT appears at output locus 86 when voltage at input locus 76 exceeds a predetermined threshold. The threshold determination is inherent in comparator apparatus 10 and is related to the base-emitter voltage V_be of transistor Q₁.

[0020] The base-emitter voltage of a bipolar transistor V_be equals: $\begin{array}{cc}{V}_{\mathrm{be}}=V_T\ln \left(\frac{I_C}{A_e{J}_s}\right)& \left[2\right]\end{array}$

[0021] Where V_T is the thermal voltage,

• • [0022] I_C is the collector current,
  • [0023] A_e is the emitter area, and
  • [0024] J_S is the saturation current density.

[0025] If one assumes that current mirrors 12, 14, 16 all implement 1:1 current ratios, then a balance point is attainable at which currents through transistors M1, M2, M3, M4 are equal. That is, referring to FIG. 1,
I_M1=I_M2=I_M3=I_M4  [3]

[0026] In such a balance point condition, collector currents I_C1, I_C2 of transistors Q₁, Q₂, respectively, are equal. That is, referring to FIG. 1,
I_C1=I_C2  [4]

[0027] The voltage across resistor R₃ is therefore equal to the difference between base-emitter voltages V_be1, V_be2 for transistors Q₁, Q₂, respectively. That is,
ΔV_be=V_be1−V_be2  [5]

[0028] Substituting expression [2] into expression [5]: $\begin{array}{cc}\Delta V_{\mathrm{be}}=V_T\ln \left(\frac{I_{\mathrm{C1}}}{A_{\mathrm{e1}}{J}_s}\right)-V_T\ln \left(\frac{I_{\mathrm{C2}}}{A_{\mathrm{e2}}{J}_s}\right)& \left[6\right]\end{array}$

[0029] Combining terms in expression [6], and canceling out the term J_S: $\begin{array}{cc}\Delta V_{\mathrm{be}}=V_T\ln \left(\frac{I_{\mathrm{C1}}{A}_{\mathrm{e2}}}{I_{\mathrm{C2}}{A}_{\mathrm{e1}}}\right)& \left[7\right]\end{array}$

[0030] Recalling that in balance point conditions I_C1=I_C2, expression [4] becomes, $\begin{array}{cc}\Delta V_{\mathrm{be}}=V_T\ln \left(\frac{A_{\mathrm{e2}}}{A_{\mathrm{e1}}}\right)& \left[8\right]\end{array}$

[0031] Recalling further that A_e2=N·A_e1, expression [8] becomes,

 ΔV_be=V_T ln(N)  [9]

[0032] The voltage at input locus 76 may be regarded as the bandgap voltage V_bg for comparator apparatus 10. If one ignores base currents (for the sake of simplicity) one may observe that: $\begin{array}{cc}{V}_{\mathrm{bg}}=2V_T\frac{R_4}{R_3}\ln \left(N\right)+V_{\mathrm{be2}}& \left[10\right]\end{array}$

[0033] The factor (2V_T) is necessary because currents from both transistors Q₁, Q₂ flows through resistor R₄, thus doubling the total voltage seen there. Resistors R₃, R₄ are derived from matched devices, so any temperature dependencies they may possess are equal and therefore cancel. Emitter area ratio N is substantially independent of temperature. The thermal voltage V_T equals: $\begin{array}{cc}{V}_T=\frac{\mathrm{kT}}{q}& \left[11\right]\end{array}$

[0034] Where k is Boltzmann's constant,

• • [0035] T is absolute temperature, and
  • [0036] q is the charge on an electron.

[0037] Since k and q are temperature-invariant fundamental constants, the thermal voltage V_T depends linearly upon absolute temperature. The first term on the right side of expression [10] therefore has a linear positive temperature coefficient. The second term on the right side of expression [10] is base-emitter voltage of transistor Q₁. The base-emitter voltage of a silicon bipolar transistor has a negative temperature coefficient of approximately −2 millivolts per degree Kelvin $\left(-\frac{2mV}{K}\right).$
The base-emitter voltage of a bipolar transistor varies nonlinearly with temperature. If this relationship is expanded as a power series upon temperature, the linear term will be found to dominate. If resistors R₃, R₄ are properly selected, the positive temperature coefficient of the first term of expression [10] will exactly cancel the linear contribution of the second term of expression [10]. Under these conditions, bandgap voltage V_bg becomes substantially independent of temperature. If transistors Q₁, Q₂ are fabricated from silicon, then the value of bandgap voltage V_bg for which the temperature coefficient is smallest typically equals about 1.25 volts, a value that is indirectly related to the bandgap voltage of silicon. It is for this reason that voltage V_bg is referred to as the bandgap voltage.

[0038] Adjustment of a threshold voltage at which the output of comparator apparatus 10 transition may be effected by appropriate selection of values for resistors R₁, R₂. Thus, a threshold voltage V_th at input node 78 can be established by the relation: $\begin{array}{cc}{V}_{\mathrm{th}}=\frac{R_1+R_2}{R_2}\left[2V_T\frac{R_4}{R_3}\ln \left(N\right)+V_{\mathrm{be2}}\right]& \left[12\right]\end{array}$

[0039] If one assumes that V_bg≡1.25 volts, then $\begin{array}{cc}{V}_{\mathrm{th}}\cong 1.25\left(\frac{R_1+R_2}{R_2}\right)& \left[13\right]\end{array}$

[0040] That is, comparator apparatus 10 may be configured to switch (i.e., to present an output signal V_OUT at output locus 86) at any threshold greater than approximately 1.25 volts by appropriately selecting values for resistors R₁, R₂.

[0041] As mentioned briefly earlier herein, a significant problem with comparator apparatus 10 is that it draws significant current when input voltage V_IN at input node 78 exceeds threshold voltage V_th. Comparator apparatus 10 has five circuit limbs, so its minimum quiescent supply current I_Q at the threshold is 5·I_min, where I_min is the minimum current that can be conducted through a path from supply to ground without experiencing significant variations due to junction leakage, thermal noise or other parasitics. I_Q increases rapidly at higher input voltages. There is a need for a lower current comparator apparatus. A good way to realize this goal is to reduce the number of paths through which current flows from supply to ground.

[0042] FIG. 2 is an electrical schematic diagram of a comparator apparatus constructed according to the teachings of the present invention. In FIG. 2, a comparator apparatus 110 includes a first current mirror 112 and a second current mirror structure 114. Second current mirror structure 114 is coupled with an impedance, preferably a resistor R₂, to establish a current generating circuit 162. First current mirror 112 includes bipolar PNP transistors Q₁, Q₂; second current mirror 114 includes bipolar NPN transistors Q₃, Q₄. Bipolar transistor Q₁ has an emitter 120, a base 122 and a collector 124. Bipolar transistor Q₂ has an emitter 126, a base 128 and a collector 130. Bipolar transistor Q₃ has an emitter 164, a base 166 and a collector 168. Bipolar transistor Q₄ has an emitter 170, a base 172 and a collector 174. Base 122 of bipolar transistor Q₁ is coupled with collector 124 of transistor Q₁ and base 128 of transistor Q₂ to establish first current mirror 112. Base 166 of bipolar transistor Q₃ is coupled with collector 168 of transistor Q₃ and base 172 of transistor Q₄ to establish second current mirror structure 114. Resistor R₂ is coupled with emitter 170 of transistor Q₄ and to ground 180 to establish current generating circuit 162.

[0043] Comparator apparatus 110 also includes scaling unit or element 161. Scaling element 161 determines the threshold voltage V_th of comparator unit 110, as will be described hereinafter. Scaling element 161 includes an impedance, preferably a resistor R₁ coupled in series between collector 124 and collector 168. Emitter area A_e2 of emitter 170 of transistor Q₄ is larger than emitter area A_e1 of emitter 164 of transistor Q₃ by a factor of N. That is, from expression [1]:
A_e2=N·A_e1  [1]

[0044] Emitter 164 is coupled with ground 182. Collectors 130, 174 are coupled together. An input locus 178 is coupled with emitters 120, 126. An output locus 186 is coupled with collectors 130, 174; an output voltage V_OUT appears at output locus 186 when voltage at input locus 178 exceeds a predetermined threshold. The threshold determination is inherent in comparator apparatus 110 and is related to the base-emitter voltages V_be of transistors Q₁, Q₃.

[0045] At threshold voltage V_th, current mirror 112 ensures that collector currents are equal at collectors 124, 130. In such circumstances, the voltage across impedance R₂ (from expression [9]) is:
ΔV_be=V_T ln(N)  [9]

[0046] As a consequence, current through transistor Q₂ (and, thus, current through transistor Q₁) equals: $\begin{array}{cc}{I}_{\mathrm{C2}}=\frac{V_T\ln \left(N\right)}{R_2}& \left[14\right]\end{array}$

[0047] Where I_C2 is the current through collector 130 of transistor Q₂.

[0048] Thus, at the balance point, the threshold voltage V_th must equal: $\begin{array}{cc}{V}_{\mathrm{th}}=V_T\frac{R_1}{R_2}\ln \left(N\right)+2V_{\mathrm{be}}& \left[15\right]\end{array}$

[0049] If threshold voltage V_th is set to equal twice the bandgap voltage V_bg, then the temperature dependence of the threshold voltage V_th is minimized. In such a configuration, comparator apparatus 110 acts as a bandgap comparator with a threshold of approximately 2.5 volts, a figure that is associated with the semiconductor material used in manufacturing transistors Q₁, Q₂, Q₃, Q₄. For purposes of this illustrative example, transistors Q₁, Q₂, Q₃, Q₄ are presumed to have been manufactured in silicon. In contrast to comparator apparatus 10 (FIG. 1), comparator apparatus 110 has only two circuit limb currents, so its minimum quiescent supply current I_Q at the threshold is 2·I_min. Thus, comparator apparatus 110 is a lower current comparator apparatus than is comparator apparatus 10 (FIG. 1).

[0050] Output impedance characteristics of comparator apparatus 110 may be improved by adding impedances 119, 125 in series with emitters 120, 126, thereby improving accuracy of threshold switching. Impedances 119, 125 are indicated in dotted line format to indicate their characterization as an alternate embodiment.

[0051] Comparator apparatus 110 has a fixed threshold voltage V_th=2·V_bg. Providing an impedance 121 connected between base 122 and emitter 120 of transistor Q₁ allows selection of a threshold voltage V_th>2·V_bg. Adjusting of threshold voltage V_th is effected by appropriate selection of the value of the impedance 121. Impedance 121 is indicated in dotted line format to indicate its characterization as an alternate embodiment.

[0052] Comparator apparatus 110 does not provide a definite output signal when V_IN is less than 2·V_be. This may be corrected by injecting a small startup current I_start from a current source 165 connected with base 166 of transistor Q₃. So long as startup current I_start is much smaller than minimum current I_min in the various current limbs of comparator apparatus 110, startup current I_start will have little effect upon threshold voltage V_th of comparator apparatus 110. With the addition of current source 165, the output 186 will be asserted for input voltages V_IN as low as one V_be. Current source 165 can be implemented as a depletion-mode FET (field effect transistor), a pinched-off JFET (junction field effect transistor), or similar structure. Current source 165 is indicated in dotted line format to indicate its characterization as an alternate embodiment.

[0053] FIG. 3 is an electrical schematic diagram of a second embodiment of a comparator apparatus constructed according to the teachings of the present invention. In FIG. 3, a comparator apparatus 210 is substantially similar to comparator apparatus 110 (FIG. 2) with additional structure that permits establishing higher threshold voltages. Comparator apparatus 210 includes a first current mirror 212 and a second current mirror structure 214. Second current mirror structure 214 is coupled with an impedance, preferably a resistor R₂ to establish a current generating circuit 262. First current mirror 212 includes bipolar PNP transistors Q₁, Q₂; second current mirror structure 214 includes bipolar NPN transistors Q₃, Q₄. Bipolar transistor Q₁ has an emitter 220, a base 222 and a collector 224. Bipolar transistor Q₂ has an emitter 226, a base 228 and a collector 230. Bipolar transistor Q₃ has an emitter 264, a base 266 and a collector 268. Bipolar transistor Q₄ has an emitter 270, a base 272 and a collector 274. Base 222 of bipolar transistor Q₁ is coupled with collector 224 of transistor Q₁ and base 228 of transistor Q₂ to establish first current mirror 212. Base 266 of bipolar transistor Q₃ is coupled with collector 268 of transistor Q₃ and base 272 of transistor Q₄ to establish second current mirror structure 214. Resistor R₂ is coupled with emitter 270 of transistor Q₄ and with ground 280 to establish current generating circuit 262.

[0054] Comparator apparatus 210 also includes a scaling unit or element 261. Scaling element 261 determines the threshold voltage V_th of comparator unit 210. Scaling element 261 includes an impedance, preferably a resistor R₁ coupled in series between collector 224 and collector 268. Emitter area A_e2 of emitter 270 of transistor Q₄ is larger than emitter area A_e1 of emitter 264 of transistor Q₃ by a factor of N. That is:
A_e2=N·A_e1  [16]

[0055] Emitter 264 is coupled with ground 282. Collectors 230, 274 are coupled together. An input locus 278 is coupled with emitters 220, 226. An output locus 286 is coupled with collectors 230, 274; an output voltage V_OUT appears at output locus 286 when voltage at input locus 278 exceeds a predetermined threshold. The threshold determination is inherent in comparator apparatus 210 and is related to the base-emitter voltages V_be of transistors Q₁, Q₃. Threshold determination is also further related to diode elements 290a, 290b, . . . 290n coupled in series with impedance R₁. Diode elements 290n each add a voltage drop substantially equal to V_be to threshold determinations for comparator apparatus 210. Diode elements 290n are illustrated as coupled between impedance R₁ and collector 268. Diode elements 290n could as well be coupled in series with impedance R₁ between collector 224 and impedance R₁. Diode elements 290n could be employed with the alternate embodiments indicated in comparator apparatus 110 (FIG. 2) in dotted line format.

[0056] FIG. 4 is an electrical schematic diagram of a circuit segment that may be employed for constructing a third embodiment of a comparator apparatus according to the teachings of the present invention. In FIG. 4, a circuit segment 300 is illustrated for effecting adjustment of threshold voltage V_th. Circuit segment 300 includes an NPN bipolar transistor 302 having a collector 304, a base 306 and an emitter 308 and connection terminals 310, 312. Resistors R_A, R_B affect the voltage drop V_d between terminals 310, 312 as: $\begin{array}{cc}{V}_d=\frac{R_A+R_B}{R_B}·V_{\mathrm{be}}& \left[17\right]\end{array}$

[0057] Circuit segment 300 may be included for providing threshold level control, for example, in a comparator apparatus such as comparator apparatus 210 (FIG. 3) using connection terminals 310, 312 to couple circuit segment 300 in series with resistor R₁. The advantage of circuit segment 300 is that it allows continuous adjustment of the threshold voltage V_th rather than adjustment by discrete steps, as is the case when using diodes 290n (FIG. 3). Circuit segment 300 could be employed with the alternate embodiments indicated in comparator apparatus 110 (FIG. 2) in dotted line format.

[0058] FIG. 5 is an electrical schematic diagram of a circuit segment that may be employed for constructing a fourth embodiment of a comparator apparatus according to the teachings of the present invention. In FIG. 5, a circuit segment 400 is illustrated for effecting contribution to threshold voltage V_th. Circuit segment 400 includes an NPN bipolar transistor 402 having a collector 404, a base 406 and an emitter 408 and connection termini 410, 412. Collector 404 and base 406 are coupled together to configure collector 404 as a diode-connected collector. Circuit segment 400 may be included for contributing to threshold level, for example, in a comparator apparatus such as comparator apparatus 210 (FIG. 3) using connection terminals 410, 412 to couple circuit segment 400 in series with in series with impedance R₁ as a substitute for a diode element 290n (FIG. 3) or in addition to a diode element 290n. Circuit segment 400 could be employed with the alternate embodiments indicated in comparator apparatus 110 (FIG. 2) in dotted line format or with circuit segment 300 (FIG. 4).

[0059] FIG. 6 is an electrical schematic diagram of a fifth embodiment of a comparator apparatus constructed according to the teachings of the present invention. In FIG. 6, a comparator apparatus 510 is substantially similar to comparator apparatus 110 (FIG. 2) with additional structure that provides hysteresis to avoid multiple transitions of the output V_OUT during operation of comparator apparatus 510 when input V_IN is near threshold voltage V_th. Comparator apparatus 510 includes a first current mirror 512 and a second current mirror structure 514. Second current mirror structure 514 is coupled with an impedance, preferably a resistor R₂ to establish a current generating circuit 562. First current mirror 512 includes bipolar PNP transistors Q₁, Q₂; second current mirror structure 514 includes bipolar NPN transistors Q₃, Q₄. Bipolar transistor Q₁ has an emitter 520, a base 522 and a collector 524. Bipolar transistor Q₂ has an emitter 526, a base 528 and a collector 530. Bipolar transistor Q₃ has an emitter 564, a base 566 and a collector 568. Bipolar transistor Q₄ has an emitter 570, a base 572 and a collector 574. Base 522 of bipolar transistor Q₁ is coupled with collector 524 of transistor Q₁ and base 528 of transistor Q₂ to establish first current mirror 512. Base 566 of bipolar transistor Q₃ is coupled with collector 568 of transistor Q₃ and base 572 of bipolar transistor Q₄ to establish second current mirror structure 514. Resistor R₂ is coupled to emitter 570 of transistor Q₄ and to ground 580 to establish current generating circuit 562.

[0060] Comparator apparatus 510 also includes a scaling unit or element 561. Scaling element 561 determines the threshold voltage V_th of comparator unit 510. Scaling unit 561 includes an impedance, preferably a resistor R₁ coupled in series between collector 524 and collector 568. Emitter area A_e2 of emitter 570 of transistor Q₄ is larger than emitter area A_e1 of emitter 564 of transistor Q₃ by a factor of N. That is:
A_e2=N·A_e1  [18]

[0061] Emitter 564 is coupled with ground 582. Collectors 530, 574 are coupled together. An input locus 578 is coupled with emitters 520, 526. An output locus 586 is coupled with collectors 530, 574; an output voltage V_OUT appears at output locus 586 when voltage at input locus 578 exceeds a predetermined threshold. The threshold determination is inherent in comparator apparatus 510 and is related to the base-emitter voltages V_be of transistors Q₁, Q₃.

[0062] A hysteresis circuit 600 includes a digital buffer 612 for providing an indication of output voltage V_OUT to a MOS transistor 604. Digital buffer 612 could be constructed, for example, by a series connection of two CMOS (complementary metal oxide semiconductor) inverters. MOS transistor 604 includes a drain 606, a gate 608 and a source 610. Drain 606 is coupled in common with resistor R₁ and a second impedance, preferably a resistor R₃. Source 610 is coupled with connection 567 that commonly couples base 566 with collector 568 of bipolar transistor Q₁. When output 586 is low, hysteresis circuit 600 operates to raise the threshold voltage V_th. Because output 586 is low, transistor 604 does not conduct and resistor R₃ is placed in series with resistor R₁. This, in turn, increases threshold voltage V_th, as represented by expression [15]. When output 586 is high, transistor 604 conducts, effectively shorting resistor R₃ and reducing threshold voltage V_th. The amount of hysteresis V_hys provided by hysteresis circuit 600 equals: $\begin{array}{cc}{V}_{\mathrm{hys}}=\frac{R_3·V_t\ln \left(N\right)}{R_2}& \left[19\right]\end{array}$

[0063] Therefore, hysteresis circuit 600 can be adjusted to provide an arbitrarily small hysteresis. A small hysteresis is desirable because it is not possible to adjust both thresholds of apparatus 510 (FIG. 6) to zero temperature coefficients. So long as the hysteresis is less than or equal to about 50 mV, this shortcoming has little impact upon the operation of apparatus 510. Hysteresis circuit 600 could be employed with the alternate embodiments indicated in comparator apparatus 110 (FIG. 2) in dotted line format, or with circuit segment 300 (FIG. 4) or with circuit segment 400 (FIG. 5).

[0064] It is to be understood that, while the detailed drawings and specific examples given describe preferred embodiments of the invention, they are for the purpose of illustration only, that the apparatus of the invention is not limited to the precise details and conditions disclosed and that various changes may be made therein without departing from the spirit of the invention which is defined by the following claims: