Low-distortion transimpedance amplifier structures and methods

[0002] 1. Field of the Invention

[0003] The present invention relates generally to transimpedance amplifiers.

[0004] 2. Description of the Related Art

[0005] Transimpedance amplifiers typically deliver a single-ended output voltage signal in response to a differential input current signal. For effective use in a variety of systems (e.g., cable communication systems and optical transmission systems), they must generally drive low-impedance loads and provide output signals with extremely low distortion (e.g., harmonic spectral lines reduced <50 dB below the carrier).

[0006] In order to meet these requirements, the output circuit of an exemplary upstream cable modem amplifier comprises a differential pair of transistors and a 2:1 (voltage ratio) transformer. A resistor across the collectors of the differential pair provides a nominal output impedance of 300 ohms and the transformer is required to provide a single-ended output, to match this output to a 75 ohm load and to maintain an acceptable bandwidth. The output circuit of another exemplary transimpedance amplifier requires a complex feedback system that comprises a number of diodes, resistors, capacitors and transistors.

[0007] In contrast to these exemplary amplifiers, a preferred transimpedance amplifier should provide a low-distortion output voltage signal across low-impedance loads and be formed with simple, small, inexpensive and reliable structures that are compatible with integrated-circuit fabrication processes.

[0008] The present invention is directed to transimpedance amplifiers that generate low-distortion output voltage signals with simple, inexpensive structures that are compatible with integrated-circuit fabrication processes.

[0009] These amplifiers include a current processor and a complementary output stage. The processor provides in-phase upper and lower current signals S_uprand S_lwrin response to a differential input current signal S_diffand alters amplitudes of these signals in response to a common-mode input current signal S_cm. The complementary output stage has upper and lower transistors that provide the output voltage signal in respective response to the upper and lower current signals and with distortion that is reduced by the altered first and second amplitudes. The common-mode input current signal S_cmcan be set to a predetermined value.

[0010] The novel features of the invention are set forth with particularity in the appended claims. The invention will be best understood from the following description when read in conjunction with the accompanying drawings.

[0011] FIG. 1 is a block diagram of a transimpedance amplifier of the present invention and of an embodiment of a current processor in the amplifier;

[0012] FIG. 2 is a schematic of another embodiment of the transimpedance amplifier of FIG. 1;

[0013] FIG. 3 is a flow chart that illustrates transimpedance process steps that can be practiced with the amplifiers of FIGS. 1 and 2; and

[0014] FIG. 4 is a block diagram of a programmable amplifier of the present invention that includes the transimpedance amplifier of FIG. 1.

[0015] FIG. 1 illustrates a low-distortion transimpedance amplifier 20 that generates an output voltage signal S_outat an output port 22 in response to a differential input current signal S_diffand a common-mode input current signal S_cmthat are applied at a differential input port 24. The transimpedance amplifier includes a current processor 26 that receives the signals S_diffand S_cmand a complementary output stage 28 that generates the output voltage signal S_outin response to the current processor.

[0016] In particular, the current processor 26 responds to the differential input current signal S_diffby providing in-phase upper and lower current signals S_uprand S_lwrthat drive the complementary output stage 28. The current processor 26 also responds to the common-mode input current signal S_cmby differentially altering respective amplitudes of the upper and lower current signals S_uprand S_lwrto thereby reduce harmonic distortion in the complementary output stage 28. If desired, the common-mode input current signal S_cmcan be set to a predetermined value that best reduces the distortion.

[0017] Accordingly, the transimpedance amplifier 20 replaces prior art output structures (e.g., transformers) that have undesirable characteristics (e.g., expensive, large size and unreliable) with complementary transistor structures that have desirable characteristics (e.g., inexpensive, small, reliable and compatible with integrated circuit processes) and provides structure and method that realizes low harmonic distortion.

[0018] FIG. 1 also shows a current processor embodiment 26A which can be substituted in the transimpedance amplifier 20 as indicated by substitution arrow 30. This embodiment includes a current generator 34 positioned between a first current mirror 36 and a second current mirror 38. In response to the differential input current signal S_diff, the current generator provides first and second differential currents 40 and 44 wherein the first differential current 40 comprises first and second currents 41 and 42 and the second differential current 44 comprises third and fourth currents 45 and 46.

[0019] The first current mirror 36 mirrors the first current 41 to thereby generate a first mirrored current 50 that is differenced with the second current 42 at an upper current node 51. The upper current signal S_upris therefore the difference between the first mirrored current 50 and the second current 42. Similarly, the second current mirror 38 mirrors the third current 45 to thereby generate a second mirrored current 52 that is differenced with the fourth current 46 at a lower current node 53. The lower current signal S_lwris therefore the difference between the second mirrored current 52 and the fourth current 46.

[0020] The current generator 34 is biased to maintain constant internal bias currents that respectively couple the first and third currents 41 and 45 and couple the second and fourth currents 42 and 46. Accordingly, when a first side of the differential input current signal S_diffinduces a current increase δi in the first current 41, the internal current bias insures that the third current 45 has a corresponding current loss δi. Corresponding responses occur between the second current signal 42 and the fourth current signal 46 except they are substantially 180° out of phase with the responses in the first and third currents because they are responsive to the other side of the differential input current signal S_diff.

[0021] Consequently, the first and second currents 41 and 42 form the first differential current 40, the third and fourth currents 45 and 46 form the second differential current 44, the first and fourth currents 41 and 46 are substantially in phase and the second and third currents 42 and 45 are also substantially in phase.

[0022] Assuming the first and second current mirrors are 1:1 current mirrors, the first mirrored current 50 substantially equals the first current 41. A current increase δi in the first current 41, therefore, mirrors a current increase δi in the first mirrored current 50 and is accompanied by a corresponding current decrease i_δ in the second current signal 42. Consequently, a current of 2δi flows from the upper current node 51 and forms the upper current signal S_upr.

[0023] Because the second and third currents are in phase, the current decrease i_δ in the second current 42 is accompanied by a current decrease δi in the third current 45. This latter current decrease δi mirrors a current decrease δi in the second mirrored current 52 and is accompanied by a corresponding current increase δi in the fourth current signal 46. Consequently, a current of 2δi also flows from the current node 53 and forms the lower current signal S_lwr. The current processor 26A, therefore, provides in-phase upper and lower current signals S_uprand S_lwrwhose amplitudes substantially match the amplitude of the differential input current signal S_diff(i.e., their amplitudes are substantially twice the signal amplitudes at each side of the differential input port 24).

[0024] If the common-mode input current signal S_cmsources a current ΔI into the current generator 34, it will cause a current increase ΔI in the third and fourth currents 45 and 46 and, because of the constant internal bias currents, a corresponding current decrease ΔI in the first and second currents 41 and 42. The current decrease ΔI in the first current 41 mirrors a current decrease ΔI in the first mirrored current 50 so that current changes cancel at the upper current node 51 and, therefore, the common-mode input current signal S_cmdoes not directly alter the upper current signal S_upr. Similarly, the current increase ΔI in the third current 45 mirrors a current increase ΔI in the second mirrored current 52 so that current changes also cancel at the lower current node 53 and, therefore, the common-mode input current signal S_cmdoes not directly alter the lower current signal S_lwr.

[0025] However, the increased current flows in the third and fourth currents 45 and 46 effect impedance decreases in transistors associated with these currents and the decreased current flows in the first and second currents 41 and 42 effect impedance increases in transistors associated with these currents. These impedance changes steer current portions from the first differential current 40 to the second differential current 44. Accordingly, the amplitude of the lower current signal S_lwris increased and the amplitude of the upper current signal S_upris correspondingly decreased. This amplitude change can be reversed by reversing the common-mode input current signal S_cm, i.e., by sinking a current ΔI from the current generator 34 rather than inserting a current ΔI.

[0026] Transistors in the upper portion of the complementary output stage 28 differ in polarity from transistors in the lower portion and, accordingly, they differ in their production of distortion signals when responding to like amplitude signals. The current processor 26A facilitates, therefore, a distortion reduction in the output voltage signal S_outbecause it can realize an appropriate amplitude difference in the upper current signal S_uprand the lower current signal S_lwrin response to a corresponding change in the common-mode input current signal S_cm.

[0027] An understanding of the structure and operation of the transimpedance amplifier 20 is further enhanced by directing attention to the transimpedance amplifier embodiment 20A of FIG. 2. This embodiment includes a current bias generator 60, a current processor 26B and a complementary output stage 28A.

[0028] The current processor 26B positions a current generator 34A between a first current mirror 36A and a second current mirror 38A. Although the invention may be practiced with various current mirror structures, the exemplary first current mirror of FIG. 2 has a diode-connected transistor 61 coupled base-to-base with a transistor 62 and these transistors are coupled through respective resistors 63 and 64 to a supply voltage V_cc. in addition, an exemplary second current mirror is formed with similar elements 65, 66, 67 and 68 (wherein the transistors are of opposite polarity) and is coupled to GND.

[0029] The current generator 34A includes series-coupled resistors 71 and 72 that are respectively coupled to first sides of the first and second current mirrors by cascode transistors 73 and 74 and includes series-coupled resistors 75 and 76 that are respectively coupled to second sides of the first and second current mirrors by cascode transistors 77 and 78. The junction between the resistors 71 and 72 is coupled to one side of a differential input port 24 and the junction between the resistors 75 and 76 is coupled to the other side.

[0030] A junction between the cascode transistor 77 and the current mirror transistor 62 is the upper current node (51 in FIG. 1) and the junction between the cascode transistor 78 and the current mirror transistor 66 is the lower current node (53 in FIG. 1). From these nodes, the upper current S_uprand the lower current S_lwrpass through respective ports 80 and 82 to the complementary output stage 28A.

[0031] The complementary output stage 28A includes an exemplary upper current mirror 90 that has an input transistor 91 base-coupled to an output transistor 92 and these transistors are respectively coupled to V_ccthrough resistors 93 and 94. To enhance the current mirror's mirror accuracy, the input transistor 91 is diode-connected by a boost transistor 95. The complementary output stage 28A includes similar elements 100, 101, 102, 103, 104 and 105 except the transistors are of opposite polarity.

[0032] The output transistors 92 and 102 are complementarily coupled to provide the output voltage signal S_outto an output port 22 through a coupling capacitor 108. A feedback resistor 109 is coupled back to the amplifier port 80 via a resistor 110 and a cascode transistor 112 and is also coupled back to the amplifier port 82 via a resistor 114 and a cascode transistor 116.

[0033] The current bias generator 60 includes series-connected resistors 120 and 121 that are coupled between V_ccand GND by bias transistors 122 and 123 and current sources 124 and 125. The bias transistors are respectively diode-connected by boost transistors 126 and 127 which are respectively coupled to V_ccand GND. Leakage current to insure sufficient current flow through the boost transistors 126 and 127 is provided by a series connection of diode-connected transistors 128 and 129 and resistor 130. Finally, a bias voltage (e.g., V_cc/2) is established between the resistors 120 and 121 by a voltage source 132.

[0034] In operation of the transimpedance amplifier 20A, the current generator 34A establishes a bias current through resistors 120 and 121 and a bias voltage (e.g., V_cc/2) at their common junction. Cascode transistors 73 and 74, 77 and 78 and 112 and 116 form “floating current mirrors” that mirror the bias current and the bias voltage to resistors 71 and 72, 75 and 76 and 110 and 114. These mirrors maintain constant internal bias currents in the current generator 34A so that a current increase δi in cascode transistor 73 imposes a substantially equal current decrease δi in cascode transistor 74. Because of the differential current drive at the input port 24, these increases and decreases will be accompanied by a current decrease δi in cascode transistor 77 and a current increase δi in cascode transistor 78.

[0035] In response, mirror transistor 62 of the current mirror 36A will provide a current increase δi to the differencing node 51 and, accordingly, the upper current signal S_uprhas a current increase 2δi. At the same time, mirror transistor 66 of the current mirror 38A will provide a current decrease δi to the differencing node 53 and, accordingly, the lower current signal S_lwralso has a current increase 2δi. In response to the differential input current signal S_diff, therefore, the current processor 26B supplies in-phase upper and lower currents S_uprand S_lwrthat substantially equal S_diff(i.e., their amplitudes substantially equal the signal amplitudes at each side of the differential input port 24).

[0036] A positive common-mode input current signal S_cmat the input port 24 of FIG. 2 will decrease currents through cascode transistors 73 and 77 and increase currents through cascode transistors 74 and 78. Because these changes in cascode transistors 73 and 77 are in the same direction, a current cancellation will occur at the difference node 51 between the current through cascode transistor 77 and a mirrored current from mirror transistor 62. A similar current cancellation will occur at the difference node 53 and, as a result, the common-mode input current signal S_cmdoes not directly effect a direct current (dc) change in the upper and lower current signals S_uprand S_lwr.

[0037] A positive common-mode input current signal S_cmwill, however, effect a change in the input impedances of the cascode transistors 73, 74, 77 and 78. The emitter impedance of each of these transistors approximates 1/g_mwhich equals kT/qI_cwherein g_mis transistor transconductance, k is Boltzmann's constant, T is degrees Kelvin, q is electron charge and I_cis collector current. A current increase in cascode transistors 74 and 78 will, therefore, lower their emitter resistances while a current decrease in cascode transistors 73 and 77 will raise their emitter resistances.

[0038] Assuming that resistors 71, 72, 75 and 76 have equal resistance, this difference in emitter resistances steers a greater portion of the differential input current signal S_diffthrough the lower-impedance cascode transistors 74 and 78 than through the higher-impedance cascode transistors 71 and 75. Accordingly, the lower current S_lwris somewhat increased and the upper current S_upris decreased by a like amount.

[0039] Because the complementary output transistors 92 and 102 have finite Early voltages V_a, their output impedances V_a/I_c(in which I_cis instantaneous collector current) include second-order nonlinearities and this nonlinearity increases as the Early voltage V_adecreases. A substantial voltage swing generally occurs across the collectors of output transistors 92 and 102 and this voltage swing across nonlinear output impedances in combination with the output current swing causes nonlinear variations of transistor output impedance R_o. These variations result in 2nd order harmonic distortions in the output voltage signal S_outat the output port 22. These distortions increase with Early voltage V_adifferences in different transistor types.

[0040] For example, PNP transistors typically have lower Early voltages than do NPN transistors so, for the same voltage swing, output transistor 92 will produce more distortion than output transistor 102. The greater distortion of the output transistor 92 can be reduced, however, by steering a greater portion of the signal currents generated in the current processor 26B to the output transistor 102.

[0041] The invention therefore realizes that harmonic distortion in the complementary output stage 28A can be substantially controlled via the amplitudes of the upper and lower current signals S_uprand S_lwrand these amplitudes, in turn, can be controlled by the common-mode input current signal S_cm. Accordingly, harmonic distortion of the transimpedance amplifier 20A is reduced by an appropriate selection of the common-mode input current signal S_cm(which can be set, for example, to a predetermined level).

[0042] It was previously assumed that the current mirrors 90 and 100 of the complementary output stage 28A were 1:1 current mirrors. With appropriate selection of transistor sizes and resistor values, however, current mirrors 90 and 100 can be set to mirror currents that are N times greater than their corresponding input currents.

[0043] In general, therefore, the in-phase upper and lower current signals S_uprand S_lwr(prior to adjustments effected by the common-mode input current signal S_cm) are equal to (½)S_diffand currents out of the complementary output transistors 92 and 102 are equal to (N/2)S_diff. Harmonic distortion in the transimpedance amplifier 20A is reduced because these currents are differentially altered by application of an appropriate common-mode input current signal S_cmwhich changes transistor impedances in the current generator 34A.

[0044] Feedback through the feedback resistor 109 sets the output impedance Z_outof the transimpedance amplifier 20A to be substantially equal to the impedance of the feedback resistor Z₁₀₉divided by N+1 wherein N is the current gain of the current mirrors 90 and 100. Thus, the output impedance Z_outof the transimpedance amplifier 20A can be adjusted to substantially match the load impedance R_ld.

[0045] The flow chart 120 of FIG. 3 illustrates process steps that can be practiced with the transimpedance amplifiers of FIGS. 1 and 2. In particular, the flow chart 120 shows process steps in a method for generating a low-distortion output voltage signal S_outin response to a differential input current signal S_diff.

[0046] In a first process step 122, in-phase upper and lower current signals S_uprand S_lwrare provided in response to the differential input current signal S_diff. The output voltage signal S_outis generated in process step 124 with complementary upper and lower transistors that respectively respond to the upper and lower current signals S_uprand S_lwr. Finally, first and second amplitudes of the upper and lower current signals S_uprand S_lwrare altered in process step 126 to thereby reduce distortion in the output voltage signal S_out.

[0047] FIG. 4 illustrates a programmable amplifier 140 that positions a programmable attenuator 142 between an input buffer amplifier 144 and the transimpedance amplifier 20 of FIG. 1. Attenuation of the programmable attenuator is set by a digital word that is applied from a data latch 146 which receives the digital word from a shift register 148 in response to a latch signal at a latch port. Words are applied to a data port of the shift register and clocked into the shift register in response to a clock signal applied to a clock port.

[0048] The buffer amplifier 144 isolates the programmable attenuator 142 from input impedance changes and provides a low noise signal to the programmable attenuator. In response, the programmable attenuator provides, to the transimpedance amplifier, a differential input signal S_diffthat has been attenuated to a level commanded by the digital word that is provided from the latch 146. The transimpedance amplifier 20 then converts the differential input signal S_diffinto an output voltage signal S_outin previously-described processes. The programmable amplifier 140 thus comprises a low-noise fixed amplifier, a digitally-controlled attenuator and a low-distortion high-power amplifier.

[0049] The transimpedance amplifiers and programmable amplifiers of the invention provide low distortion amplification and the ability to drive single-ended low-impedance loads (e.g., a 75 ohm CATV cable) without the need for large, expensive output devices (e.g., transformers). They are especially suited for use in a variety of systems (e.g., cable communication systems and optical transmission systems) and for a variety of specific amplifier applications (e.g., as upstream power amplifiers in cable modems and CATV set-top boxes).

[0050] In the transimpedance amplifier 20A of FIG. 2, output transistors 92 and 102 (of the current mirrors 90 and 100) are arranged as a complementary common emitter output stage. Other embodiments of the invention, however, may be formed with different complementary arrangements (e.g., a complementary common collector stage).

[0051] Although embodiments of the transimpedance amplifiers of the invention have been described with respect to bipolar transistors, other embodiments can be formed with different transistors (e.g., metal oxide transistors as exemplified in FIG. 2 by the metal oxide transistor 150 and the substitution arrow 152).

[0052] The embodiments of the invention described herein are exemplary and numerous modifications, variations and rearrangements can be readily envisioned to achieve substantially equivalent results, all of which are intended to be embraced within the spirit and scope of the invention as defined in the appended claims.