Constant current source having a controlled temperature coefficient

[0002] The present invention relates to a constant current source for use in radio frequency circuits. Specifically, a current source having a controllable temperature coefficient is described.

[0003] Radio frequency circuit applications for the cellular telephone field may require circuits which can operate over a wide temperature range. In the case of a transmitter circuit for a radio telephone, it is desirable to maintain a power output characteristic constant so that the compression point is stable with temperature. However, temperature changes typically decrease the gain or transconductance of active devices in the circuit, even when current is maintained constant over temperature. The loss in gain will decrease the compression point for an amplifier biased to operate in a class A mode of operation. As the compression point decreases, increased input signal levels do not increase the output signal level proportionally. It may be desirable in some applications to increase the bias current supplied to the amplifier to offset the loss in transconductance using a current source with a controllable temperature coefficient. A current source having a small positive temperature coefficient makes it possible to maintain the device gain and improve the overall stability of the RF circuit gain, noise figure and power output over an operating temperature range.

[0004] In accordance with the invention, a current source is provided which has a temperature coefficient which can be invariant with respect to temperature, or which may provide some small selectable temperature coefficient to offset component degradation with temperature. The invention generates a bandgap voltage which is coupled to a current source. The temperature coefficient of the bandgap voltage is selected by the value of a first resistor and the value of a second resistor of the bandgap generator. The bandgap voltage applied to the current source substantially determines the level of current produced by the current source. By controlling the relative resistance values, the temperature coefficient for the current source is also established.

[0005] The FIGURE in the application illustrates a current source having a controllable temperature coefficient in accordance with a preferred embodiment of the invention.

[0006] The schematic circuit drawing of the FIGURE illustrates a bandgap voltage generator connected to a current source. The bandgap voltage generator comprises a pair of bipolar transistors 15 and 16 fed from a current mirror comprising a PFET 12 and PFET 13. The current mirror produces first and second identical currents I₁and I₂. I₁is supplied to the collector connection of NPN bipolar transistor 16, and I₂is supplied through a bipolar NPN transistor 14 to the collector connection of NPN bipolar transistor 15 of the bandgap voltage generator. Resistor 19 having a resistance value R₁is connected across the emitter connection of NPN bipolar transistors 15 and 16, and resistor 18 having resistance value R₀receives currents I₁and I₂and is connected to the common terminal 11 of the circuit. A power supply voltage is connected across terminal 10 and 11 to provide operating current for the device. The bandgap voltage generated at the base connection of NPN bipolar transistors 15 and 16 follows the general formula of:

[0007] V_Bg=V_BE+KΔV_BE

[0008] where $K=\left(\ln \frac{A_2}{A_1}\right)\frac{R_0}{R_1};$

[0009] A₂, and A₁being the area of the base-emitters junctions of transistor 15 and 16, respectively.

[0010] ΔV_Be≈kT/q_V≈VBE15−VBE16, where VBE15 and VBE16 are the base emitter voltages of transistors 15 and 16. $\begin{array}{cc}\begin{array}{c}\mathrm{since}{V}_{\mathrm{BE1}}=V_{T}l\frac{I_1}{A_1{I}_2}\mathrm{and}\\ V_{\mathrm{BE2}}=V_{T}l\frac{I_2}{A_2{I}_1},\mathrm{then}\Delta V_{\mathrm{BE}}=V_T\ln \frac{A_2}{A_1}\end{array}& \left(1\right)\end{array}$

[0011] The current through the collector emitter connection s is generally:

[0012] I=I_sAeV/V_T

[0013] Therefore,

[0014] I₁=I_sA_1e^V/V

[0015] I₂=I_sA₂eV^BE2

[0016] The bandgap voltage V_Bgcan be made substantially temperature invariant by selecting the values of resistors 19 and 18, R₁and R₀, so that the bandgap voltage follows the formula, $\begin{array}{cc}\begin{array}{c}{V}_{\mathrm{Bg}}=V_{\mathrm{BE1}}+2I·R_0\\ =V_{\mathrm{BE}}+2·\frac{\Delta V_{\mathrm{BE}}}{\mathrm{R1}}·\mathrm{R0}\end{array}& \left(2\right)\end{array}$

[0017] where I is the total current through both branches (I₁+I₂) of the bandgap voltage generator. Since the temperature coefficient for silicon has a known negative temperature coefficient of minus 2 MV/° C., the negative temperature coefficient is effectively compensated for by the term 2IR₀, recognizing that the current I through one branch of the bandgap generator is: $\begin{array}{cc}I=\frac{\Delta V_{\mathrm{BE}}}{R_1}& \left(3\right)\end{array}$

[0018] Accordingly, equation (2) becomes $\begin{array}{cc}{V}_{\mathrm{Bg}}=V_{\mathrm{BE}}+2\frac{R_0}{R_1}·\Delta V_{\mathrm{BE}}& \left(4\right)\end{array}$

[0019] ΔV_BE, is the difference between base emitter voltages of transistors 15 and 16, or $\begin{array}{cc}\begin{array}{c}\Delta V_{\mathrm{BE}}=V_{\mathrm{BE1}}-V_{\mathrm{BE2}}\\ =V_T\mathrm{In}\frac{A_2}{A_1}\end{array}& \left(5\right)\end{array}$

[0020] Since ΔV_BEequals $V_T\ln \frac{A_2}{A_1},$

[0021] the bandgap voltage V_BGcan be represented by $\begin{array}{cc}{V}_{\mathrm{BG}}=V_{\mathrm{BE}}+2\frac{R_0}{R_1}·\mathrm{In}\frac{A_2}{A_1}·\frac{\mathrm{KT}}{q}& \left(6\right)\end{array}$

[0022] Since V_BEwill have a negative coefficient, the remaining terms of equation 6 can be adjusted by selecting the ratio of R₀/R₁to provide a positive temperature coefficient to offset the negative coefficient of the base emitter voltage of NPN bipolar transistors 15 and 16.

[0023] The substantially temperature invariant bandgap voltage developed at the base of bipolar transistors 15 and 16 is coupled through bipolar transistor 14 to the input of a current source comprising bipolar transistor 21 and resistor 22. The value of resistor 22 establishes for a given bandgap voltage applied to the base of transistor 21 a bias current 13 for the RF circuits of the cellular telephone.

[0024] Bipolar transistor 14 is connected in a diode configuration (base to collector) in one of the current paths of the bandgap voltage generator. As the transistors 14 and 21 have substantially the same base emitter junction area A₁, A₂and are of the same material, the voltage drops across the base emitter connections of transistors 14 and 21 essentially offset each other so that the voltage applied to resistor 22, shown as V_out, is essentially the bandgap voltage.

[0025] Control over the temperature coefficient of current I₃can therefore be affected by selecting the values R₁, R₀of resistors 19 and 18 so that they either provide for total compensation of the negative temperature coefficient of the bandgap generator, or to provide a slightly positive temperature coefficient which may be helpful for offsetting the effects of temperature on other circuits which operate from bias current I₃.

[0026] As is common in bandgap voltage generators, a start up circuit is provided to make certain the circuit wakes up when power is supplied and assumes a stable bandgap voltage producing state. It is possible that the current mirror comprising PFET 12 and PFET 13 may start in a zero current conduction mode. In order to force the bandgap voltage generator into operation in a stable state, a start up circuit is provided which injects current into the branch of the bandgap generator comprising PFET 12 and bipolar transistor 15.

[0027] If the bandgap voltage circuit has not reached a stable state, a PFET 30 will inject current into the branch comprising PFET 12 and bipolar transistor 15. In effect, transistor 29 operates as a comparator to determine whether or not the voltage level at the gate of PFETS 12 and 13 is sufficient to render PFET 29 non-conducting. PFET 29 is included in a current mirror comprising NFET 27 and NFET 28. The current mirror circuit of NFET 27, 28 is kept in a conduction mode by PFET 26. In operation, if the current mirror comprising PFET 12, 13 is producing current for maintaining the bandgap voltage, current is diverted by PFET 29 so that PFET 30 no longer injects current into the branch of the bandgap circuit comprising PFET 12 and bipolar transistor 15.

[0028] The foregoing description of the invention illustrates and describes the present invention. Additionally, the disclosure shows and describes only the preferred embodiments of the invention but, as mentioned above, it is to be understood that the invention is capable of use in various other combinations, modifications, and environments and is capable of changes or modifications within the scope of the inventive concept as expressed herein, commensurate with the above teachings and/or the skill or knowledge of the relevant art. The embodiments described hereinabove are further intended to explain best modes known of practicing the invention and to enable others skilled in the art to utilize the invention in such, or other, embodiments and with the various modifications required by the particular applications or uses of the invention. Accordingly, the description is not intended to limit the invention to the form or application disclosed herein. Also, it is intended that the appended claims be construed to include alternative embodiments.