Inverter Circuit Having Transistors Operable in a Shallow Saturation Region for Avoiding Fluctuation of Electrical Characteristics

1. Field of the Invention

The present invention relates to an inverter circuit which is, for example, used as an output stage of a word driver in metal-oxide semiconductor (MOS) memory devices.

2. Description of the Prior Art

In recent years, in MOS (broadly, metal-insulated semiconductor (MIS)) memory devices, the potential of a power supply has been gradually decreased. Therefore, if a signal having a particularly high potential is required, such a high potential signal is generated by using a charge-pumping circuit or the like. For example, such a high-potential signal is supplied as a clock signal to an inverter circuit which serves as an output stage of a word driver in MOS memory devices.

One prior art inverter circuit comprises two MOS transistors: a load transistor on the side of a high-potential power supply and a driving transistor on the side of a low-potential power supply. In this case, the two transistors are connected in series between the high-and low-potential power supplies. Two clock signals, which are approximately opposite in phase are supplied to the gates of the two transistors, respectively, while the connection node between the two transistors serves as an output terminal of the inverter circuit.

In the above-mentioned prior art inverter circuit, however, when the load transistor and the driving transistor are in an off state and an on state, respectively, the potential at the connection node is low. Therefore, a high potential is applied directly to the load transistor, which, in turn, operates in a deep saturation region. As a result, a large amount of hot electrons and hot holes having high energy are generated and captured by a gate oxide layer so that the electrical characteristics, such as the threshold voltage (V.sub.th) value, of the load transistor fluctuate. In addition, since the load transistor operates in a state near to a punch-through state during the conduction mode in the deep saturation region, the load transistor may break down.

Similarly, when the load transistor and the driving transistor are in an on state and an off state, respectively, the potential at the connection node is high. Therefore, a high potential is also applied directly to the driving transistor, which, in turn, operates in a deep saturation region. Therefore, the above-mentioned phenomenon also occurs in the driving transistor.

It is, therefore, an object of the present invention to provide an inverter circuit comprising a load transistor and a driving transistor connected in series, in which both of the transistors can be operated in a shallow saturation region, thereby avoiding fluctuation of the electrical characteristics, such as threshold voltage (V.sub.th) values, of these transistors and the breakdown of these transistors.

According to the present invention, there is provided an inverter circuit comprising: a low-potential power supply; a first high-potential power supply; a second high-potential power supply, the potential of the second power supply being higher than that of the first high-potential power supply; a clock signal generating means for generating first and second clock signals approximately opposite in phase; and a load MIS transistor and a driving MIS transistor connected in series between the low-potential power supply and the second high-potential power supply, the load MIS transistor and the driving MIS transistor each having a gate for receiving one of the clock signals. The present invention also comprises a connection node, between the load MIS transistor and the driving MIS transistor, generating an output signal; at least one first MIS transistor connected between the first high-potential power supply and the load MIS transistor; and a means for raising a gate potential of the first MIS transistor to a potential level higher than that of the second high-potential power supply.

The present invention will be more clearly understood from the description as set forth below contrasting the present invention with the prior art and with reference to the accompanying drawings.

FIG. 1 is a circuit diagram of a prior art inverter circuit;

FIG. 2 is a timing diagram of the signals appearing in the circuit of FIG. 1;

FIG. 3 is a circuit diagram of an embodiment of the inverter circuit according to the present invention;

FIG. 4 is a timing diagram of the signals appearing in the circuit of FIG. 3;

FIG. 5 is a circuit diagram of another embodiment of the inverter according to the present invention;

FIG. 6 is a circuit diagram in which the circuit of FIG. 3 is applied to a word driver circuit for generating a word line driving signal; and

FIG. 7 is a timing diagram of the signals appearing in the circuit of FIG. 6.

First, the prior art inverter circuit will be explained with reference to FIGS. 1 and 2. In FIG. 1, V.sub.SS, V.sub.CC, and V.sub.PP are power supplies. In this case, V.sub.SS, V.sub.CC, and V.sub.PP indicate also the potential values of the power supplies V.sub.SS, V.sub.CC, and V.sub.PP, respectively. In more detail, the potential V.sub.SS is low while the potential V.sub.CC is high. Further, the potential V.sub.PP is higher than the potential V.sub.CC. For example, the potential V.sub.PP is generated by a charge-pumping circuit or the like.

The inverter circuit of FIG. 1 comprises a load MIS transistor Q.sub.1 and a driving transistor Q.sub.2 connected in series between the power supply V.sub.SS and the power supply V.sub.PP. A clock signal .phi..sub.0 and its inverted signal .phi..sub.0 are supplied to the gates of the transistors Q.sub.1 and Q.sub.2, respectively, and an output signal .phi..sub.1 is obtained at the connection node of the two transistors. In this case, the high-level potential of the signal .phi..sub.0 is set at a higher value than the power supply potential V.sub.PP by using the bootstrap effect or the like. Therefore, the high-level potential of the output signal .phi..sub.1 becomes the same as the power supply potential V.sub.PP.

The operation of the circuit of FIG. 1 will be explained with reference to FIG. 2. Note that V.sub.th indicates a common theshold voltage of the transistors. When the potentials of the clock signals .phi..sub.0 and .phi..sub.0 are low (V.sub.SS) and high (V.sub.CC -V.sub.th), respectively, the transistors Q.sub.1 and Q.sub.2 are in an off state and an on state, respectively, so that the potential of the output signal 100.sub.1 is low (V.sub.SS). Next, when the potential of the signal .phi..sub.0 is changed from low (V.sub.SS) to high (>V.sub.PP) and, simultaneously, the potential of the signal .phi..sub.0 is changed from high (V.sub.CC -V.sub.th) to low (V.sub.SS), the transistors Q.sub.1 and Q.sub.2 change to an on state and an off state, respectively. As a result, the potential of the output signal .phi..sub.1 increases. In this case, since the potential of the signal .phi..sub.0 is set at a higher value than V.sub.PP by using the bootstrap effect or the like, the potential of the output signal .phi..sub.1 is pulled up to the high power supply potential V.sub.PP, as illustrated in FIG. 2.

In FIG. 1, however, when the potential of each of the signals .phi..sub.0 and .phi..sub.0 remains low (V.sub.SS) and high (V.sub.CC -V.sub.th), respectively, the large difference V.sub.PP in potential is applied between the drain and source of the load transistor Q.sub.1, which causes the transistor Q.sub.1 to be operated in a deep saturation region. Therefore, as explained above, the electrical characteristics, such as the threshold voltage (V.sub.th) value, of the load transistor Q.sub.1 may fluctuate, and, in addition, breakdown of the load transistor Q.sub.1 may occur. Similarly, when the potential of each of the signals .phi..sub.0 and .phi..sub.0 remains high (>V.sub.PP) and low (V.sub.SS), respectively, the large difference V.sub.PP in potential is applied to the drain and source of the driving transistor Q.sub.2. Therefore, the above-mentioned phenomenon also occurs in the driving transistor Q.sub.2.

FIG. 3 is a circuit diagram illustrating an embodiment of the inverter circuit according to the present invention. In FIG. 3, the transistors Q.sub.3, Q.sub.4, and Q.sub.5 and a capacitor C.sub.1 are added to the elements of FIG. 1. Note that the transistor Q.sub.4 and the capacitor C.sub.1 form a bootstrap circuit.

By the addition of the transistor Q.sub.3, when the signals .phi..sub.0 and 100.sub.0 are low (V.sub.SS) and high (V.sub.CC -V.sub.th), respectively, the large difference V.sub.PP in potential is divided by the two transistors Q.sub.1 and Q.sub.3, thereby reducing the load on the load transistor Q.sub.1. By the addition of the transistor Q.sub.5, when the signals .phi..sub.0 and .phi..sub.0 are high (>V.sub.PP) and low (V.sub.SS), respectively, the large difference V.sub.PP in potential is divided by the two transistors Q.sub.2 and Q.sub.5, thereby reducing the load on the driving transistor Q.sub.2.

The operation of the circuit of FIG. 3 will be explained with reference to FIG. 4. When the potentials of the signals .phi..sub.0 and .phi..sub.0 are low (V.sub.SS) and high (V.sub.CC -V.sub.th), respectively, the transistors Q.sub.1 and Q.sub.2 are in an off state and an on state, respectively, so that the potential at the node N.sub.1, that is, the potential of the output signal .phi..sub.1, becomes low (V.sub.SS). Simultaneously, current flows from the power supply V.sub.CC through the transistor Q.sub.4 to the node N.sub.2, so that the capacitor C.sub.1 is charged. In this case, the potential at the node N.sub.2 equals V.sub.CC -V.sub.th. Note that, in this state, the large difference V.sub.PP in potential is divided by the two transistors Q.sub.1 and Q.sub.3, and, accordingly, the drain-source voltage of the load transistor Q.sub.1 is decreased.

Next, when the potential of the signal .phi..sub.0 is changed from low (V.sub.SS) to high (>V.sub.PP) and, in addition, the signal .phi..sub.0 is changed from high (V.sub.CC -V.sub.th) to low, (V.sub.SS) the transistors Q.sub.1 and Q.sub.2 change to an on state and an off state, respectively, and, simultaneously, the transistor Q.sub.3 changes to an on state due to the high potential at the node N.sub.2. Therefore, current flows from the power supply V.sub.PP through the transistors Q.sub.3 and Q.sub.1 to the node N.sub.1, which, in turn, increases the potential of the node N.sub.1. At this time, since the transistor Q.sub.4 is cut off due to the fact that the gate-source voltage of the transistor Q.sub.4 is less than the threshold voltage (V.sub.th) value thereof, the potential at the node N.sub.2 increases as the potential at the node N.sub.1 increases. Therefore, the potential at the node N.sub.2 becomes higher than the potential V.sub.CC and, finally, this potential becomes substantially higher than the potential V.sub.PP. Such pushing up of the potential at the node N.sub.2 is called the bootstrap effect. Thus, since a substantially higher potential than the potential V.sub.PP is applied to the gates Q.sub.3 and Q.sub.1, the potential at the node N.sub.1, that is, the potential of the output signal .phi..sub.0, equals the potential V.sub.PP. Note that, even in this state, the large difference V.sub.PP in potential is divided by the two transistors Q.sub.2 and Q.sub.5, and, accordingly, the drain-source voltage of the driving transistor Q.sub.2 is decreased.

That is, the output waveform (FIG. 4) of the inverter circuit of FIG. 3 is about the same as the output waveform (FIG. 2) of the circuit of FIG. 1.

FIG. 5 is a circuit diagram illustrating another embodiment of the inverter circuit according to the present invention. In FIG. 5, the capacitor C.sub.1 is connected to the gate of the load transistor Q.sub.1, not to the node N.sub.1. Therefore, in this case, the potential at the node N.sub.2 is pushed up by the increase of the potential of the signal .phi..sub.0 to cause the bootstrap effect. FIG. 6 is a circuit diagram in which the circuit of FIG. 3 is applied to a word driver circuit for generating a word line driving signal. In FIG. 6, C.sub.1 indicates the circuit of FIG. 3, and C.sub.2 is a circuit for generating the clock signals .phi..sub.0 and .phi..sub.0.

In the circuit C.sub.2, the transistors Q.sub.11 and Q.sub.12 comprise an inverter which is operated by receiving clock signals .phi..sub.11 and .phi..sub.11 which are opposite in phase. The transistors Q.sub.13 and Q.sub.14 also comprise an inverter circuit which is operated by receiving the clock signal .phi..sub.11 and the potential at the node N.sub.11 of the pre-stage inverter circuit. A transistor Q.sub.15 is interposed between the second stage inverter circuit and a bootstrap circuit. This bootstrap circuit is formed by the transistors Q.sub.16 through Q.sub.20 and a capacitor C.sub.2. Among them, the transistor Q.sub.17 is used to protect the transistor Q.sub.18 when the bootstrap potential node N.sub.14 is discharged. That is, when the bootstrap potential at the node N.sub.14 is paticularly high, such high potential is divided by the two transistors Q.sub.17 and Q.sub.18, thereby reducing the drain-source voltage of the transistor Q.sub.18.

The operation of the circuit of FIG. 6 will now be explained with reference to FIG. 7. First, when the potentials of the clock signals .phi..sub.11 and .phi..sub.11 are low (V.sub.SS) and high (V.sub.CC), respectively, the transistors Q.sub.12 and Q.sub.13 are in an on state so that the potentials at the nodes N.sub.11 and N.sub.12 are low (V.sub.SS) and high (V.sub.CC -V.sub.th), respectively. In addition, since the transistor Q.sub.15 is in an on state, the node N.sub.13 is charged up to the potential V.sub.CC -V.sub.th. Next, when the potentials of the signals .phi..sub.11 and .phi..sub.11 becomes high (V.sub.CC) and low (V.sub.SS), respectively, the transistors Q.sub.11 and Q.sub.12 turn on and off, respectively. Therefore, the potential at the node N.sub.11 becomes high, and, accordingly, the transistor Q.sub.14 turns on so that the potential at the node N.sub.12 becomes low since the transistor Q.sub.13 has already turned off. Thus, after the potential of the signal .phi..sub.11 becomes high (V.sub.CC), the potential at the node N.sub.12 is changed from high (V.sub.CC -V.sub.th) to low (V.sub.SS) with some delay. In addition, the node N.sub.14 is charged by the signal .phi..sub.11. In this case, the discharging of the node N.sub.12 is carried out later than the charging of the node N.sub.14. Therefore, the node N.sub.13 is discharged by the transistor Q.sub.15, which cuts off the transistor Q.sub.16. Thus, at this time, the node N.sub.14 changes to a floating state. Simultaneously, the transistor Q.sub.20 is cut off so that the potential at the node N.sub.16 changes from low (V.sub.SS) to high (V.sub.CC). As a result, the potential at the node N.sub.14 is pushed up to a higher level than the potential V.sub.CC by the capacitive coupling of the capacitor C.sub.2. Thus, a particularly higher potential than the potential V.sub.CC is obtained at the node N.sub.14. Note that such particularly high potential at the node N.sub.14 is divided by the transistors Q.sub.17 and Q.sub.18, thereby reducing the load of the transistor Q.sub.18. Next, when the signals .phi..sub.11 and .phi..sub.11 are reversed, the transistor Q.sub.18 turns on, and, as a result, the charges stored at the node N.sub.14 are discharged through the on-state transistors Q.sub.17 and Q.sub.18.

In the above-mentioned embodiments, only the transistor Q.sub.3 (or Q.sub.5) is connected in series to the transistor Q.sub.1 (or Q.sub.2); however, two or more transistors can be connected in series to the transistor Q.sub.1 (or Q.sub.2).

As explained hereinbefore, the inverter circuit according to the present invention is advantageous in that the load transistor and the driving transistor can be operated in a shallow saturation region, thereby avoiding fluctuation of the electrical characteristics, such as the threshold voltage value, and the breakdown of the transistors.