Switching power converter controller with direct current transformer sensing

[0001] 1. Field of the Invention

[0002] The present invention relates in general to the field of electronics, and more specifically to a switching power converter control system and method with direct current transformer sensing.

[0003] 2. Description of the Related Art

[0004] Switching power converters provide regulated output voltages for numerous electronic products including lamps, such as light emitting diode and gas discharge type lamps, cellular telephones, computing devices, personal digital assistants. As the name suggests, switching power converters include a switch that controls the output voltage. Sensing current in the switch provides feedback to a controller to allow the controller to control the switch and, thus, regulate the output voltage of the switching power converter. The switch current can be sensed in any number of ways. Switch current can be sensed as a voltage across a sense resistor in series with the switch. However, conducting a high switch current across the sense resistor of the switching power converter can result in consequential power losses. Current transformers provide a more efficient way of sensing the switch current. A primary-side of the current transformer is placed in series with the switch to conduct the switch current. The secondary-side conducts a stepped-down switch current. A sense resistor across the secondary-side of the current transformer develops a voltage corresponding to the stepped-down switch current, thus, lowering the power loss across the sense resistor.

[0005] FIG. 1 depicts a power control system 100, which includes switching power converter 102. In at least one embodiment, switching power converter 102 operates in a continuous conduction mode (CCM). In CCM, the inductor current i_L in inductor 104 does not go to zero during operation of switching power converter 102. As subsequently described in more detail, in CCM operation, PFC and output voltage controller 106 senses switch current i_DP in switch 108 by sensing a voltage V_CTS corresponding to a current i_DS across a secondary winding 110 of current transformer 112. Switch 108 is implemented as a field effect transistor (FET).

[0006] Switching power converter 102 converts voltage sourced from voltage source 114 into a regulated link voltage V_LINK across link capacitor 116. Voltage source 114 supplies an alternating current (AC) input voltage V_IN to a full bridge diode rectifier 117. The voltage source 114 is, for example, a public utility, and the AC voltage V_IN is, for example, a 60 Hz/110 V line voltage in the United States of America or a 50 Hz/220 V line voltage in Europe. The rectifier 117 rectifies the input voltage V_IN and supplies a rectified, time-varying, line input voltage V_X to the switching power converter 102. Capacitor 115 filters high reference components of input voltage V_X and, thus, provides electromagnetic interference (EMI) protection.

[0007] The switching power converter 102 includes at least two switching operations. The first operation controls switch 108 to provide power factor correction (PFC), and the second operation controls switch 108 to provide regulation of link voltage V_LINK. The goal of power factor correction technology is to make the switching power converter 102 appear resistive to the voltage source 114 so that the real power provided to switching power converter 102 is equal to the apparent power provided to switching power converter 102.

[0008] During operation of switching power converter 102, the inductor current i_L ramps ‘up’ when switch 108 conducts, i.e. is “ON”, and ramps down when switch 108 is nonconductive, i.e. is “OFF”. When switch 108 is ON, inductor current i_L magnetizes inductor 104 and also magnetizes the primary-side winding 118 of current transformer 112. In at least one embodiment, current transformer 112 is configured as a toroid with a primary-side to secondary-side winding ratio of 1:N. “N” is, for example, 10. The primary-side transformer current i_DP equals the inductor current i_L when switch 108 is ON. Thus, the second-side current i_DS equals i_DP/N when switch 108 is ON.

[0009] When switch 108 is ON, secondary-side current i_DS flows through diode 120 and is filtered by filter 122 to reduce noise. Resistor 121 provides a number of functions, such as reverse current protection for diode 120. The resistance value of resistor 121 is a matter of design choice. In at least one embodiment, resistor 121 has a resistance greater than an input resistance of load 132. In at least one embodiment, the resistance of resistor 121 is approximately equal to at least ten times the input resistance of load 132. A secondary-side current transformer voltage V_CTS develops across filter 122 when switch 108 is ON. In embodiment, filter 122 is designed so that the transformer voltage V_CTS is 1V at the peak of secondary-side current i_DS. Current synthesizer 126 charges capacitor 128 with a current proportional to the switch current i_DS. The transformer voltage V_CTS is level shifted up one ‘diode drop’ voltage across diode 124 relative to the voltage V_CTC across capacitor 128. (A “diode drop” voltage refers to a voltage drop across a forward biased diode. A typical diode drop voltage is 0.7V). The inductor current i_L is the same as the switch current i_DS when switch 108 is ON. Thus, control signal generator 130 determines the inductor current i_L when switch 108 is ON based on the secondary-side voltage V_CTS.

[0010] When switch 108 is OFF, secondary-side current i_DS is zero, and inductor current i_L flows through diode 127 to charge link capacitor 116. Diode 127 prevents reverse current flow into inductor 104. PFC and output voltage controller 106 includes a current synthesizer 128 to reconstruct the inductor current i_L during the OFF time of switch 108. In general, current synthesizer 126 controls the discharge of capacitor 128 in proportion to the input voltage V_X and the link voltage V_LINK to reconstruct the down slope of inductor current i_L and, thus, determine the inductor current i_L when switch 108 is OFF. Upon determining the switch current i_DS while switch 108 is ON using the secondary-side voltage V_CTS of current transformer 110 and reconstructing the inductor current i_L when switch 108 is OFF, control signal generator 130 generates control signal CS₀ to regulate the link voltage V_LINK and provide power factor correction for switching power converter 102. The link voltage V_LINK is supplied to load 132. A more complete explanation of an exemplary operation of PFC and output voltage controller 106 is described in “High Performance Power Factor Preregulator”, Unitrode Products from Texas Instruments, part numbers UC2855A/B and UC3855A/B, Data Sheet June 1998, revised October 2005 and Application Report SLUA146 May 1996, revised April 2004.

[0011] The cost of external components is generally higher than the cost of integrated components. For example, if filter 122 could be eliminated, the cost of power control system 100 could potentially be reduced.

[0012] In one embodiment of the present invention, an apparatus includes an integrated circuit. The integrated circuit includes a controller configured to:

• • [0013] receive a current from a secondary-side of a current transformer, wherein the secondary-side current represents an inductor current in a switching power converter; and
  • [0014] utilize the received secondary-side current to control power factor correction of the switching power converter.

[0015] In another embodiment of present invention, a method includes receiving a current from a secondary-side of a current transformer at a node of a switching power converter controller. The secondary-side current represents an inductor current in a switching power converter. The method includes utilizing the received secondary-side current to control power factor correction of the switching power converter.

[0016] In a further embodiment of the present invention, an apparatus includes an integrated circuit. The integrated circuit includes a controller configured to:

• • [0017] receive a current from a secondary-side of a current transformer, wherein the secondary-side current represents an inductor current in switching power converter;
  • [0018] compare the current received from the current transformer with a reference current; and
  • [0019] generate a switch control signal to control conductivity of a switch in the switching power converter to control power factor correction and regulate the link voltage;
  • [0020] determine an elapsed time between a beginning of a conductive time of the switch and when the current received from the current transformer exceeds the reference current; and
  • [0021] utilize the elapsed time to control conductivity of the switch.

[0029] A power control system includes a current transformer to step down a switch current of a switching power converter. In at least one embodiment, the stepped down current is received by a switching power converter controller. Since the current is received by the controller, the current is not converted into a voltage prior to receipt by the controller in order for the controller to monitor an inductor current of the switching power converter. In at least one embodiment, The controller utilizes current sensing of the stepped down switch current to control at power factor correction of a switching power converter. In at least one embodiment, the controller includes a current comparator that compares the stepped down switch current with a reference current. The reference current can be fixed or programmable. In at least one embodiment, current sensing provides very high accuracy using a relatively inexpensive comparator. In at least one embodiment, the controller includes a voltage converter to convert the switch current into a voltage within the controller. The controller includes a comparator to compare the voltage representing the switch current with a reference voltage. In at least one embodiment, the controller uses the current or voltage comparisons to control power factor correction and output voltage regulation of a switching power converter. The reference voltage can be fixed or programmable.

[0030] FIG. 2 depicts power control system 200 having a controller 202 to control switching power converter 204. As subsequently described in more detail, controller 202 senses secondary-side current i_CTS, which is derived from switch current i_CTP of switching power converter 204. In at least one embodiment, the secondary-side current i_CTS is received by the controller 202 from the switching power converter 204. Thus, the secondary-side current i_CTS is not converted into a voltage prior to receipt of the secondary-side current i_CTS. Controller 202 uses the sensed current i_CTS to determine switch control signal CS₁, and switch control signal CS₁ controls the conductivity of switch 206. By directly sensing the switch related current i_CTS, in at least one embodiment, power control system 200 reduces costs associated with external components, such as components of filter 122 (FIG. 1).

[0031] Power control system 200 includes a current transformer 208 that steps down the primary-side switch current i_CTP. The switch current i_CTP magnetizes the primary-side 210 of current transformer 208 and induces current i_CTS in the secondary-side 212 of current transformer 208. Current transformer 208 can be any type of transformer and is, in at least one embodiment, a toroid configured transformer. The ratio of the primary-side, switch current i_CTP to secondary-side current i_CTS is 1:N, where N is the ratio of turns on the secondary-side 212 to the primary-side 210 of current transformer 208. N can be any number and is, for example, 10. Thus, for a switch current i_CTP of 10 A, the secondary-side current i_CTS is 1 A, i.e 10 Amps/N=10/10 Amps=1 A. Diode 214 prevents reverse current flow through the secondary-side 212. Resistor 215 provides a number of functions, such as reverse current protection for diode 214. The resistance value of resistor 215 is a matter of design choice. In at least one embodiment, resistor 215 has a resistance greater than an input resistance of load 224. In at least one embodiment, the resistance of resistor 215 is approximately equal to at least ten times the input resistance of load 224. In at least one embodiment, the switch current i_CTP and, thus, the secondary-side current flow only when switch 206 is ON. The inductor current i_L equals the switch current i_CTP when switch 206 is ON. Thus, the inductor current i_L equals the secondary-side current i_CTS divided by N when switch 206 is ON, i.e. i_L=i_CTS/N.

[0032] As previously discussed in the context of PFC and voltage controller 106, controller 202 uses information about the inductor current i_L in switching power converter 204 to control a duty cycle of control signal CS₁. The switch current i_CTP on the primary-side of current transformer 208 is the inductor current i_L when switch 206 is ON. The secondary-side current i_CTS is a stepped down version of the switch current i_CTP, and, thus, secondary-side current i_CTS represents the inductor current i_L when switch 206 is ON. Since the relationship between the inductor current i_L and switch current i_CTS is known when switch 206 is ON, i.e. i_CTS=i_L/N, sensing secondary-side current i_CTS when switch 206 is ON provides controller 202 with the information about inductor current i_L to generate control signal CS₁.

[0033] Controller 202 directly senses secondary-side current i_CTS at an input of current comparator 216. The process by which controller 202 senses the secondary-side current i_CTS is a matter of design choice. In at least one embodiment, controller 202 performs a direct current comparison between secondary-side current i_CTS and a reference current. In at least one embodiment, controller 202 converts the secondary-side current i_CTS into a voltage and compares the voltage with a reference voltage. For example, in at least one embodiment, reference REF is a reference current, and signal CTS is the secondary-side current i_CTS. Comparator 202 compares signal CTS directly with reference REF. In at least one embodiment, reference REF is a voltage reference source, and signal CTS is a voltage representing secondary-side current i_CTS. In this embodiment, controller 202 includes optional voltage converter 219 to receive secondary-side current i_CTS and convert secondary-side current i_CTS into signal CTS. Comparator 216 then compares the signal CTS, which represents secondary-side current i_CTS as a voltage, with the voltage of reference REF. The output signal COMP₀ represents the result of the comparison by comparator 216. In at least one embodiment, the reference 218 is set to a desired average value of signal CTS. Control signal generator 220 monitors the output signal COMP₀ to determine the time T1 that elapses from when the control signal generator 220 turns switch 2060N until the output signal COMP₀ indicates that the secondary-side current i_CTS equals the reference current. For example, in at least one embodiment, control signal generator 220 initiates timer 222 when control signal generator 220 generates a pulse of control signal CS₁ to turn switch 2060N.

[0034] In at least one embodiment, when the signal CTS is less than the reference REF, the output signal COMP₀ of comparator 216 is a logical 0. When the signal CTS increases to a value greater than the reference REF, the output signal COMP₀ switches to a logical 1. When the output signal COMP₀ switches to logical 1, control signal generator 220 observes the elapsed time T1 determined by timer 222. Control signal generator 220 utilizes the elapsed time T1 to generate control signal CS₁. Thus, in at least one embodiment, determination of the control signal CS₁ is derived from the value of secondary-side current i_CTS, and the secondary-side current i_CTS is derived from the inductor current i_L. Commonly assigned U.S. patent application Ser. No. 11/864,366 filed on Sep. 28, 2007, entitled “Time-Based Control of a System Having Integration Response”, and inventor John. L. Melanson describes exemplary systems and methods for generating control signal CS₁ using the elapsed time T1. U.S. patent application Ser. No. 11/864,366, referred to herein as Melanson I, is hereby incorporated by reference in its entirety. The particular implementation of controller 202 is a matter of design choice. For example, controller 202 can be implemented using analog and/or digital circuits or using analog or digital circuits in combination with a processor executing software instructions stored in a memory.

[0035] In at least one embodiment, controller 202 generates control signal CS₁ and controls switch 206 to operate switching power converter 202 in continuous conduction mode. In at least one embodiment, controller 202 also controls the conductivity of switch 206 to regulate link voltage V_LINK and control power factor correction of switching power converter 204. Switching power converter 204 can be any type of switching power converter such as boost-type converter (such as switching power converter 102 of FIG. 1), a buck-type converter, a buck-boost converter, or a Cúk converter. Switching power converter 204 supplies the link voltage V_LINK to load 224. Load 224 can be any type of load. For example, in at least one embodiment, load 224 is a power supply, such as a DC-DC or DC-AC converter. “DC” stands for direct current. “AC” stands for alternating current. Load 224 can also include, for example, a lighting system that includes one or more light emitting diodes or gas discharge lamps. In at least one embodiment, switch 206 is a field effect transistor (FET), and control signal CS₁ is a duty cycle modulated signal, such as signal 207. Voltage source 226 can be any type of voltage source, such as AC voltage source 101. In at least one embodiment, voltage source 226 includes a rectifier (not shown) and supply voltage V_SUPPLY is a rectified AC voltage.

[0036] FIG. 3 depicts power control system 300, which represents one embodiment of power control system 200 (FIG. 2). Power control system 300 includes controller 302, and controller 302 includes current comparator 304, which respectively represent embodiments of controller 202 and current comparator 216 (FIG. 2). Controller 302 is an integrated circuit that generates a control signal CS₂ to control power factor correction and regulate link voltage V_LINK of switching power converter 308. In at least one embodiment, control signal CS₂ is a duty cycle modulated signal, such as signal 303. As previously discussed in the context of PFC and voltage controller 106 and controller 202, controller 302 uses information about the inductor current i_L in switching power converter 308 to control a duty cycle of control signal CS₂. The switch current i_CTP in the primary-side winding 309 of current transformer 310 is the inductor current i_L when switch 312 is ON.

[0037] The secondary-side current i_CTS in the secondary-side winding 311 is a stepped down version of the switch current i_CTP and, thus, secondary-side current i_CTS represents a stepped down version of the inductor current i_L when switch 312 is ON. Diode 313 prevents reverse current flow through the secondary-side winding 311. Resistor 307 provides a number of functions, such as reverse current protection for diode 313. The resistance value of resistor 307 is a matter of design choice. In at least one embodiment, resistor 307 has a resistance greater than an input resistance of load 352. In at least one embodiment, the resistance of resistor 307 is approximately equal to at least ten times the input resistance of load 352. The secondary-side current i_CTS is received by controller 302 at pin 306. The secondary-side current i_CTS is related to the switch current i_CTP by the winding ratio N between the primary-side winding 309 and secondary-side winding 311 of current transformer 310. Since the switch current i_CTP equals the inductor current i_L when switch 312 is ON, the secondary-side current i_CTS is related to the inductor current i_L when switch 312 is ON by Equation [1]:
i_CTS=i_L/N  [1].

[0038] Current comparator 304 allows controller 302 to generate control signal CS₂ without converting the switch current i_CTP or the secondary-side current i_CTS into a voltage. Thus, power control system 300 avoids power losses associated with otherwise converting the switch current i_CTP or the secondary-side current i_CTS into a voltage.

[0039] The current comparator 304 receives secondary-side current i_CTS at pin 306. FETs 314 and 316 are configured as a current mirror. Diode 315 holds node 322 near the source voltage of FET 314 if the source voltage of FET 314 rises above the voltage at node 322 by more than the diode drop voltage of diode 315. The use of FETs 314 and 316 is a matter of design choice. Other components, such as bipolar junction transistors can be substituted for FETs 314 and 316 to implement the current mirror. Constant current supply 318 provides a constant drain current i_M to FET 316. Constant current supply 318 can be, for example, a fixed or programmable current supply. An exemplary programmable current supply is discussed in more detail with reference to FIG. 5. The current mirror provides a constant mirrored drain current i_M/R of FET 314 as a reference current for comparison with secondary-side current i_CTS. “R” is the current mirror ratio and is established by a ratio of the relative size of FET 316. A control node 317 of switch 312 is connected to pin 319 to receive control signal CS₂. As subsequently discussed in more detail, in at least one embodiment, the value R is set so that the timer 328 of control signal generator 320 can determine an elapsed time from a beginning of a pulse of control signal CS₂ to turn switch 3120N and a desired average value of inductor current i_L.

[0040] In accordance with Kirchoff's current law, at node 322:
i_S=i_CTS−i_M/R  [2];
where “i_S” is the current at node 322 and a result of the comparison between secondary-side current i_CTS and reference current i_M/R, i_Crs is the secondary-side current i_CTS into node 322, and i_M/R is the drain current reference of FET 314 out of node 322. Thus, in accordance with Equation [2], current comparator 304 compares the secondary-side current i_CTS with the reference current i_M/R, and current i_S represents a result of the comparison. In this embodiment, “R” is set so that i_M/R equals i_L—TARGET/N, where N is the winding ratio of current transformer 310. Thus, in at least one embodiment, R is determined in accordance with Equation [3]:
R=N·i_M/i_L—TARGET  [3],
where R is the ratio of mirror current i_M to the drain current of FET 314 and i_L—TARGET is a target average value of inductor current i_L. For example, if the target inductor current i_L—TARGET is 1 A, the winding ratio N of current transformer 310 is 10, and current i_M is 1 A, then R equals 10.

[0041] Comparison current i_S charges node 326 of comparator 324 to a voltage V_S0. Comparator 324 compares the voltage V_S0 to a reference voltage V_REF0 and generates comparison signal COMP₁. In at least one embodiment, the value of a reference voltage V_REF0 input to comparator 324 is set so that so that: (i) voltage V_S0 at comparator 324 exceeds reference voltage V_REF0 when secondary-side current i_CTS is greater than reference current i_M/R and (ii) voltage V_S0 is less than voltage V_REF0 when secondary-side current i_CTS is less than the reference current i_M/R. Equations [4] and [5] represent one embodiment of the relationships between secondary-side current i_CTS and voltage V_S0:
if i_CTS>i_M/R, then V_S0>V_REF0  [4], and
if i_CTS<i_M/R, then V_S0<V_REF0  [5].
The value of voltage V_REF0 is set so that Equations [4] and [5] are true. In at least one embodiment, comparator 324 includes a high input impedance and is, thus, very responsive to even small changes in current i_S. However, in at least one embodiment, comparator 324 is relatively insensitive to voltage offsets. Thus, in at least one embodiment, comparator 324 can be implemented as a relatively simple comparator, such as a flip-flop. Therefore, implementing comparator 324 can decrease manufacturing costs of controller 302.

[0042] Control signal generator 320 includes a timer 328. Control signal generator 320 monitors the output signal COMP₁ to determine the time T1 that elapses from when the control signal generator 320 turns switch 3120N until the output signal COMP₁ indicates that voltage V_S0 is greater than the reference voltage V_REF0. Because of the relationships in Equations [1], [2], [3], [4], and [5], when the voltage V_S is greater than the reference voltage V_REF0, the inductor current i_L has reached the target average value i_L—TARGET. For example, in at least one embodiment, control signal generator 320 initiates timer 328 when control signal generator 320 generates a pulse of control signal CS₂ to turn switch 3120N.

[0043] When the secondary-side current i_CTS is less than the reference current i_M/R and, thus, voltage V_S0 is less than the reference voltage V_REF0, the output signal COMP₁ of comparator 324 is a logical 0. When the secondary-side current i_CTS rises to a value greater than the reference current i_REF and, thus, voltage V_S0 is greater than the reference voltage V_REF0, the output signal COMP₁ switches to a logical 1. When the output signal COMP₁ switches to logical 1, control signal generator 320 observes the elapsed time T1 determined by timer 328. Control signal generator 320 utilizes the elapsed time T1, feedback of the input voltage V_X at pin 330, and the link voltage WINK at pin 332 to generate control signal CS₂. Melanson I describes exemplary systems and methods for generating control signal CS₂ using the elapsed time T1. For example, controller 302 can be implemented using analog and/or digital circuits or using analog or digital circuits in combination with a processor executing software instructions stored in a memory.

[0044] The secondary-side winding of current transformer 310 is also connected to pin 334 of controller 302. Diode 336 and voltage V_DD protect pin 334 from rising above V_DD plus a diode voltage drop across diode 336. Diode 338 connects the secondary-side winding to sources of FETs 314 and 316.

[0045] During operation of switching power converter 308, voltage source 340 provides an input voltage V_IN to full-bridge diode rectifier 342. The voltage source 340 is, for example, a public utility, and the AC voltage V_IN is, for example, a 60 Hz/110 V line voltage in the United States of America or a 50 Hz/220 V line voltage in Europe. Voltage source 340 and rectifier 342 represent one embodiment of voltage source 226 (FIG. 2). Voltage V_X is a rectified version of input voltage V_IN and is the input voltage to switching power converter 308. Switching power converter 308 includes capacitor 344 to filter high frequency components of voltage V_X and, thus, provide electromagnetic interference (EMI) protection. Control signal CS₂ turns switch 3120N and OFF. When switch 312 is ON, the inductor current i_L in inductor 346 ramps up, diode 348 is reverse biased, and inductor current i_L flows through the primary-side winding 309 and through switch 312. In at least one embodiment, switch 312 is a FET. When switch 312 is OFF, the inductor current i_L flows through forward biased diode 348, and charges link capacitor 350. Link capacitor 350 holds link voltage V_LINK to an approximately constant value for load 352. Load 352 can be any load. For example, in at least one embodiment, load 352 is a power supply, such as a DC-DC or DC-AC converter. Load 352 can also include, for example, a lighting system that includes one or more light emitting diodes or gas discharge lamps. The specific component values of switching power converter 308 are a matter of design choice and are selected to, for example, allow controller 302 to provide power factor correction and regulate link voltage V_LINK for switching power converter 308 and to allow switching power converter 308 to provide a regulated output voltage to load 352.

[0046] FIG. 4 depicts power control system 400. Power control system 400 includes controller 402, and controller 402 is an integrated circuit that generates a control signal CS₂ to control power factor correction and regulate link voltage V_LINK of switching power converter 308. In at least one embodiment, power control system 400 is identical to power control system 300 except that controller 402 converts the received secondary-side current i_CTS into a voltage V_S1 to sense the secondary-side current i_CTS.

[0047] Comparator 404 compares voltage V_S1 with a reference voltage V_REF1 and generates a comparison signal COMP₂. The voltage V_S1 represents the secondary-side current i_CTS. Thus, controller 402 detects the secondary-side current i_CTS via comparison signal COMP₂. In at least one embodiment, the input impedance of comparator 404 is much larger than the resistance of resistor circuit 406. Resistor circuit 406 represents one embodiment of the voltage converter 219 (FIG. 2). Thus, the voltage V_S1 is approximately the product of the secondary-side current i_CTS and the resistance R of resistor circuit 406, i.e. V_S1≈i_CTS ·R. Resistor circuit 406 represents one embodiment of the voltage converter 219 (FIG. 2).

[0048] The value of the resistance R₄₀₆ of resistor circuit 406 is a matter of design choice and, in at least one embodiment, is set so that the comparison signal COMP₂ changes logical states when the secondary-side current i_CTS equals an average target value i_TARGET. Thus, in at least one embodiment, Equation [6]:
R₄₀₆=V_REF1/i_TARGET  [6].
wherein R₄₀₆ is the resistance of resistor circuit 406, V_REF1 is a reference voltage, and i_TARGET is an average target value of secondary-side current i_CTS. For example, if the target average value of i_CTS is 250 mA and V_REF1 is 1V, R=V_REF1/i_TARGET=4 ohms.

[0049] When the secondary-side current i_CTS is less than the target current i_TARGET, voltage V_S1 is less than the reference voltage V_REF1, and the output signal COMP₁ of comparator 324 is a logical 0. When the secondary-side current i_CTS rises to a value greater than the average target value i_TARGET, voltage V_S1 is greater than the reference voltage V_REF, and the output signal COMP₁ switches to a logical 1. When the output signal COMP₂ switches to logical 1, control signal generator 328 observes the elapsed time T1 determined by timer 328. Control signal generator 320 utilizes the elapsed time T1, feedback of the input voltage V_X at pin 330, and the link voltage V_LINK at pin 332 to generate control signal CS₂ as previously described with reference to controller 302 (FIG. 3).

[0050] Resistor circuit 406 can be any type of resistor circuit. In at least one embodiment, resistor circuit 406 is a single resistor or combination of resistors having a fixed value. In at least one embodiment, resistor circuit 406 is a variable resistor, such as a programmable, variable resistor, whose value can be adjusted in accordance with changes to the target current value i_TARGET.

[0051] FIG. 5 depicts a programmable current source 500, which represents one embodiment of current source 318 (FIG. 3). Programmable current source 500 includes a digital-to-analog converter (DAC) 502 that receives reference current set point data D_ISET. In at least one embodiment, the reference current set point data D_ISET is a digital signal representing a desired value of drain current i_M (FIG. 3). In at least one embodiment, the desired value of drain current i_M is programmed so that switching power converter 308 (FIG. 3) is power factor corrected and regulates the link voltage V_LINK to a desired value. In at least one embodiment, the drain current i_M is calibrated based on the characteristics of power control system 300 that affect the value of, for example, secondary-side current i_CTS. For example, the turns ratio between the primary-side and secondary-side windings of transformer 310 affect the value of secondary-side current i_CTS. Different switching power converters may have different turns ratios. Thus, the drain current i_M can be programmed based on knowledge of the specific switching power converter that will be connected to controller 302. For example, the reference current set point data D_ISET can be programmed to increase or decrease drain current i_M as the respective turns ratios increase and decrease. Additionally, the desired average target inductor current i_L—TARGET may change depending upon, for example, power usage of different loads. Thus, in at least one embodiment, the drain current i_M can be programmed to account for various desired values of target inductor current i_L—TARGET so that Equations [3], [4], and [5] are satisfied to allow the switching power converter 308 to generate the desired average target inductor current i_L—TARGET.

[0052] DAC 502 converts the reference current set point data D_ISET into the analog, constant drain current i_M. The DAC 502 can be any DAC, and the particular implementation of DAC 502 is a matter of design choice. The manner of providing the reference current set point data D_ISET to DAC 502 is a matter of design choice. For example, in at least one embodiment, the reference current set point data D_ISET can be programmed into a memory (not shown) of controller 302 (FIG. 3). In at least one embodiment, the reference current set point data D_ISET is preset in controller 302 or supplied through a data pin (not shown) of controller 302.

[0053] FIG. 6 depicts a programmable resistor circuit 600, which represents one embodiment of resistor circuit 406. Programmable resistor circuit 600 includes N resistors 602.0-602.N−1, where N is an index representing the number of resistors in programmable resistor circuit 600 and N is greater than or equal to 2. In at least one embodiment, the values of resistors 602.0-602.N−1 are set to provide a desired range of values for the target current i_TARGET. From equation [6], since, in at least one embodiment, the reference voltage V_REF is fixed, adjusting the resistance R of resistor circuit 600 adjusts the target current i_TARGET. Switches 604.1-604.N−1 are configured in parallel with respective resistors 602.1-602.N−1. The resistance R of programmable circuit 600 is programmed using switch control signals CR₁-CR_N-1 to control the conductivity of switches 604.1-604.N−1. Controlling the conductivity of switches 604.1-604.N−1 controls the series resistance of programmable resistor circuit 600, and, thus, sets the value of resistance R. In at least one embodiment, switches 604.1-604.N−1 are n-channel field effect transistors, and control signals CR₁-CR_N-1 are gate voltages.

[0054] The manner of setting values control signals CR₁-CR_N-1 is a matter of design choice. In at least one embodiment, controller 402 (FIG. 4) sets the values of control signals CR₁-CR_N-1 in accordance with data received via a data pin (not shown) of controller 302. In at least one embodiment, the values of control signals CR₁-CR_N-1 are fixed upon manufacture of controller 402. The configuration of resistors 602.0-602.N−1 is a matter of design choice. Resistors 602.0-602.N−1 are configured in series but can also be configured in parallel or a combination of series and parallel.

[0055] Thus, in at least one embodiment, a power control system includes a current transformer to step down a switch current of a switching power converter. In at least one embodiment, a controller utilizes current sensing of the stepped down switch current to control a switching power converter. By utilizing current sensing, in at least one embodiment, the controller can reduce component costs.

[0056] Although embodiments have been described in detail, it should be understood that various changes, substitutions, and alterations can be made hereto without departing from the spirit and scope of the invention as defined by the appended claims.