RF power detector

[0001] 1. Field of the Invention

[0002] This following invention relates to power detectors such as those used for wireless applications.

[0003] 2. Description of the Related Art

[0004] Radio frequency (RF) power amplifiers (PAs) are commonly used within the transmit path for wireless applications to boost the transmitted power provided to the antenna. They are typically designed to provide power into a fixed impedance load, such as 50 ohms, and are designed to operate from a fixed supply voltage, such as 3.5V. At its output, the power amplifier (PA) has an output network that is formed from passive components, such as inductors, capacitors, and transformers that are arranged in a tuned configuration. The networks formed from these passive elements are resonant at frequencies near that of the desired radio frequency (RF) carrier. Within this narrow frequency range, the circuit provides impedance transformation. Without impedance transformation, the maximum power that could be delivered by the power amplifier to a fixed load would be limited by the amplifier's supply voltage. With impedance transformation, the power amplifier can provide an output power that is higher than this limit.

[0005] FIG. 1 shows an example of an RF power amplifier and associated output network that are well known in the field. The amplifier's final stage is formed from transistor 102 and inductor 103. Capacitor 109 acts as a DC blocking element and elements 105, 106, and 107 perform impedance transformation, transforming the load impedance 111 from its actual value, R_L, to new impedance, Z_T, seen by the final PA stage. The ratio R_L/Z_T is referred to as the transformation ratio. The load impedance R_L is usually purely real. Components in the amplifier's output network are usually chosen so that Z_T is strictly real as well.

[0006] FIG. 2 shows an example of a common transmit chain for wireless transmitters. The PA's amplification stages, 202, are followed by an output network, 204, which is followed by an isolator, 206, and then a directional coupler, 208. An isolator is a passive device that provides the PA with a more controlled load impedance despite large variations in antenna impedance. The directional coupler 208 generates an output signal, 210, that represents the level of power being transmitted to the antenna. This measure is fed back to the amplifier stages where it is used to control power levels being generated. In some cases, the purpose of this loop is to reduce variations in transmitted power from device to device and over conditions such as temperature. In other cases, this loop is used to assure that the maximum power provided to the antenna remains below a maximum limit as set by safety considerations.

[0007] RF systems have in the past measured forward power of an amplifier. For example, as shown in FIG. 2, the directional coupler 208 may be placed between the power amplifier and the load and the forward power supplied by the power amplifier is measured and used to control a gain of the power amplifier. However, the forward power value does not equate to the real power consumed by the load.

[0008] Thus, the power at the load can be an important criterion in controlling amplifier output.

[0009] Accordingly, an embodiment of the invention provides a method that includes measuring real power delivered to a load in an RF transmit path and utilizing the measured real power to adjust a power amplifier in the RF transmit path. Measuring the real power may further include supplying a first voltage from a first side of a passive network disposed between the power amplifier and the load; supplying a second voltage from a second side of the passive network; and using the first and second voltages and a phase delayed voltage to determine the real power delivered to the load. Determining the real power may further include determining a voltage difference between the first and second voltages and determining a multiplication result of the voltage difference multiplied by the phase shifted third voltage.

[0010] In another embodiment, an apparatus includes one or more passive networks disposed between a power amplifier and the load in an RF transmit path. A real power detector is coupled to utilize a first voltage present on a first side of the one or more passive networks with respect to the power amplifier, and a second voltage present on a second side of the one or more passive networks with respect to the power amplifier, and a phase delayed voltage, to determine the real power consumed by the load. The phase delayed voltage may be derived from one of the first and second voltages or a separate voltage. The apparatus may further include a difference circuit to determine a voltage difference between the first and second voltages and a circuit to determine a multiplication result of the voltage difference multiplied by the phase shifted voltage. The apparatus may further include a low pass filter to filter the multiplication result and supply the real power consumed by the load.

[0011] In another embodiment, a method is provided that includes supplying a first voltage present on a first side of a passive network disposed between the power amplifier and a load in an RF transmit path to a power detector circuit and supplying a second voltage present on a second side of the passive network to the power detector circuit, and determining a voltage difference between the first and second voltages. A voltage is phase shifted to provide a phase shifted voltage, which is then multiplied by the voltage difference between the first and second voltages.

[0028] The use of the same reference symbols in different drawings indicates similar or identical items.

[0029] A methodology to determine the real power consumed by the load according to an embodiment of the invention is illustrated in FIG. 3, which shows a portion of an RF integrated circuit that includes the DC blocking capacitor 301 at the output of the BALUN 303. The real power consumed by the real part of the load 307 is the average of v1(t)*I(t). All the information needed to determine the real power consumed by the load is in v1(t) and v2(t), which is accessible on the RF integrated circuit. Note that CS 301 and LS 309 can be any reactive lossless network.

[0030] Referring now to FIG. 4, generic reactive networks 401 and 403 are shown. The current I(t) passes through nodes 402 and 404. The voltage at node 402 is v1(t) and at node 404 is v2(t). The average of v1(t)*I(t) is the same as v2(t)*I(t) since there is no power loss at reactive networks Z1 or Z2. The average of v1(t)*I(t) is the real power consumed at ZLOAD 307. Real power is the power consumed by the resistive part (real part) of ZLOAD. Note that I(t) has a 90 degree phase relation with v2(t)−v1(t). Note that Z2 can just be a short. V1(t) and I(t) can be defined as follows:
v1(t)=sin(ωt) and  (1)
I(t)=sin(ωt+φ)  (2)
where ω is the carrier frequency and φ is the phase angle between current and voltage.
So the real power is the average of
v1(t)*I(t)=1/2 cos(φ).  (3)
Referring to FIGS. 3 and 4,

[0031] $\begin{array}{cc}\frac{d}{dt}\left(v1-v2\right)=\frac{I\left(t\right)}{C},\mathrm{so}& \left(4\right)\\ v1-v2=\int \frac{I\left(t\right)}{C}dt=-\frac{1}{\omega C}\cos \left(\omega t+\varphi \right).& \left(5\right)\end{array}$

[0032] So (v1−v2) is a 90 degree delayed version of I(t). The real power can be calculated if v1(t) is phase delayed by 90 degrees or advanced by 90 degrees so it properly aligns with v1−v2 (the phase delayed version of I(t)). Referring to FIG. 5, that can be accomplished by passing v1(t) through an integrator 501 to provide the desired 90 degree phase shift. So, the voltage v1(t) is passed through integrator 501 (or a differentiator) and multiplied in multiplier 503 with the difference (v1−v2) of voltages v1 and v2 to give:

[0033] $\begin{array}{cc}\frac{1}{{\omega }^{2}C}\cos \left(2\omega t+\varphi \right)+\frac{1}{{\omega }^{2}C}\cos \left(\varphi \right)& \left(6\right)\end{array}$

[0034] The first term in equation 6 is a high frequency portion and the second term reflects the real power consumed by the load. So passing the multiplication value through the low pass filter 505 filters out the first term and so provides the real power.

[0035] If v1−v2 is across an inductive impedance L instead of C,

[0036] $\begin{array}{cc}\left(v1-v2\right)=L\frac{dI}{dt}=L\omega \cos \left(\omega t+\varphi \right).& \left(7\right)\end{array}$
Integrating v1 we obtain

[0037] $\begin{array}{cc}\frac{1}{\omega }\cos \left(\omega t\right)& \left(8\right)\end{array}$
Multiplying (7) and (8), we obtain
L cos(2ωt+φ)+L cos(φ).  (9)
The first term represents the high frequency component that can be filtered out by a low pass filter.

[0038] Referring to FIG. 6, it can be hard to build a true RF multiplier, so in an alternative approach to obtain the multiplication result, two RMS detectors 601 and 603 are used, which can be easier to implement. As shown in FIG. 6, the value (v1(t)−v2(t)) is summed with the phase delayed v1 in summer 605 and then provided to RMS detector 601. In summer 607, (v2(t)−v1(t)) is summed with the phased delayed v1 and provided to RMS detector 603. The values from the RMS detectors are then subtracted in summer 609 to provide the multiplication result, which is then supplied to low pass filter 505. Thus, block 611 provides the multiply function of multiplier 503 in FIG. 5 as shown in equation 10 below:
(v1_delayed+(v1−v2))²−(v1_delayed−(v1−v2))²=4*v1_delayed*(v1−v2).   (10)
The output from block 611 can be scaled appropriately to obtain the desired multiplication result v1_delayed*(v1−v2) without the fixed gain term of four.

[0039] While the phase delayed voltage shown in FIGS. 3-6 was assumed to be v1, in fact, the choice for the phase delayed voltage can be flexible and can be any of the sampled voltages shown in FIGS. 7-15. FIG. 7 shows the voltage v1(t) being supplied to phase shift block 501 and subsequently to analog processing block 701, which performs the functions shown in blocks 611 and 505 in FIG. 6. Note that the figure is simplified in that the summer 703 provides both the positive and negative values for v1(t)−v2(t). As shown in FIG. 8, the real power calculation can also utilize v2(t) for the phase shifted voltage. FIG. 9 shows the phase shifted voltage can be selected as v(3) at the load side of the second network Z2.

[0040] In addition, the choice for the v1 and v2 can also be flexible. For example, FIG. 10 shows an embodiment in which v1(t) is supplied to the phase shifter 501 and v2(t)−v3(t) is supplied to the analog processing block 701. FIG. 11 shows an embodiment in which v2(t) is supplied to the phase shifter 501 and v2(t)−v3(t) is supplied to the analog processing block 701. FIG. 12 shows an embodiment in which v3(t) is supplied to the phase shifter 501 and v2(t)−v3(t) is supplied to the analog processing block 701.

[0041] Referring to FIG. 13, additional details are shown for the embodiment shown in FIG. 6. More specifically, separate summers 1301 and 1303 are shown, which provide, respectively, v1(t)−v2(t) and v2(t)−v1(t). In another embodiment shown in FIG. 14, v2(t) is supplied to phase shifter 501. FIG. 15 illustrates an embodiment corresponding to FIG. 10.

[0042] Referring again to FIG. 3, once the real power is determined, the determined real power at ZLOAD can be provided to the power amplifier stage 315 in FIG. 3 to adjust the output of the power amplifier. In that way better control of the amplifier using the real power consumed by the load impedance can be achieved.

[0043] The description of the invention set forth herein is illustrative, and is not intended to limit the scope of the invention as set forth in the following claims. For example, although the RF power amplifier described herein is particularly well suited to power amplifiers built in CMOS, it could also be applied to power amplifiers built in other technologies such as GaAs, SiGe, SOI, and LDMOS. Variations and modifications of the embodiments disclosed herein may be made based on the description set forth herein without departing from the scope of the invention as set forth in the following claims.