Synchronous rectifier type series resonant converter for operating in intermittence mode

[0001] This application is the National Phase application of International Application No. PCT/KR2006/001862, filed May 18, 2006, which designates the United States and was published in English. This application, in its entirety, is incorporated herein by reference.

[0002] The present invention relates to a synchronous rectifier type Series Resonant Converter (hereinafter, referred to as SRC) for operating in an intermittence mode, and more particularly to a synchronous rectifier type SRC for operating in an intermittence mode, in which, in a DC/DC converter for converting DC voltage to DC voltage of a predetermined level, a high frequency rectifier of the DC/DC converter is replaced with a synchronous rectifier, a dead time generator is realized by a simple method in order to reduce the switching loss of the synchronous rectifier, conversion efficiency is maximized in low output voltage, and a series resonant circuit of the DC/DC converter is controlled in the intermittence mode, so that no-load characteristic can be improved, conversion efficiency can be improved, and a manufacturing cost can be reduced.

[0003] Nowadays, there exist various circuits for realizing an isolation DC/DC converter. One of them is an SRC.

[0004] FIG. 1 is a diagram illustrating a conventional SRC and a resonant current characteristic curve.

[0005] FIG. 1a illustrates the conventional SRC. The conventional SRC uses resonance of an inductor and a capacitor, and shows good conversion efficiency. The conventional SRC includes DC input power 110, four switching devices (Q1 to Q4) 120, an LC resonance circuit 130 comprised of a resonance inductor 132 and a resonance capacitor 134, a transformer 140, a bridge rectification circuit 150, a capacitor 160, and an output terminal 170. The four switching devices 120 alternatively switch the input voltage of the input power 110 to convert the DC voltage to AC voltage, and transfer the AC voltage to the LC resonance circuit 130. The LC resonance circuit 130 changes the frequency characteristic of the received AC voltage. The transformer 140 has a predetermined turn ratio, converts the AC voltage (i.e. primary voltage), which is received from the LC resonance circuit 130, to voltage of a predetermined level, and thus outputs secondary voltage. The bridge rectification circuit 150 rectifies the AC voltage induced at a secondary-side of the transformer 140 into DC voltage. The capacitor 160 filters the rectified DC voltage and the output terminal 170 outputs the filtered DC voltage. Herein, the inductor constituting the LC resonance circuit 130 may also be separately added, but increased leakage inductance of the transformer 140 may also be used.

[0006] Hereinafter, a process for performing DC/DC conversion by means of such an SRC will be briefly described. The DC voltage inputted through the input power V is converted to the spherical wave pulse voltage of AC, which has positive voltage and negative voltage alternating with each other, through periodic repetition of a process in which one pair of switching devices Q1 and Q4 are turned on during a half period and the other pair of switching devices Q2 and Q3 are turned on during the other half period.

[0007] Then, the spherical wave pulse voltage of AC is transferred to the LC resonance circuit 130 comprised of the resonance inductor 132 and the resonance capacitor 134, and the LC resonance circuit 130 stores and transfers energy.

[0008] Herein, the resonance voltage and the resonance current of the LC resonance circuit 130 have amplitudes changing according to frequencies of the applied spherical wave.

[0009] The transformer 140 converts input current to output current according to its turn ratio, transfers the output current to a secondary-side. The secondary current of the transformer 140 is rectified to DC through the bridge rectification circuit 150, is filtered by the capacitor 160, and then is outputted as output voltage through the output terminal 170.

[0010] Herein, the frequency switched through the afore-described switching devices will be referred to as a switching frequency. As such a switching frequency increases, the sizes of the capacitor 160 for output filtering, the resonance inductor 132, the resonance capacitor 134, etc., can be reduced. Accordingly, it is possible to reduce the size and area of an entire circuit. In the meantime, there exists loss due to overlap of electric current and voltage during an excessive switching interval. Such loss will be referred to as switching loss. As the switching frequency increases, the switching loss also increases and the efficiency of a circuit deteriorates.

[0011] FIG. 1b illustrates the characteristic of resonant current Ir according to a switching frequency F_sw.

[0012] If the switching frequency reaches a resonance frequency

[0013] $\mathrm{Fr}=\frac{1}{2\pi \sqrt{L_r{C}_r}},$
the resonant current Ir is maximized. As the switching frequency becomes greater than the resonance frequency, the resonant current Ir is reduced. However, as the switching frequency becomes less than the resonance frequency, the resonant current Ir is reduced. Due to such a characteristic, the conventional SRC controls output by means of a Pulse Frequency Modulation (hereinafter, referred to as PFM) scheme.

[0014] However, as it can be understood in the characteristic curve of FIG. 1b, one of the disadvantages of such an SRC lies in that the resonant current does not become zero no matter how the frequency increases. That is, the SRC cannot include a no-load state, in which output current becomes zero during switching, as an operation area. In order to improve control characteristics in such a no-load state, an LLC SRC and an LCC SRC have been mainly used in industrial fields.

[0015] FIG. 2 is a circuit diagram illustrating a conventional LLC SRC and LCC SRC for improving a no-load characteristic. FIG. 2a illustrates the conventional LLC SRC, and FIG. 2b illustrates the conventional LCC SRC. As illustrated in FIGS. 2a and 2b, in the LLC SRC, a parallel inductor 210 is added in parallel with the primary winding of the transformer 140. In the LCC SRC, a parallel capacitor 220 is added in parallel with the primary winding of the transformer 140.

[0016] In such a case, entire resonance voltage is divided in proportion to impedance of each part. If the primary voltage of the transformer 140 is less than output voltage, output control in a no-load state is possible because a secondary rectifier diode is not turned on.

[0017] In the LLC SRC, the parallel inductor 210 is added in parallel with the transformer 140. In the LCC SRC, the parallel capacitor 220 is added in parallel with the transformer 140. Each of the LLC SRC and the LCC SRC has the following advantages and disadvantages.

[0018] In an actual the LLC SRC, the leakage inductance of the transformer 140 is used as the resonance inductor L_r. Further, a gap is inserted into the core of the transformer 140 and magnetizing inductance thereof is adjusted, so that the parallel inductor 210 is achieved. That is, the LLC SRC is advantageous in that no additional parts exist except for the resonance capacitor 134.

[0019] However, since electric current flowing through magnetizing inductance always flows in the primary winding of the transformer 140, the primary winding must be relatively thicker than the secondary winding. Therefore, the size of the transformer 140 increases. That is, primary-secondary winding current is not in proportion to the turn ratio.

[0020] In the LCC SRC, if the parallel capacitor 220 is disposed in the secondary winding of the transformer 140, electric current flowing in the parallel capacitor 220 flows in both the primary winding and the secondary winding of the transformer 140. Therefore, the size of the transformer 140 increases as compared with that in the LLC SRC. Accordingly, if the resonance inductor 132 is separately added and the resonance capacitor 134 is disposed in the primary winding, the transformer 140 can be optimally manufactured. However, even in such a case, since the leakage inductance of the transformer 140 is not used, a manufacturing cost may increase.

[0021] As described above, the conventional SRC does not show a no-load state, i.e. cannot control output voltage due to abnormal increase in the output voltage. Further, an SRC capable of controlling a no-load state has been currently provided, but the size of a necessary power device may increase and a manufacturing cost may increase. Accordingly, it is necessary to provide an SRC capable of controlling a no-load state even without additional separate parts, and maximizing power conversion efficiency.

[0022] Therefore, the present invention has been made in view of the above-mentioned problems, and it is an object of the present invention to provide a synchronous rectifier type SRC for operating in an intermittence mode, in which, in a DC/DC converter for converting DC voltage to DC voltage of a predetermined level, a high frequency rectifier of the DC/DC converter is replaced with a synchronous rectifier, a dead time generator is realized by a simple method in order to reduce the switching loss of the synchronous rectifier, conversion efficiency is maximized in low output voltage, and a series resonant circuit of the DC/DC converter is controlled in the intermittence mode, so that no-load characteristic can be improved, conversion efficiency can be improved, and a manufacturing cost can be reduced.

[0023] In order to accomplish this object, according to an aspect of the present invention, A synchronous rectifier type Series Resonant Converter (SRC) for converting input DC voltage to output DC voltage having a level different from that of the input DC voltage, the synchronous rectifier type SRC comprising: an input power for supplying the input DC voltage; an input-side switching unit for switching four input-side switching devices to convert the input DC voltage to AC voltage, the input-side switching unit being connected to the input power, an LC resonance circuit for storing energy in a resonance inductor and a resonance capacitor by means of LC resonance, and outputting resonance voltage, the LC resonance circuit being connected to the input-side switching unit; a transformer with a primary winding and a secondary winding for converting the resonance voltage to voltage of a predetermined level according to a turn ratio to generate secondary voltage, and transferring the secondary voltage through the secondary winding, the primary winding being connected to the LC resonance circuit; an output-side switching unit for switching four output-side switching devices to convert the secondary voltage to the output DC voltage, the output-side switching unit being connected to the secondary winding of the transformer; and a gate driving circuit for detecting a polarity of secondary current flowing in the secondary winding, generating dead time, generating driving signals for driving the output-side switching device of the output-side switching unit according to the polarity, and then controlling turn-on or turn-off of the output-side switching devices, the gate driving circuit being connected to both the secondary winding of the transformer and the output-side switching unit.

[0024] According to another aspect of the present invention, there is provided A method for controlling output in a no-load characteristic or a light-load characteristic in a synchronous rectifier type Series Resonant Converter (SRC) for converting input DC voltage to output DC voltage having a level different from that of the input DC voltage by using a switching device and an LC resonance circuit, the method comprising the steps of: switching the switching device by using an intermittence mode; and controlling the output in the no-load characteristic or the light-load characteristic so that electric current inputted to the LC resonance circuit approximates zero, wherein the intermittence mode corresponds to a switching method for operating the switching device by changing at least one of turn-on time, turn-off time and turn-on period of the switching device.

[0033] The configuration of the present invention may include a control method of an SRC for controlling a no-load state, and a realization method of a synchronous rectifier for maximizing power conversion efficiency in a simple manner. For convenience of description, a method for realizing a synchronous rectifier will be described with reference to FIGS. 3 to 6, and then a method for improving the no-load characteristic of an SRC by means of an intermittence mode control scheme will be described with reference to FIG. 7.

[0034] Herein, a method for realizing a synchronous rectifier type SRC for improving conversion efficiency of an SRC according to a preferred embodiment of the present invention will be described in detail with reference to FIGS. 3 to 6.

[0035] FIG. 3 is a circuit diagram schematically illustrating a synchronous rectifier type SRC according to a preferred embodiment of the present invention.

[0036] Referring to FIG. 3, the synchronous rectifier type SRC includes input power 310, an input-side switching unit 320, an LC resonance circuit 330, a transformer 340, a gate driving circuit 350, an output-side switching unit 360, an output capacitor 370 and an output terminal 380.

[0037] The input power 310 is power supply for supplying DC power.

[0038] The input-side switching unit 320 includes both four switching devices Q1 to Q4 connected in a full bridge configuration, and input-side body diodes D1 to D4 embedded in each of the switching devices Q1 to Q4. The input-side switching unit 320 performs a switching operation with a period corresponding to an applied frequency, converts DC power, which is transferred from the input power 310, to AC power, and transfers the AC power to the LC resonance circuit 330.

[0039] That is, the DC voltage transferred from the input power 310 is converted to voltage of a spherical wave pulse, which alternatively shows positive polarity and negative polarity, through a periodic process in which one pair of switching devices Q1 and Q4 are turned on during a half period and the other pair of switching devices Q2 and Q3 are turned on during the other half period. That is, the secondary current of the transformer 340 flows in the input-side body diodes D1 to D4 embedded in each of the switching devices Q1 to Q4 when each of the switching devices Q1 and Q4 is turned off.

[0040] The input-side switching unit 320 includes a gate driving circuit (not shown) for driving the switching devices Q1 and Q4 in order to cause one pair of switching devices Q1 and Q4 to be turned on during the half period and cause the other pair of switching devices Q2 and Q3 to be turned on during the other half period. Since the gate driving circuit is a conventional gate driving circuit, and is well known to those skilled in the art, details will be omitted. Further, the gate driving circuit is used for switching for an intermittence mode control method according to the preferred embodiment of the present invention that will be described later.

[0041] The LC resonance circuit 330 is connected to the input-side switching unit 320, causes the spherical wave pulse voltage generated by the switching devices Q1 and Q4 of the input-side switching unit 320 to pass, and controls the primary voltage of the transformer 340 by means of LC resonance. That is, the LC resonance circuit 330 adjusts the switching frequency by means of a PFM scheme, controls resonance current I_r and resonance voltage V_r, and thus adjusts the secondary voltage of the transformer 340. Further, in the present invention, the LC resonance circuit is comprised of a resonance inductor 332 and a resonance capacitor 334, which are connected in series with the primary winding of the transformer 340. However, the present invention is not limited to this construction.

[0042] The transformer 340 converts the primary voltage, i.e. the resonance voltage transferred from the LC resonance circuit 330, to secondary voltage according to a turn ratio, and transfers the secondary voltage to the output-side switching unit 360. That is, the transformer 340 converts the input voltage to the voltage of a predetermined level.

[0043] The gate driving circuit 350 detects the polarity of secondary current I_o flowing in the secondary winding of the transformer 340, and controls the output-side switching unit 360 according to the detected polarity. That is, the gate driving circuit 350 automatically turns on or off the switching device Q11, Q12, Q13 or Q14 of the output-side switching unit 360 according to the polarity of the secondary current I_o, instead of turning on or off the switching device Q11, Q12, Q13 or Q14 by an external operation as the gate driving circuit (not shown) of the input-side switching unit 320. This will be described in detail with reference to FIG. 4.

[0044] The output-side switching unit 360 converts the secondary voltage of the transformer 340 to DC voltage. That is, the output-side switching unit 360 rectifies the AC voltage to the DC voltage while the switching devices Q11 and Q14 and the switching devices Q12 and Q13 are periodically turned on or off under the control of the gate driving circuit 350. Specifically, in the preferred embodiment of the present invention, a reverse conductive Metal-Oxide Semiconductor Field Effect Transistor (hereinafter, referred to as MOSFET) is used as the switching device Q11, Q12, Q13 or Q14, instead of a general diode rectifier.

[0045] That is, the synchronous rectifier type SRC for improving the conversion efficiency of an SRC according to the preferred embodiment of the present invention employs the reverse conductive MOSFET as a rectifier, instead of a general diode rectifier, so that it is possible to reduce the conduction loss of a rectifier by using the fact that the MOSFET shows the characteristic of a reverse conductive resistor R_dson if the MOSFET is turned on. For example, if electric current of 10 A flows through a general silicon diode, voltage drop of about 1V occurs, and thus conduction loss of about 10 W (=10 A×1V) occurs. If electric current of 10 A flows through a Schottky diode, voltage drop of about 0.4V occurs, and thus conduction loss of about 4 W (=10 A×0.4V) occurs. However, if synchronous rectification is performed for the MOSFET, reverse conductive resistance of which is 0.01Ω, conduction loss of 1 W (=10 A×10 A×0.01Ω) occurs. Accordingly, it is possible to effectively reduce conduction loss as compared to the case of using the conventional silicon diode.

[0046] As described above, in the output-side switching unit 360, output-side Schottky diodes D11 to D14 are connected in parallel with the switching devices Q11 to Q14, respectively. The reason for respectively connecting the output-side Schottky diodes D11 to D14 to the switching devices Q11 to Q14 as illustrated in FIG. 3 is for preventing the body diodes embedded in the switching devices from being turned on when the switching devices are turned off before the resonance current I_r, which is the secondary current of the transformer 340, becomes zero.

[0047] In FIG. 3, the input-side Schottky diodes are connected to each switching device of the input-side switching unit 320. However, this is for convenience of description. Actually, the input-side Schottky diodes are embedded in the switching devices, respectively. Further, the output-side Schottky diodes are respectively connected in parallel with the switching devices of the output-side switching unit 360. Actually, each of the switching devices includes each of the output-side Schottky diodes therein as in the case of the switching device in the input-side switching unit 320, and the Schottky diodes are respectively connected in parallel with the switching devices in order to reduce reverse recovery time as described later.

[0048] As described above, the body diode of the MOSFET used as the switching device has low reverse conduction loss as compared to the conventional diode switching device, but has long reverse recovery time. Accordingly, it is apparent that rectification loss due to the long reverse recovery time is loss that cannot be passed over. In the meantime, when the Schottky diode is turned on, forward voltage drop is less than that of the body diode in the MOSFET.

[0049] According to the preferred embodiment of the present invention, if the Schottky diode is connected in parallel with the MOSFET that is the switching device, the secondary current of the transformer 140 flows through the Schottky diode, instead of the body diode in the MOSFET, when the MOSFET is turned off. Herein, since the Schottky diode has reverse recovery time approximating zero, rectification loss can be reduced.

[0050] The output capacitor 370 performs filtering for smoothing the output voltage of the output-side switching unit 360, and transfers the filtered voltage to the output terminal 380. In FIG. 3, the capacitor C_o is connected, but the present invention is not limited to this construction.

[0051] The output terminal 380 is connected to a load, and outputs load voltage. In FIG. 3, the resistor R_o is connected, but the present invention is not limited to this construction.

[0052] The synchronous rectifier type SRC according to the preferred embodiment of the present invention includes the gate driving circuit 350 as described above.

[0053] FIG. 4 is a circuit diagram schematically illustrating the gate driving circuit according to the preferred embodiment of the present invention.

[0054] As illustrated in FIG. 4, the gate driving circuit 350 is connected to both the secondary winding of the transformer 340 in the synchronous rectifier type SRC and the gates and sources of the switching devices Q11 to Q14 in the output-side switching unit 360, and controls the operations of each of the switching devices Q11 to Q14.

[0055] The gate driving circuit 350 according to the preferred embodiment of the present invention includes a Current Transformer (CT) 410, a dead time generator 420, a bridge diode 430, a comparator 440 and a gate driver 450.

[0056] The primary winding of the current transformer 410 is connected to the secondary winding of the transformer 340, and converts the secondary current I_o of the transformer 340 to electric current of a predetermined level according to the turn ratio of the current transformer 410. The secondary winding of the current transformer 410 is connected to both the four diodes Dg1 to Dg4 of the bridge diode 430 and the comparator 440, and converts the converted current to the bridge diode 430 and the comparator 440. That is, the current transformer 410 converts the secondary current I_o of the transformer 340 to electric current suitable for being used by the bridge diode 430, the comparator 440 and the gate driver 450, and transfers the converted current to the bridge diode 430 and the comparator 440.

[0057] The dead time generator 420 is a resistor, is connected in parallel with the secondary winding of the current transformer 410, and reduces voltage transferred to the bridge diode 430, thereby adjusting input voltage of each diode in the bridge diode 430, which are turned on or off by itself, according to the polarity of the secondary current of the current transformer 410.

[0058] That is, voltage inputted to the bridge diode 430 is reduced according to the resistance value of the dead time generator 420, and thus each of the diodes Dg1 to Dg4 in the bridge diode 430 is alternatively turned on according to the polarity of input voltage, which is greater than the threshold voltage of the comparator 440, so that electric current flows in the secondary winding of the current transformer 410.

[0059] Accordingly, only when the secondary current I_o of the transformer 340, i.e. the primary current of the current transformer 410, has an absolute value of more than a predetermined level from zero, the secondary current of the current transformer 410 based on its turn ratio applies voltage of more than threshold voltage to each of the diodes Dg1 to Dg4 in the bridge diode 430. As a result, it is possible to prevent both the switching devices Q11 and Q14 and the switching devices Q12 and Q13 in the output-side switching unit 360 from being simultaneously turned on. That is, it is possible to generate dead time at which both the switching devices Q11 and Q14 and the switching devices Q12 and Q13 are simultaneously turned off.

[0060] In the conventional synchronous rectifier type SRC, a complicated circuit for generating dead time must be separately embedded in the gate driver 450. However, in the present invention, a resistor R_g is connected in parallel with the secondary-side of the current transformer in order to generate dead time, so that it is possible to simplify the construction of a gate driving circuit (i.e. the gate driving circuit 350) for driving the output-side switching unit 360. Consequently, it is possible to obtain the effect of cost reduction.

[0061] The dead time generator 420 according to the preferred embodiment of the present invention may also be realized by directly connecting a resistor R_dg in parallel with the current transformer 410 as illustrated in FIG. 4a. However, the dead time generator 420 may also be realized by connecting a resistor 422 in parallel with the diode Dg3 and connecting a resistor 424 in parallel with the diode Dg4 as illustrated in FIG. 4b.

[0062] In the bridge diode 430, four diodes are connected to the secondary winding of the current transformer 410. That is, the Zener diodes Dg1 and Dg3 are connected to one part of the secondary winding of the current transformer 410, and the diodes Dg2 and Dg4 are connected to the other part of the secondary winding of the current transformer 410. If the voltage having passed through the dead time generator 420 has an absolute value greater than the threshold voltage of the comparator 440, the bridge diode 430 causes the comparator 440 to detect the polarity of the secondary current of the current transformer 410, thereby determining the polarity of the secondary current I_o of the transformer 340.

[0063] That is, in a case in which the input voltage is greater than the threshold voltage of the comparator 440, if the input voltage has a positive polarity, the bridge diode 430 causes the switching devices Q11 and Q14 to be turned on. If the input voltage has a negative polarity, the bridge diode 430 causes the switching devices Q12 and Q13 to be turned on. In this way, the comparator 440 operates.

[0064] The comparator 440 is connected to both the diodes Dg1 to Dg4 in the bridge diode 430 and the gate driver 450, detects the polarity of the secondary current, and transfers driving signals to the gate driver 450 according to the detected polarity.

[0065] That is, the comparator 440 detects the polarity of the secondary current of the current transformer 410. If the secondary current has a positive polarity, the comparator 440 simultaneously transfers driving signals to a Q11 gate driver 452 and a Q14 gate driver 458 in order to turn on the switching devices Q11 and Q14 of the output-side switching unit 360. If the secondary current has a negative polarity, the comparator 440 simultaneously transfers driving signals to a Q12 gate driver 454 and a Q13 gate driver 456 in order to turn on the switching devices Q12 and Q13 of the output-side switching unit 360.

[0066] The gate driver 450 includes the Q11 gate driver 452, the Q12 gate driver 454, the Q13 gate driver 456 and the Q14 gate driver 458. The gate drivers 452, 454, 456 and 458 are connected to the comparator 440, are connected to the gates of the switching devices Q11 to Q14 of the output-side switching unit 360, respectively, and receive the driving signals from the comparator 440, thereby turning on or off the switching devices Q11 to Q14 of the output-side switching unit 360 according to the driving signals as described above.

[0067] To this end, each of the gate drivers 452, 454, 456 and 458 includes an amplifier in order to generate large gate driving current from the small driving signals received from the comparator 440. Further, each of the gate drivers 452, 454, 456 and 458 includes an isolator for isolating the switching devices Q11 and Q12, and isolating the driving signals when source voltage is different from the ground voltage of a control circuit in the case of the switching devices Q13 and Q14. The conventional gate driver 450 must have a separate dead time generator for generating dead time. However, the gate driver 450 according to the preferred embodiment of the present invention generates dead time through the dead time generator 420 without a separate device for generating the dead time.

[0068] FIG. 5 is a circuit diagram illustrating an operation waveform of the synchronous rectifier type SRC according to the preferred embodiment of the present invention.

[0069] FIG. 5 illustrates voltage and electric current outputted from main elements of the synchronous rectifier type SRC according to the preferred embodiment of the present invention as shown in FIG. 3. For convenience of description, it is assumed that the input power 310 outputs voltage V_dc, the gate-source voltage of the switching devices Q1 to Q4 of the input-side switching unit 320 are V_q1, V_q2, V_q3 and V_q4, respectively, the drain-source voltage of the switching device Q3 is V_ds3, the drain-source voltage of the switching device Q4 is V_ds4, the output voltage of the LC resonance circuit 330 is resonance voltage V_r, the output current of the LC resonance circuit 330 is resonance current I_r, the secondary voltage of the transformer 340 is V_tx, the secondary current of the transformer 340 is I_o, the gate-source voltage of the switching devices Q11 to Q14 of the output-side switching unit 360 are V_q11, V_q12, V_q13 and V_q14, respectively, and the output voltage of the output-side switching unit 360 is V_o.

[0070] FIG. 6 is a diagram illustrating an operation waveform of the synchronous rectifier type SRC according to the preferred embodiment of the present invention.

[0071] Hereinafter, the operation of the synchronous rectifier type SRC through output waveforms from the main elements of the synchronous rectifier type SRC according to the preferred embodiment of the present invention will be described with reference to FIGS. 5 and 6.

[0072] As illustrated in FIG. 6, the time frame is divided into t₀ to t12 based on points of time when the input-side switching unit 320 and the output-side switching unit 360 are switched, respectively.

[0073] As input power voltage Vdc is applied to the input-side switching unit 320 and thus the switching devices Q1 and Q4 of the input-side switching unit 320 are turned on, V_ds3 is outputted as V_dc and V_ds4 is outputted as 0. At the time point t₀, since the resonance current I_r has a positive value, Q11 and Q14 are turned on and V_o is outputted as V_tx. Herein, if Q1 and Q4 are turned off, V_ds3 starts to fall and V_ds4 starts to rise. At the time point t₁, V_ds3 has a value of 0 and V_ds4 becomes V_dc.

[0074] At the time point t₁, the resonance current I_r flows through the input-side body diodes D2 and D3. While the resonance current I_r flows through the input-side body diodes D2 and D3, if the switching devices Q2 and Q3 are turned on, zero voltage switching having switching loss of 0 is achieved (since the conduction voltage of the input-side body diode is sufficiently less than V_dc, switching loss is regarded as 0).

[0075] At the time point t₂, if Q2 and Q3 are turned on, since the resonance current I_r has a positive value, Q11 and Q14 remains in a turned-on state. However, as energy stored in the resonance inductor L_r is reduced, the value of the resonance current I_r is reduced. Then, just before the value of the resonance current I_r becomes zero, i.e. at the time point t₃, Q11 and Q14 are turned off by the dead time generator 420. That is, since the voltage applied to the bridge diode 430 becomes less than the turn-on threshold voltage of the comparator 440 by the dead time generator 420, the comparator 440 is turned off, and thus Q11 and Q14 are turned off. Herein, the secondary current I_o of the transformer 340 flows through the output-side Schottky diodes D11 and D14 accordingly, and thus zero voltage switching having switching loss of 0 is achieved.

[0076] At the time point t₃, Q11 and Q14 are turned off, and the resonance current I_r is gradually reduced until it reaches a value of 0. This time point corresponds to the time point t₄. Up to this time point, the secondary current I_o of the transformer 340 flows through the output-side Schottky diodes D11 and D14.

[0077] If the time point t₄ is reached and the resonance current I_r has a value of 0, the resonance current I_r has a negative value. Therefore, the secondary current I_o of the transformer 340 does not flow through the output-side Schottky diodes D11 and D14 of the output-side switching unit 360. Then, if V_tx is reduced until it reaches a value of 0, the secondary current I_o of the transformer 340 flows through the output-side Schottky diodes D12 and D13.

[0078] While the secondary current I_o of the transformer 340 flows through the output-side Schottky diodes D12 and D13, if the value (negative value) of the resonance current I_r increases and the voltage inputted to the bridge diode 430 becomes greater than the threshold voltage of the comparator 440, the switching devices Q12 and Q13 of the output-side switching unit 360 are turned on. This time corresponds to the time point t₅.

[0079] At the time point t₅, if the switching devices Q12 and Q13 of the output-side switching unit 360 are turned on, V_tx is outputted as a value of −V_o. While this value is maintained, if the switching devices Q2 and Q3 of the input-side switching unit 320 are turned off at the time point t₆, V_ds3 increases and V_ds4 decreases because the resonance current I_r has a negative value. Accordingly, at the time point t₇, V_ds3 becomes V_dc and V_ds4 becomes zero.

[0080] From the time point t₇, the resonance current I_r flows through the input-side body diodes D1 and D4. At the time point t₇, if the switching devices Q1 and Q4 of the input-side switching unit 320 are turned on, zero voltage switching is possible. The time point at which the switching devices Q1 and Q4 of the input-side switching unit 320 are turned on corresponds to the time point t₈.

[0081] From the time points t₅ to t₉, V_tx is outputted as a value of −V_o because the resonance current I_r has a negative value. If the resonance current I_r has a positive value, V_tx is outputted as a value of +V_o. Herein, if the resonance current I_r approximates zero from the negative value (t₉ to t₁₀) the switching devices Q12 and Q13 of the output-side switching unit 360 are turned off by the dead time generator 420. If the resonance current I_r increases to a positive value from zero, V_tx also increases. If V_tx becomes +V_o, D11 and D14 are turned on. If the resonance current has a value of more than a predetermined level, the switching devices Q11 and Q14 of the output-side switching unit 360 are turned on by the comparator 440. This time corresponds to the time point t₁₁. From this time point, V_tx outputs a value of +V_o.

[0082] While V_tx outputs +V_o, the switching devices Q1 and Q4 of the input-side switching unit 320 are turned off the time point t₁₂. Then, the afore-described processes are repeated, and details will be omitted.

[0083] Hereinafter, an intermittence mode control method for controlling the no-load characteristic in the synchronous rectifier type SRC according to the preferred embodiment of the present invention will be described in detail with reference to FIG. 7.

[0084] The most general method for controlling the SRC as described in FIG. 1 corresponds to the PFM scheme as described above. However, when the SRC is controlled by the PFM scheme, it is impossible to control the SRC in a no-load state as described above. Of course, if a frequency is allowed to increase, the operation range of electric current is widened. However, since it is very difficult to infinitely increase a switching frequency in the switching device 120, which is a Switch Mode Power Supply (SMPS), and a manufacturing cost also increases considerably, it is impossible to infinitely increase the switching frequency in the switching device 120. In order to solve these problems, the present invention employs an intermittence mode for use.

[0085] FIG. 7 is a diagram illustrating an operation method of an intermittence mode according to the preferred embodiment of the present invention.

[0086] A frequency control method according to the preferred embodiment of the present invention basically uses a PFM scheme. However, in order to accomplish control in a no-load state, an intermittence mode or a hiccup mode has been introduced.

[0087] In an intermittence mode, an entire interval is divided into an operation interval and a non-operation interval, and a switching device operates, instead of consequently operating the switching device by changing turn-on time and period of the switching device. For example, if output current corresponds to 20% of maximum current in a maximum operable frequency, the switching device operates with output current, which corresponds to 20% of the maximum current, during 50% of entire time and does not operate during remaining 50% of the entire time in order to reduce the output current to 10% of the maximum current. That is, in order to reduce the output current of the SRC, when the same switching frequency is used, one or more of the turn-on time, the turn-off time and the turn-on period of the switching device are altered, so that the running time of the SRC can be reduced by 50%.

[0088] In this way, when the SRC is controlled in the intermittence mode, invalid power flowing in the parallel parts (L_p and C_p) of the LLC SRC and the LCC SRC as described in FIG. 2 becomes zero, so that conduction loss can be reduced and thus conversion efficiency can be improved. In addition, the size of the transformer can be reduced and thus a manufacturing cost can be saved.

[0089] Such an intermittence mode may be accomplished in various ways as illustrated in FIG. 7. FIG. 7a shows a constant period scheme. In the constant period scheme, the turn-on periods of the switching devices Q1 to Q4 in the input-side switching unit 320 are constantly maintained, and only ratios of turn-on time and turn-off time are altered.

[0090] FIG. 7b shows a constant on time scheme. In the constant on time scheme, the turn-on periods and turn-on time of the switching devices Q1 to Q4 in the input-side switching unit 320 are constantly maintained, and the turn-off time is altered.

[0091] FIG. 7c shows a constant off time scheme. In the constant off time scheme, the turn-on periods and turn-off time of the switching devices Q1 to Q4 in the input-side switching unit 320 are constantly maintained, and the turn-on time is altered.

[0092] FIG. 7d shows a variable period/time scheme. In the variable period/time scheme, all of the turn-on periods, turn-off time and turn-on time of the switching devices Q1 to Q4 in the input-side switching unit 320 are altered.

[0093] In the intermittence mode control method according to the preferred embodiment of the present invention, the switching devices Q1 to Q4 in the input-side switching unit 320 are turned on or off by using the constant period scheme, the constant on time scheme, the constant off time scheme or the variable period/time scheme, so that electric current inputted to the LC resonance circuit 330 is controlled to approach zero and the no-load characteristic and light-load characteristic are controlled. Consequently, the intermittence mode control method is used for the conventional SRC as well as the synchronous rectifier type SRC as described in FIGS. 3 to 6, so that the no-load characteristic can be controlled.

[0094] According to the present invention, an intermittence mode control method is accomplished only with a simple operation of a switching device. That is, a no-load characteristic can be controlled with an easy scheme and a simple construction by solving disadvantages in that the conventional SRC cannot control the no-load characteristic or a predetermined apparatus has a complicated construction and becomes expensive in order to control the no-load characteristic.

[0095] Further, for achieving zero voltage switching, a simple resistor is added to a gate driving circuit for driving a switching device, instead of embedding a separate circuit for generating dead time in the gate driving circuit, so that it is possible to generate the dead time. Consequently, it is possible to simply reduce switching loss that may occur in the zero voltage switching.