High attenuation filtering circuit for power converters

[0001] This application claims the benefit of U.S. Provisional Patent Application No. 60/873,823, filed on Dec. 8, 2006, the entire disclosure of which is incorporated herein by reference.

[0002] This invention relates to power supplies, and more particularly to low-noise or low-ripple power converters.

[0003] Much of the advance in standard of living over the past twenty or so years results from the use of advanced communications, data processing, and environmental sensing techniques. The devices used in such communications, processing, and sensing generally become more useful as their sizes are decreased, such that more of them can be used. For example, computers and cellular phones require ever-smaller elements, and become more capable as the number of devices, which can be accommodated, increases. Similarly, lightweight and reliable sensors can be used in large numbers in vehicles to aid in control and, in the case of spacecraft and military vehicles, to aid in carrying out their missions.

[0004] Most modern semiconductor devices, and other devices important for the above purposes, are generally energized or biased by direct voltages. As devices have become smaller, their powering requirements also advantageously decrease. Unfortunately, a concomitant of low power requirements is often sensitivity to unintended noise or fluctuations in the applied power. It is easy to understand that extremely small transistors, which ordinarily operate at two or three volts, could be destroyed by application of tens of volts. It is less apparent but true that small-percentage variations or noise on the applied powering voltage may result in degradation of the operating characteristics of semiconductor and other devices and the circuits in which they operate, which may adversely affect the performance. It is a commonplace that conventional radio and television receivers will respond to noise on or sudden changes in their supply voltages with aural or visual distortions, or both.

[0005] In general, electronic equipments require direct voltages for their power sources. There are two general sources of electrical energy which can be used to provide the power, and these two sources are batteries, which provide direct voltage, and power mains of an alternating voltage. When power mains are the source of electrical energy, it is common to rectify the alternating voltage to achieve a direct voltage. The power mains are used to drive machine motors in addition to electronic equipment, so the mains voltages tend to be higher than the voltages required for electronic equipment, and rectified voltages also tend to be higher than desired or usable. In the past, transformers have been used to convert the mains power to voltages more compatible with electronic equipment. However, transformers operating at 60 Hz tend to be much larger than is desirable in modern miniaturized equipment. It might be thought that there are no problems with the powering of electronic equipment from batteries, which directly provide direct voltage. However, batteries have the same general problem as that of mains powering, namely that available direct voltage does not necessarily correspond with the desired operating voltage. One modern technique for producing voltages for powering electronic equipment is that of use of a switching power supply or switching converter, which changes a direct source voltage to a different direct voltage.

[0006] A switching power converter can operate from a direct voltage derived from the power mains or from a battery, and can either increase or decrease the output voltage relative to the input voltage. These switching power converters take many different forms, some examples of which include those described in U.S. Pat. No. 4,163,926 issued Aug. 7, 1979 in the name of Willis; U.S. Pat. No. 4,190,791, issued Feb. 26, 1980 in the name of Hicks; U.S. Pat. No. 4,298,892 issued Nov. 3, 1981 in the name of Scott; U.S. Pat. No. 4,761,722 issued Aug. 2, 1988 in the name of Pruitt; and U.S. Pat. No. 5,602,464 issued Feb. 11, 1997 in the name of Linkowski et al.

[0007] Radar, sensor, imaging, and other systems require power converters that produce very clean power. Ripple attenuation and cancelling circuits have been developed to achieve such ultra-low noise power converters. These prior art filter circuits, however, use many matched inductors and capacitors that must be cascaded together to achieve high attenuation filtering. Hence, undesirable parasitic components are difficult-to-impossible to cancel out in these circuits. Moreover, prior art filter circuits require many electrical components, and matching of components. All inductors in such prior art filter circuits must carry the full load current, and these filter circuits are extremely difficult to damp out to assure stable operation.

[0008] U.S. Pat. No. 6,693,805 discloses a ripple cancellation circuit that addresses many of the problems described above and achieves relatively high attenuation filtering. Unfortunately, the ripple cancellation circuit described in U.S. Pat. No. 6,693,805 is sensitive to magnetic component value matching and the attenuation thereof could be improved.

[0009] Accordingly, a power converter ripple attenuation or cancellation filtering circuit with very high ripple attenuation or cancellation filtering is needed, which avoids the problems associated with prior art ripple filtering circuits.

[0010] Disclosed herein is a power converter. The power converter comprises an output filter circuit including first and second inductive elements; a voltage source coupled to the output filter circuit, the voltage source for generating a voltage across the output filter circuit, the voltage including an alternating voltage component, the alternating voltage component causing the application of an alternating current to the first inductive element of the output filter circuit; and an attenuation filtering circuit.

[0011] The attenuation filtering circuit disclosed herein comprises a first magnetically coupled inductive arrangement for generating a first opposing alternating current in response to the alternating voltage applied to the first inductive element of the output filter circuit; a first combining element for combining the first opposing alternating current with the alternating current to partially cancel the alternating current, an intermediate capacitor, thereby producing an attenuated alternating current with a direct current component applied to the second inductive element of the output filter circuit; a second magnetically coupled inductive arrangement for generating a second opposing alternating current in response to the attenuated alternating current applied to the second inductive element of the output filter circuit; and a second combining element for combining the second opposing alternating current with the attenuated alternating current to substantially cancel the attenuated alternating current, leaving a large direct component of output voltage, with a small alternating component of voltage.

[0012] Also disclosed herein is a method for attenuation of an alternating current component of a power converter. The method comprises generating a first opposing alternating current in response to an alternating current applied to a first inductive element of an output filter circuit of the power converter circuit; combining the first opposing alternating current with the alternating current applied to the first inductive element to partially cancel the alternating current, an intermediate capacitor connected between the first and second stages, thereby producing an attenuated alternating current applied to a second inductive element of the output filter circuit of the power converter circuit; generating a second opposing alternating current in response to the attenuated alternating current applied to the second inductive element of the output filter circuit of the power conversion circuit; and combining the second opposing alternating current with the attenuated alternating current to substantially cancel the attenuated alternating current.

[0022] FIG. 1 is a simplified schematic diagram showing an exemplary embodiment of a power converter. The power converter, denoted generally by reference numeral 10, comprises a voltage source 20 and an output filter circuit 30 including an attenuation filtering circuit 40. The switching action of the power converter connected to the voltage source 20 produces a pulsating voltage V_x(t), which is applied to the output filter circuit 30. The pulsating voltage V_x(t) includes a DC voltage component with an AC voltage (ripple voltage) component superimposed thereon. The output filter circuit 30 includes a high attenuation filtering circuit 40 for removing the ripple voltage.

[0023] As shown in FIG. 1, the voltage source 20 comprises a voltage supply 21, a phase-shifted full-wave switched bridge circuit 22, a transformer 23, and a rectifying arrangement 24. The phase-shifted full-wave switched bridge circuit 22 includes first and second tap points 22₁, 22₂ across which an alternating voltage is generated. The transformer 23 includes a primary winding 23₁ connected to the first and second tap points 22₁, 22₂. The transformer 23 also includes a secondary winding formed by first and second windings 23_2a, 23_2b across which a varying voltage is generated in response to the alternating voltage. More specifically, the phase-shifted full-wave bridge circuit 22 includes a plurality of controllable switches Q1, Q2, Q3, and Q4. As known to those skilled in the art, the switches Q1, Q2, Q3, and Q4 can be operated in a number of modes and are operated by a controller (not illustrated) in a phase-shifted mode, in which the switches are rendered conductive in a manner that minimizes the voltages across the switches during at least one of turn-on and turn-off. The result of these operations produces the alternating voltage across the first and second tap points 22₁, 22₂ and therefore, across the primary winding 23₁ of the transformer 23, which in turn, produces the varying voltage across the secondary windings 23_2a, 23_2b of the transformer 23

[0024] The rectifying arrangement 24 includes first and second diodes or rectifiers 24₁, 24₂ having cathodes connected together at a third tap point 25₁. Anodes of the rectifiers 24₁, 24₂ are connected to ends of the secondary windings 23_2a 23_2b, which are remote from a fourth tap point 25₂. The rectifying arrangement 24 converts the varying voltage across the first and second windings 23_2a, 23_2b into the pulsating direct voltage V_x(t) at the third and fourth tap points 25₁ 25₂.

[0025] In one exemplary embodiment, a 0 volt/100 volt 1 MHz pulsating direct voltage V_x(t) can be generated at the third and fourth tap points 25₁ 25₂ with a 300 volt voltage supply 21 and a transformer 23 having a turns ratio of 3:1.

[0026] The pulsating direct voltage V_x(t) at the third and fourth tap points 25₁ 25₂ is applied across terminals 31₁, and 31₂ of the output filter circuit 30. FIGS. 2A, 2B, and 2C show plots representing the pulsating direct voltage V_x(t) applied across terminals 31₁, and 31₂ of the output filter circuit 30 over time, at different time intervals to t₀ t₁, t₀′, to t₁′, t₀″ to t₁″, and t₀′″ to t₁′″, corresponding respectively to the on/off switching states of the switches Q1, Q2, Q3, and Q4 of the phase-shifted full-wave bridge circuit 22 of the voltage source 20.

[0027] Referring again to the exemplary embodiment of FIG. 1, the output filter circuit 30 comprises an LC output filter formed by an inductive arrangement connected in series with an output filter. The series connected inductive arrangement and an output filter are connected across terminals 31₁, and 31₂ for receiving the pulsating direct voltage V_x(t). In the exemplary embodiment of FIG. 1, the inductive arrangement comprises a first main inductor L_M1 connected in series with a second main inductor L_M2 and the output filter comprises a capacitor C_OUT. In other embodiments, the inductive arrangement may comprise more than two main inductors and/or the output filter may comprise other types of filter structures.

[0028] The attenuation filtering circuit 40 includes first and second transformers T₁ and T₂, and an intermediate capacitor C₁. The first transformer T₁ includes a primary winding N_PT1 and a secondary winding N_ST1, poled as indicated in FIG. 1. The primary winding N_PT1 is connected across the first main inductor L_M1. The first main inductor L_M1 and the transformer T₁ may be constructed as a single coupled inductor element. A first blocking capacitor C_B1, a first auxiliary inductor L_A1, and a first resistor R₁, used for damping, are serially connected with the secondary winding N_ST1. The transformer T₂ includes a primary winding N_PT2 and a secondary winding N_ST2, poled as indicated in FIG. 1. The primary winding N_PT2 is connected across the second main inductor L_M2. A second blocking capacitor C_B2, a second auxiliary inductor L_A2, and a second resistor R₂, used for damping, are serially connected with the secondary winding N_ST2. The first main inductor L_M1, first transformer T₁, first blocking capacitor C_B1, first auxiliary inductor L_A1 and first resistor R₁ form a first filter stage 40_S1, of the attenuation filtering circuit 40. The second main inductor L_M2, second transformer T₂, second blocking capacitor C_B2, second auxiliary inductor L_A2 and second resistor R₂ form a second filter stage 40_S2 of the attenuation filtering circuit 40. The first and second filter stages 40_S1, 40_S2 of the attenuation filtering circuit 40 are connected by the series connected intermediate capacitor C₁ and a damping resistor R₁ arrangement, which form an intermediate filter stage.

[0029] As shown in FIGS. 2A, 2B, and 2C, each voltage pulse in the intervals to t₀ t₁, t₀′ to t₁′, t₀″ to t₁″, and t₀′″ to t₁′″, increases the electrical currents through the first and second main inductors L_M1 and L_M2, as illustrated by plots IL_M1 (FIG. 2A) and IL_M2 (FIG. 2B and FIG. 2C (shows voltage and currents on an expanded voltage and current scales)), respectively. In this context, IL_M1 represents the magnetizing or inductive current component flowing in the first main inductor L_M1, and IL_M2 represents the magnetizing or inductive current component flowing in the second main inductor L_M2. The increasing current flows through the first and second main inductors L_M1 and L_M2 in the intervals t₀ to t₁, t₀′ to t₁′, t₀″ to t₁″, and t₀′″ to t₁′″, flows as current I_{STAGE 2 OUT} through the output filter capacitor C_OUT. Since the output filter capacitor C_OUT is relatively large, its AC voltage is small and most time varying currents flow therethrough. As known to those skilled in the art, the flow of increasing current generally results in an increasing output voltage V_OUT across the output filter capacitor C_OUT, although the current drawn by the load, represented by resistor R_L in FIG. 1, may under some conditions exceed the inductor current, thereby resulting in a net reduction of V_OUT. The voltage across output filter capacitor C_OUT is the voltage available to supply the load represented by resistor R_L.

[0030] Referring collectively to FIG. 1 and FIGS. 2A and B, the pulsating direct voltage V_x(t) applied across the series combination of the first and second main inductors L_M1 and L_M2 and output filter capacitor C_OUT may be viewed as including a DC voltage component with an AC voltage component superposed thereon. The inductive arrangement and output filter capacitor C_OUT may be viewed as a voltage divider, in which case the direct voltage component of pulsating direct voltage V_x(t) may be viewed as being developed solely across the output filter capacitor C_OUT, as in steady-state operation, the inductive arrangement cannot develop or withstand a direct voltage. In this voltage divider view, the alternating component of the pulsating direct voltage V_x(t) may be viewed as appearing across the inductance of inductive arrangement, assuming that output filter capacitor C_OUT has zero impedance. However, filter capacitors have undesirable equivalent series resistance (ESR) and equivalent series inductance (ESL), so some portion of the applied alternating voltage component of pulsating direct voltage V_x(t) will appear across output filter capacitor C_OUT. This alternating voltage component portion of the pulsatory direct voltage V_x(t), i.e., ripple voltage undesirably manifests itself across the load R_L. Viewed another way, the alternating voltage component portion of the pulsatory direct voltage V_x(t) applied to or across the inductive arrangement, results in an alternating current flow I_A therein that flows mostly through the internal impedance of output filter capacitor C_OUT. This alternating current flow I_A generates an undesired ripple voltage which appears across the load R_L.

[0031] In accordance with the principles of the disclosure, the alternating current flow I_A resulting from the alternating voltage component of the pulsating direct voltage V_x(t), developing an alternating voltage component across L_M1. This alternating component of voltage is applied across the winding N_PT1, and by transformer action, an alternating component of voltage appears across N_ST1, this in turn develops an alternating voltage across L_A1. The alternating voltage across L_A1 induces a current IL_A1, and by transformer action develops a current IN_PT1. The alternating current IL_A1 has a magnitude and phase (or polarity) which will cancel the alternating characteristic of the sum of the alternating currents IL_M1 and IN_PT1 respectively flowing through the first main inductor L_M1 and primary winding N_PT1. Accordingly, the first additional alternating current IL_A1 and the alternating currents IL_M1 and IN_PT1 combine together to form current I_STAGE1OUT with a substantially reduced alternating characteristic.

[0032] The current I_STAGE1OUT is filtered by the intermediate capacitor C₁ and its damping resistor R₁ of the intermediate filter stage, and an intermediate voltage V_Y across the C₁-R₁ intermediate filter stage is the input voltage to the second filter stage. The voltage V_Y across the C₁-R₁ intermediate filter stage is predominantly a direct voltage with a small alternating voltage component. The current for the second filter stage flows through the second main inductor L_M2, developing a small alternating component of voltage across L_M2. This alternating component of voltage is applied across the winding N_PT2, and by transformer action, an alternating component of voltage appears across N_ST2, this in turn develops an alternating voltage across L_A2. The alternating voltage across L_A2 induces a current IL_A2, and by transformer action develops a current IN_PT2. The alternating current IL_A2 has a magnitude and phase (or polarity) which will operate to substantially cancel any remaining alternating characteristic in the sum of the currents IL_M2 and IN_PT2 respectively flowing through the second main inductor L_M2 and primary winding N_PT2. Accordingly, the second additional current IL_A2 and the currents IL_M2 and IN_PT2 combine together to form current I_STAGE2OUT with virtually no alternating characteristic.

[0033] In effect, the output filter capacitor C_OUT “sees” only the direct current flow I_STAGE2OUT because the alternating current components flowing in the main inductors L_M1 and L_M2, the primary windings N_PT1 and N_PT2, and the auxiliary inductors L_A1 and L_A2, add to nearly zero. Since almost no alternating current component flows through the internal impedance of output filter capacitor C_OUT, substantially no ripple voltage is generated across the output filter capacitor C_OUT.

[0034] The main inductors L_M1, L_M2 of the output filter circuit 30 carry a large DC load current, in addition to the small AC ripple current. The transformers T₁ and T₂ of the attenuation filtering circuit 40, however, are typically very small and carry virtually no DC current. In one exemplary embodiment, the transformers T₁ and T₂ and may each be formed by a single additional turn around their respective main inductor L_M1 and L_M2. In other exemplary embodiments, the transformers T₁ and T₂ and may each be formed by multiple turns around their respective main inductor L_M1 and L_M2. The capacitors C_B1, C_B2, and C₁ and resistors R₁, R₂, R₁ are small, low-power devices, and the auxiliary inductors L_A1 and L_A2 are very small, as they carry only a small AC current, and virtually no DC current, so the total added volume of the attenuation filtering circuit 40 may be much smaller than either one of the main inductors L_M1, L_M2 (which have to carry the large DC load current and the small AC ripple current) required for the output filter capacitor C_OUT of the output filter circuit 30.

[0035] Proper matching of the auxiliary inductors L_A1, L_A2 to their respective main inductors L_M1, L_M2 depends upon the turns ratio of the coupled inductor assembly.

[0036] The coupled inductor turns ratio N for each of the transformers T₁, T₂ may be calculated by dividing the number of secondary windings by the number of primary windings. The desired value for each of the auxiliary inductors L_A1, L_A2 may be calculated by multiplying the inductance value of their respective main inductor L_M1, L_M2 by the coupled inductor turns ratio N, and then multiplying this value by 1 minus the coupled inductor turns ratio N.

[0037] In one exemplary embodiment where the attenuation filtering circuit provides a 1 MHz output, the ratio of the number of primary windings to the number of secondary windings for each of the transformers T1, T2 is 4:1, the coupled inductor turns ratio N is 0.25, the first main inductor L_M1 is 4 μH, the first blocking capacitor is 0.5 μF, the first resistor is 0.3 ohms, the intermediate capacitor is 0.5 μF, the intermediate resistor is 0.3 ohms, the second main inductor L_M2 is 2 μH, the second blocking capacitor is 0.1 μF, the second resistor is 0.1 ohms, the output filter capacitor C_OUT is 200 μF and the load resistor R_L is 5 ohms. Hence, the first auxiliary inductor L_A1 ideally should be 4×0.25×(1−0.25)=0.75 μH and the second auxiliary inductor L_A2 ideally should be 2×0.25×(1−0.25)=0.375 μH. The component values in other embodiments may be varied from those described above depending upon the desired output frequency and the like.

[0038] U.S. Pat. No. 6,693,805, which is incorporated herein by reference, describes an LC output filter including a prior art ripple cancellation circuit. The LC output filer/ripple cancellation circuit of the U.S. Pat. No. 6,693,805 forms a second order filter. The high attenuation filter circuit described herein increases the order of the LC output filter from a second-order filter, as described in U.S. Pat. No. 6,693,805, to an eighth order filter. In addition, high attenuation filter circuit is much less sensitive to matching of the main and auxiliary inductors than the single stage ripple cancellation circuit described in U.S. Pat. No. 6,693,805.

[0039] FIG. 3 shows two attenuation plots 50 and 52 for the LC output filter circuit described in U.S. Pat. No. 6,693,805, where the auxiliary inductor of the single stage ripple cancellation circuit described therein is substantially matched to the main inductor of the LC output filter circuit, e.g., where the auxiliary inductor was set to 1.125 μH and the main inductor was set to 6 μH. Plot 50 shows the attenuation of the LC output filter circuit when the single stage ripple cancellation circuit was not included and plot 52 shows the attenuation of the LC output filter when the ripple cancellation circuit was included therewith. As can be seen, the attenuation added by this prior art ripple cancellation circuit at 1 MHz was 21 dB and at 10 MHz was 43 dB.

[0040] FIG. 4 shows two attenuation plots 60 and 62 for the LC output filter circuit described in U.S. Pat. No. 6,693,805, where the auxiliary inductor of the single stage ripple cancellation circuit described therein is mismatched to the main inductor of the LC output filter circuit by 10%, e.g., where the auxiliary inductor was set to 1.0125 μH and the main inductor was set to 6 μH. Plot 60 shows the attenuation of the LC output filter circuit when the single stage ripple cancellation circuit was not included and plot 62 shows the attenuation of the LC output filter circuit when the ripple cancellation circuit was included therewith. As can be seen, the attenuation added by the prior art ripple cancellation circuit of U.S. Pat. No. 6,693,805 was limited to 14 dB at 1 MHz, which is about a 7 db reduction in attenuation when compared to matched inductors. The attenuation added by the prior art ripple cancellation circuit of U.S. Pat. No. 6,693,805 was limited to only to 19 dB at 10 MHz, which is about a 24 dB loss in attenuation, when compared to the matched inductors embodiment. This indicates a practical limitation on how much added attenuation can be obtained from the prior art single stage ripple cancellation circuit of U.S. Pat. No. 6,693,805.

[0041] FIG. 5 shows two attenuation plots 70 and 72 for the LC output filter circuit 30 disclosed herein, where the auxiliary inductors L_A1, L_A2 of the two stage attenuation filtering circuit 40 disclosed herein are substantially matched to their corresponding main inductors L_M1, L_M2 of the LC output filter circuit 30, e.g., where the auxiliary inductors L_A1 and L_A2 were set to 0.75 μH and 0.375 μH respectively, and their corresponding main inductors L_M1 and L_M2 were set to 4 μH and 2 μH, respectively. Plot 70 shows the attenuation of the LC output filter 30 circuit when the two stage attenuation filtering circuit 40 (ripple cancellation circuit) was not included and plot 72 shows the attenuation of the LC output filter circuit 30 when the attenuation filtering circuit 40 was included therewith. As can be seen, the attenuation added by the attenuation filtering circuit 40 circuit at 1 MHz was 66 dB and at 10 MHz was 140 dB.

[0042] Comparing the attenuation filtering circuit 40 disclosed herein to the prior art ripple cancellation circuit of U.S. Pat. No. 6,693,805, the attenuation filtering circuit 40 added about 45 dB more attenuation to the LC output filter 30 at 1 MHz than the attenuation added by the ripple cancellation circuit of U.S. Pat. No. 6,693,805, with matched the auxiliary and main inductors. At 10 MHz, the attenuation filtering circuit 40 added almost 100 dB more attenuation to the LC output filter 30 than the attenuation added by the ripple cancellation circuit of U.S. Pat. No. 6,693,805, with matched auxiliary and main inductors.

[0043] FIG. 6 shows two attenuation plots 80 and 82 for the LC output filter circuit 30 disclosed herein, where the auxiliary inductors L_A1, L_A2 of the two stage attenuation filtering circuit 40 disclosed herein are mismatched to their corresponding main inductors L_M1, L_M2 of the LC output filter circuit 30 by 10%, e.g., where the auxiliary inductors L_A1 and L_A2 were set to 0.675 μH and 0.3375 μH respectively, and their corresponding main inductors L_M1 and L_M2 were set to 4 μH and 2 μH, respectively. Plot 80 shows the attenuation of the LC output filter 30 circuit when the two stage attenuation filtering circuit 40 was not included and plot 82 shows the attenuation of the LC output filter circuit 30 when the attenuation filtering circuit 40 was included therewith. As can be seen, the attenuation added by the attenuation filtering circuit 40 circuit at 1 MHz was still 52 dB and at 10 MHz was still 87 dB with mismatched auxiliary inductors.

[0044] Comparing again the attenuation filtering circuit 40 disclosed herein to the prior art ripple cancellation circuit of U.S. Pat. No. 6,693,805 with a perfectly matched auxiliary inductor, the attenuation filtering circuit 40 still added about 31 dB more attenuation to the LC output filter 30 at 1 MHz with mismatched auxiliary and main inductors than the attenuation added by the ripple cancellation circuit of U.S. Pat. No. 6,693,805, added about 44 dB more attenuation to the LC output filter 30 at 10 MHz with mismatched auxiliary and main inductors than the attenuation added by the ripple cancellation circuit of U.S. Pat. No. 6,693,805, with matched the auxiliary and main inductors. Thus, even with mismatched auxiliary inductors, the proposed two stage approach still provides very high attenuation.

[0045] As shown in FIGS. 7 and 8, there is very little additional phase shift associated with the attenuation filtering circuit 40, e.g., 0.7 degrees at 50 kHz with matched and 10% mismatched auxiliary and main inductors, 6.9 degrees at 100 kHz with matched auxiliary and main inductors, and 6.8 degrees at 100 kHz with 10% mismatched auxiliary and main inductors, until a breakpoint is hit. After the breakpoint, a very large phase shift occurs. The attenuation filtering circuit 40 can be used inside a power converter feedback circuit to provide high attenuation, while not affecting the converter closed-loop performance, because there is very little phase shift up to the converter power bandwidth, which would be less than 100 kHz. This allows the attenuation filter circuit 40 to be inside the feedback loop, that is, the output capacitor, Cout, voltage can still be used as the sensed voltage for the converter feedback, with very little added phase shift due to the high attenuation filter circuit.

[0046] The attenuation filtering circuit 40 is much smaller than other ripple cancellation circuits, because only some of the components carry the full load current, and is very easy to damp the using low-power components.

[0047] Although the invention has been described in terms of exemplary embodiments, it is not limited thereto. Rather, the appended claims should be construed broadly, to include other variants and embodiments of the invention, which may be made by those skilled in the art without departing from the scope and range of equivalents of the invention.