Systems and methods for immobilization with repetition rate control

[0001] This application is a continuation of and claims priority to co-pending U.S. patent application Ser. No. 11/285,945, filed Nov. 23, 2005 by Nerheim, which is a continuation of U.S. patent application Ser. No. 10/447,447, filed May 29, 2003 by Nerheim, incorporated herein by reference.

[0002] The present invention relates to electronic disabling devices, and more particularly, to electronic disabling devices with repetition rate control.

[0003] The original stun gun was invented in the 1960's by Jack Cover. Such prior art stun guns incapacitated a target by delivering a sequence of high voltage pulses into the skin of a subject such that the current flow through the subject essentially “short-circuited” the target's neuromuscular system causing a stun effect in lower power systems and involuntary muscle contractions in more powerful systems. Stun guns, or electronic disabling devices, have been made in two primary configurations. A first stun gun design requires the user to establish direct contact between the first and second stun gun output electrodes and the target. A second stun gun design operates on a remote target by launching a pair of darts which typically incorporate barbed pointed ends. The darts either indirectly engage the clothing worn by a target or directly engage the target by causing the barbs to penetrate the target's skin. In most cases, a high impedance air gap exists between one or both of the first and second stun gun electrodes and the skin of the target because one or both of the electrodes contact the target's clothing rather than establishing a direct, low impedance contact point with the target's skin.

[0004] One of the most advanced existing stun guns incorporates the circuit concept illustrated in the FIG. 1 schematic diagram. Closing safety switch S1 connects the battery power supply to a microprocessor circuit and places the stun gun in the “armed” and ready to fire configuration. Subsequent closure of the trigger switch S2 causes the microprocessor to activate the power supply which generates a pulsed voltage output on the order of 2000 volts which is coupled to charge an energy storage capacitor up to the 2000 volt power supply output voltage. Spark gap “GAP1” periodically breaks down, causing a high current pulse through transformer T1 which transforms the 2000 volt input into a 50,000 volt output pulse.

[0005] TASER International of Scottsdale, Ariz., the assignee of the present invention, has for several years manufactured sophisticated stun guns of the type illustrated in the FIG. 1 block diagram designated as the TASER® Model M18 and Model M26 stun guns. High power stun guns such as these TASER International products typically incorporate an energy storage capacitor having a capacitance rating of from 0.2 microfarads at 2000 volts on a light duty weapon up to 0.88 microfarads at 2000 volts as used on the TASER M18 and M26 stun guns.

[0006] After the trigger switch S2 is closed, the high voltage power supply begins charging the energy storage capacitor up to the 2000 volt power supply peak output voltage. When the power supply output voltage reaches the 2000 voltage spark gap breakdown voltage. A spark is generated across the spark gap designated as “GAP1.” Ionization of the spark gap reduces the spark gap impedance from a near infinite impedance to a near zero impedance and allows the energy storage capacitor to almost fully discharge through step up transformer T1. As the output voltage of the energy storage capacitor rapidly decreases from the original 2000 volt level to a much lower level, the current flow through the spark gap decreases toward zero causing the spark gap to deionize and to resume its open circuit configuration with a near infinite impedance. This “reopening” of the spark gap defines the end of the first 50,000 volt output pulse which is applied to output electrodes designated in FIG. 1 as “E1” and “E2.” A typical stun gun of the type illustrated in the FIG. 1 circuit diagram produces from 5 to 20 pulses per second.

[0007] Because a stun gun designer must assume that a target may be wearing an item of clothing such as a leather or cloth jacket which functions to establish a ¼ inch to 1 inch air gap between stun gun electrodes E1 and E2 and the target's skin, stun guns have been required to generate 50,000 volt output pulses because this extreme voltage level is capable of establishing an arc across the high impedance air gap which may be presented between the stun gun output electrodes E1 and E2 and the target's skin. As soon as this electrical arc has been established, the near infinite impedance across the air gap is promptly reduced to a very low impedance which allows current to flow between the spaced apart stun gun output electrodes E1 and E2 and through the target's skin and intervening tissue regions. By generating a significant current flow within the target across the spaced apart stun gun output electrodes, the stun gun essentially short circuits the target's electromuscular control system and induces severe muscular contractions. With high power stun guns, such as the TASER M18 and M26 stun guns, the magnitude of the current flow across the spaced apart stun gun output electrodes causes numerous groups of skeletal muscles to rigidly contract. By causing high force level skeletal muscle contractions, the stun gun causes the target to lose its ability to maintain an erect, balanced posture. As a result, the target falls to the ground and is incapacitated.

[0008] The “M26” designation of the TASER stun gun reflects the fact that, when operated, the TASER M26 stun gun delivers 26 watts of output power as measured at the output capacitor. Due to the high voltage power supply inefficiencies, the battery input power is around 35 watts at a pulse rate of 15 pulses per second. Due to the requirement to generate a high voltage, high power output signal, the TASER M26 stun gun requires a relatively large and relatively heavy 8 AA cell battery pack. In addition, the M26 power generating solid state components, its energy storage capacitor, step up transformer and related parts must function either in a high current relatively high voltage mode (2000 volts) or be able to withstand repeated exposure to 50,000 volt output pulses.

[0009] At somewhere around 50,000 volts, the M26 stun gun air gap between output electrodes E1 and E2 breaks down, the air is ionized, a blue electric arc forms between the electrodes and current begins flowing between electrodes E1 and E2. As soon as stun gun output terminals E1 and E2 are presented with a relatively low impedance load instead of the high impedance air gap, the stun gun output voltage will drop to a significantly lower voltage level. For example, with a human target and with about a 10-inch probe to probe separation, the output voltage of a TASER Model M26 might drop from an initial high level of 50,000 volts to a voltage on the order of about 5,000 volts. This rapid voltage drop phenomenon with even the most advanced conventional stun guns results because such stun guns are tuned to operate in only a single mode to consistently create an electrical arc across a very high, near infinite impedance air gap. Once the stun gun output electrodes actually form a direct low impedance circuit across the spark gap, the effective stun gun load impedance decreases to the target impedance-typically on the order of 1000 ohms or less. A typical human subject frequently presents a load impedance on the order of about 200 ohms.

[0010] Conventional stun guns have by necessity been designed to have the capability of causing voltage breakdown across a very high impedance air gap. As a result, such stun guns have been designed to produce a 50,000 to 60,000 volt output. Once the air gap has been ionized and the air gap impedance has been reduced to a very low level, the stun gun, which has by necessity been designed to have the capability of ionizing an air gap, must now continue operating in the same mode while delivering current flow or charge across the skin of a now very low impedance target. The resulting high power, high voltage stun gun circuit operates relatively inefficiently yielding low electro-muscular efficiency and with high battery power requirements.

[0048] In order to better illustrate the advantages of the invention and its contributions to the art, a preferred embodiment of the invention will now be described in detail.

[0049] Referring now to FIG. 2, an electronic disabling device for immobilizing a target according to the present invention includes a power supply, first and second energy storage capacitors, and switches S1 and S2 which operate as single pole, single throw switches and serve to selectively connect the two energy storage capacitors to down stream circuit elements. The first energy storage capacitor is selectively connected by switch S1 to a voltage multiplier which is coupled to first and second stun gun output electrodes designated E1 and E2. The first leads of the first and second energy storage capacitors are connected in parallel with the power supply output. The second leads of each capacitor are connected to ground to thereby establish an electrical connection with the grounded output electrode E2.

[0050] The stun gun trigger controls a switch controller which controls the timing and closure of switches S1 and S2.

[0051] Referring now to FIGS. 3-8 and FIG. 12, the power supply is activated at time T0. The energy storage capacitor charging takes place during time interval T0-T1 as illustrated in FIGS. 12A and 12B.

[0052] At time T1, switch controller closes switch S1 which couples the output of the first energy storage capacitor to the voltage multiplier. The FIG. 3B and FIG. 6 voltage versus time graphs illustrate that the voltage multiplier output rapidly builds from a zero voltage level to a level indicated in the FIG. 3B and FIG. 6 graphics as “V_HIGH”.

[0053] In the hypothetical situation illustrated in FIG. 5A, a high impedance air gap exists between stun gun output electrode E1 and target contact point E3. The FIG. 5A diagram illustrates the hypothetical situation where a direct contact (i.e., impedance E2-E4 equals zero) has been established between stun gun electrical output terminal E2 and the second spaced apart contact point E4 on a human target. The E1 to E2 on the target spacing is assumed to equal on the order of 10 inches. The resistor symbol and the symbol Z_LOAD represents the internal target resistance which is typically less than 1000 ohms and approximates 200 ohms for a typical human target.

[0054] Application of the V_HIGH voltage multiplied output across the E1 to E3 high impedance air gap forms an electrical arc having ionized air within the air gap. The FIG. 5C timing diagram illustrates that after a predetermined time during the T1 to T2 high voltage waveform output interval, the air gap impedance drops from a near infinite level to a near zero level. This second air gap configuration is illustrated in the FIG. 5B drawing.

[0055] Once this low impedance ionized path has been established by the short duration application of the V_HIGH output signal which resulted from the discharge of the first energy storage capacitor through the voltage multiplier, the switch controller opens switch S1 and closes switch S2 to directly connect the second energy storage capacitor across the electronic disabling device output electrodes E1 and E2. The circuit configuration for this second time interval is illustrated in the FIG. 4A block diagram. As illustrated in the FIG. 4B voltage waveform output diagram, the relatively low voltage V_LOW derived from the second output capacitor is now directly connected across the stun gun output terminals E1 and E2. Because the ionization of the air gap during time interval T1 to T2 dropped the air gap impedance to a low level, application of the relatively low second capacitor voltage “V_LOW” across the E1 to E3 air gap during time interval T2 to T3 will allow the second energy storage capacitor to continue and maintain the previously initiated discharge across the arced-over air gap for a significant additional time interval. This continuing, lower voltage discharge of the second capacitor during the interval T2 to T3 transfers a substantial amount of target-incapacitating electrical charge through the target.

[0056] As illustrated in FIGS. 4B, 5C, 6 and 8, the continuing discharge of the second capacitor through the target will exhaust the charge stored in the capacitor and will ultimately cause the output voltage from the second capacitor to drop to a voltage level at which the ionization within the air gap will revert to the non-ionized, high impedance state causing cessation of current flow through the target.

[0057] In the FIG. 2 block diagram, the switch controller can be programmed to close switch S1 for a predetermined period of time and then to close switch S2 for a predetermined period of time to control the T1 to T2 first capacitor discharge interval and the T2 to T3 second capacitor discharge interval.

[0058] During the T3 to T4 interval, the power supply will be disabled to maintain a factory present pulse repetition rate. As illustrated in the FIG. 8 timing diagram, this factory present pulse repetition rate defines the overall T0 to T4 time interval. A timing control circuit potentially implemented by a microprocessor maintains switches S1 and S2 in the open condition during the T3 to T4 time interval and disables the power supply until the desired T0 to T4 time interval has been completed. At time T0, the power supply will be reactivated to recharge the first and second capacitors to the power supply output voltage.

[0059] Referring now to the FIG. 9 schematic diagram, the FIG. 2 circuit has been modified to include a third capacitor and a load diode (or resistor) connected as shown. The operation of this enhanced circuit diagram will be explained below in connection with FIG. 10 and the related more detailed schematic diagrams.

[0060] Referring now to the FIG. 10 electrical schematic diagram, the high voltage power supply generates an output current I1 which charges capacitors C1 and C3 in parallel. While the second terminal of capacitor C2 is connected to ground, the second terminal of capacitor C3 is connected to ground through a relatively low resistance load resistor R1 or as illustrated in FIG. 9 by a diode. The first voltage output of the high voltage power supply is also connected to a 2000 volt spark gap designated as “GAP1” and to the primary winding of an output transformer having a 1:25 primary to secondary winding step up ratio.

[0061] The second equal voltage output of the high voltage power supply is connected to one terminal of capacitor C2 while the second capacitor terminal is connected to ground. The second power supply output terminal is also connected to a 3000 volt spark gap designated GAP2. The second side of spark gap GAP2 is connected in series with the secondary winding of transformer T1 and to stun gun output terminal E1.

[0062] In the FIG. 10 circuit, closure of safety switch S1 enables operation of the high voltage power supply and places the stun gun into a standby/ready to operate configuration. Closure of the trigger switch designated S2 causes the microprocessor to send a control signal to the high voltage power supply which activates the high voltage power supply and causes it to initiate current flow I1 into capacitors C1 and C3 and current flow 12 into capacitor C2. This capacitor charging time interval will now be explained in connection with the simplified FIG. 11 block diagram and in connection with the FIG. 12A and FIG. 12B voltage versus time graphs.

[0063] During the T0 to T1 capacitor charging interval illustrated in FIGS. 11 and 12, capacitors C1, C2 and C3 begin charging from a zero voltage up to the 2000 volt output generated by the high voltage power supply. Spark gaps GAP1 and GAP2 remain in the open, near infinite impedance configuration because only at the end of the T0 to T1 capacitor charging interval will the C1/C2 capacitor output voltage approach the 2000 volt breakdown rating of GAP1.

[0064] Referring now to FIGS. 13 and 14, as the voltage on capacitors C1 and C2 reaches the 2000 volt breakdown voltage of spark gap GAP1, a spark will be formed across the spark gap and the spark gap impedance will drop to a near zero level. This transition is indicated in the FIG. 14 timing diagrams as well as in the more simplified FIG. 3B and FIG. 6 timing diagrams. Beginning at time T1, capacitor C1 will begin discharging through the primary winding of transformer T1 which will rapidly ramp up the E1 to E2 secondary winding output voltage to negative 50,000 volts as shown in FIG. 14B. FIG. 14A illustrates that the voltage across capacitor C1 relatively slowly decreases from the original 2000 volt level while the FIG. 14B timing diagram illustrates that the multiplied voltage on the secondary winding of transformer T1 will rapidly build up during the time interval T1 to T2 to a voltage approaching minus 50,000 volts.

[0065] At the end of the T2 time interval, the FIG. 10 circuit transitions into the second configuration where the 3000 volt GAP2 spark gap has been ionized into a near zero impedance allowing capacitors C2 and C3 to discharge across stun gun output terminals E1 and E2 through the relatively low impedance load target. Because as illustrated in the FIG. 16 timing diagram, the voltage across C1 will have discharged to a near zero level as time approaches T2, the FIG. 15 simplification of the FIG. 10 circuit diagram which illustrates the circuit configuration during the T2 to T3 time interval shows that capacitor C1 has effectively and functionally been taken out of the circuit. As illustrated by the FIG. 16 timing diagram, during the T2 to T3 time interval, the voltage across capacitors C2 and C3 decreases to zero as these capacitors discharge through the now low impedance (target only) load seen across output terminals E1 and E2.

[0066] FIG. 17 represents another timing diagram illustrating the voltage across GAP2 and the voltage across stun gun output terminals E1 and E2 during the T2 to T3 time interval.

[0067] In one preferred embodiment of the FIG. 10 circuit, capacitor C1, the discharge of which provides the relatively high energy level required to ionize the high impedance air gap between E1 and E3, can be implemented with a capacitor rating of 0.14 microfarad and 2000 volts. As previously discussed, capacitor C1 operates only during time interval T1 to T2 which, in this preferred embodiment, approximates on the order of 1.5 microseconds in duration. Capacitors C2 and C3 in one preferred embodiment may be selected as 0.02 microfarad capacitors for a 2000 volt power supply voltage and operate during the T2 to T3 time interval to generate the relatively low voltage output as illustrated in FIG. 4B to maintain the current flow through the now low impedance dart-to-target air gap during the T2 to T3 time interval as illustrated in FIG. 5C. In this particular preferred embodiment, the duration of the T2 to T3 time interval approximates 50 microseconds.

[0068] The duration of the T1 to T2 time interval can be varied from 1.5 to 0.5 microseconds. The duration of the T2 to T3 time interval can be varied from 20 to 200 microseconds. Due to many variables, the duration of the T0 to T1 time interval charge. For example, a fresh battery may shorten the T0 to T1 time interval in comparison to circuit operation with a partially discharged battery. Similarly, operation of the stun gun in cold weather which degrades battery capacity might also increase the T0 to T1 time interval.

[0069] Since it is highly desirable to operate stun guns with a fixed pulse repetition rate as illustrated in the FIG. 8 timing diagram, the circuit of the present invention provides a microprocessor-implemented digital pulse control interval designated as the T3 to T4 interval in FIG. 8. As illustrated in the FIG. 10 block diagram, the microprocessor receives a feedback signal from the high voltage power supply via a feedback signal conditioning element which provides a circuit operating status signal to the microprocessor. The microprocessor is thus able to detect when time T3 has been reached as illustrated in the FIG. 6 timing diagram and in the FIG. 8 timing diagram. Since the commencement time T0 of the operating cycle is known, the microprocessor will maintain the high voltage power supply in a shut down or disabled operating mode from T3 until the factory preset pulse repetition rate defined by the T0 to T4 time interval has been achieved. While the duration of the T3 to T4 time interval will vary, the microprocessor will maintain the T0 to T4 time interval constant.

[0070] The FIG. 18 table entitled “Gap On/Off Timing” represents a simplified summary of the configuration of GAP1 and GAP2 during the four relevant operating time intervals. The configuration “off” represents the high impedance, non-ionized spark gap state while the configuration “on” represents the ionized state where the spark gap breakdown voltage has been reached.

[0071] FIG. 19 represents a simplified block diagram of a circuit analogous to the FIG. 10 circuit except that the circuit has been simplified to include only capacitors C1 and C2. The FIG. 19 circuit is capable of operating in a highly efficient or “tuned” dual mode configuration according to the teachings of the present invention.

[0072] FIG. 20 illustrates an alternative configuration for coupling capacitors C1 and C2 to the stun gun output electrodes E1 and E2 via an output transformer having a single primary winding and a center-tapped or two separate secondary windings. The step up ratio relative to each primary winding and each secondary winding represents a ratio of 1:12.5. This modified output transformer still accomplishes the objective of achieving a 1:25 step-up ratio for generating an approximate 50,000 volt signal with a 2000 volt power supply rating. One advantage of this double secondary transformer configuration is that the maximum voltage applied to each secondary winding is reduced by 50%. Such reduced secondary winding operating potentials may be desired in certain conditions to achieve a higher output voltage with a given amount of transformer insulation or for placing less high voltage stress on the elements of the output transformer.

[0073] Substantial and impressive benefits may be achieved by using the electronic disabling device of the present invention which provides for dual mode operation to generate a time-sequenced, shaped voltage output waveform in comparison to the most advanced prior art stun gun represented by the TASER M26 stun gun as illustrated and described in connection with the FIG. 1 block diagram.

[0074] The TASER M26 stun gun utilizes a single energy storage capacitor having a 0.88 microfarad capacitance rating. When charged to 2000 volts, that 0.88 microfarad energy storage capacitor stores and subsequently discharges 1.76 joules of energy during each output pulse. For a standard pulse repetition rate of 15 pulses per second with an output of 1.76 joules per discharge pulse, the TASER M26 stun gun requires around 35 watts of input power which, as explained above, must be provided by a large, relatively heavy battery power supply utilizing 8 series-connected AA alkaline battery cells.

[0075] For one embodiment of the electronic disabling device of the present invention which generates a time-sequenced, shaped voltage output waveform and with a C1 capacitor having a rating of 0.07 microfarads and a single capacitor C2 with a capacitance of 0.01 microfarads (for a combined rating of 0.08 microfarads), each pulse repetition consumes only 0.16 joules of energy. With a pulse repetition rate of 15 pulses per second, the two capacitors consume battery power of only 2.4 watts at the capacitors (roughly 3.5 to 4 watts at the battery), a 90% reduction, compared to the 26 watts consumed by the state of the art TASER M26 stun gun. As a result, this particular configuration of the electronic disabling device of the present invention which generates a time-sequenced, shaped voltage output waveform can readily operate with only a single AA battery due to its 2.4 watt power consumption.

[0076] Because the electronic disabling device of the present invention generates a time-sequenced, shaped voltage output waveform as illustrated in the FIG. 3B and FIG. 4B timing diagrams, the output waveform of this invention is tuned to most efficiently accommodate the two different load configurations presented: a high voltage output operating mode during the high impedance T1 to T2 first operating interval and, a relatively low voltage output operating mode during the low impedance second T2 to T3 operating interval.

[0077] As illustrated in the FIG. 5C timing diagram and in the FIGS. 2, 3A and 4A simplified schematic diagrams, the circuit of the present invention is selectively configured into a first operating configuration during the T2 to T1 time interval where a first capacitor operates in conjunction with a voltage multiplier to generate a very high voltage output signal sufficient to breakdown the high impedance target-related air gap as illustrated in FIG. 5A. Once that air gap has been transformed into a low impedance configuration as illustrated in the FIG. 5C timing diagram, the circuit is selectively reconfigures into the FIG. 3A second configuration where a second or a second and a third capacitor discharge a substantial amount of current through the now low impedance target load (typically 1000 ohms or less) to thereby transfer a substantial amount of electrical charge through the target to cause massive disruption of the target's neurological control system to maximize target incapacitation.

[0078] Accordingly, the electronic disabling device of the present invention which generates a time-sequenced, shaped voltage output waveform is automatically tuned to operate in a first circuit configuration during a first time interval to generate an optimized waveform for attacking and eliminating the otherwise blocking high impedance air gap and is then returned to subsequently operate in a second circuit configuration to operate during a second time interval at a second much lower optimized voltage level to efficiently maximize the incapacitation effect on the target's skeletal muscles. As a result, the target incapacitation capacity of the present invention is maximized while the stun gun power consumption is minimized.

[0079] As an additional benefit, the circuit elements operate at lower power levels and lower stress levels resulting in either more reliable circuit operation and can be packaged in a much more physically compact design. In a laboratory prototype embodiment of a stun gun incorporating the present invention, the prototype size in comparison to the size of present state of the art TASER M26 stun gun has been reduced by approximately 50% and the weight has been reduced by approximately 60%.

[0080] An enhanced stun gun one embodiment of which is currently designated as the TASER® X26 system includes a novel battery capacity readout system designed to create a device that is more reliable and dependable in the field. With previous battery operated stun guns, users have experienced major difficulty in determining exactly how much battery capacity remains in the batteries.

[0081] In most electronic devices the remaining battery capacity can be predicted either by measuring the battery voltage during operation or integrating the battery discharge current over time. Because the X26 system draws current at very different rates depending on the mode in which it operates, prior art battery management methods yield unreliable results. Because the X26 system is expected to function over a wide operating temperature range, non-temperature compensated prior art battery capacity prediction methods produce even less reliable results.

[0082] The battery consumption of the X26 system varies with its operating mode as described in Table 1.

[0083]

TABLE 1
Operating
Mode	Battery Consumption
1	The X26 system includes a real time clock
	which draws around 3.5 microamps.
2	If the system safety switch is armed, the
	now-activated microprocessor and its clock
	system draw around 4 milliamps.
3	If enabled, and if the safety switch is armed,
	the X26 system laser target designator will
	draw around 11 milliamps.
4	If enabled, and if the safety switch is armed,
	the forward facing low intensity twin white
	LED flashlight will draw around 63
	milliamps.
5	If the safety switch is armed and the trigger
	is pulled, the X26 system will draw about 3
	to 4 amps.

[0084] As evident from the above examples, the minimum to maximum current drain will vary in a ratio of 1,000,000:1.

[0085] To further complicate matters, the capacity of the CR123 lithium batteries packaged in the system battery model varies greatly over the operating temperature range of the X26 system. At −20° C., the X26 dual in-series CR123 battery module can deliver around 100 of the 5-second discharge cycles. At +30° C., the X26 system battery module can deliver around 350 of the 5-second discharge cycles.

[0086] From the warmest to the coldest operating temperature range and from the lowest to the highest battery drain functions, a battery life ratio of around 5,000,000:1 results. Since the wide range in battery drain makes prior art battery prediction methods unreliable, a new battery capacity assessment system was required for the X26 system. The new battery capacity assessment system predicts the remaining battery capacity based on actual laboratory measurements of critical battery parameters under different load and at different temperature conditions. These measured battery capacity parameters are stored electronically as a table (FIG. 27) in an electronic non-volatile memory device included with each battery module. (FIG. 22) As illustrated in FIGS. 21 and 22 and in FIGS. 31 and 32, appropriate data interface contacts enable the X26 microprocessor to communicate with the table electronically stored in the battery module to predict remaining battery capacity. The X26 system battery module with internal electronic non-volatile memory may be referred to as the Digital Power Magazine (DPM) or simply as the system battery module.

[0087] The data required to construct the data tables for the battery module were collected by operating the various X26 system features at selected temperatures spanning the X26 system operating temperature range while recording the battery performance and longevity at each temperature interval.

[0088] The resulting battery capacity measurements were collected and organized into a tabular spreadsheet of the type illustrated in FIG. 27. The battery drain parameters for each system feature were calculated and translated into standardized drain values in microamp-hours based on the sensible operating condition of that feature. For example, the battery drain required to keep the clock alive is represented by a number in microamp-hours that totals the current required to keep the clock alive for 24 hours. The battery drain to power up the microprocessor, the forward directed flashlight, and the laser target designator for 1 second are represented by separate table entries with values in microamp-hours. The battery drain required to operate the gun in the firing mode is represented by numbers in microamp-hours of battery drain required to fire a single power output pulse.

[0089] To enable the X26 system to be operated at all various temperatures, while keeping track of battery drain and remaining battery capacity, the total available battery capacity at each incremental temperature was measured. The battery capacity in microamp-hours at 25° C. (ambient) was programmed into the table to represent a normalized 100% battery capacity value. The battery table drain numbers at other temperatures were adjusted to coordinate with the 25° C. total (100%) battery capacity number. For example, since the total battery capacity at −20° C. was measured to approximate 35% of the battery capacity at 25° C., the microamp-hours numbers at −20° C. were multiplied by 1/0.35

[0090] A separate location in the FIG. 27 table is used by the X26 system microprocessor to keep track of used battery capacity. This number is updated every 1 second if the safety selector remains in the “armed” position, and every 24 hours if the safety selector remains in the “safe” position. Remaining battery capacity percentage is calculated by dividing this number by the total battery capacity. The X26 system will display this percentage of battery capacity remaining on the 2-digit Central Information Display (CID) 14 shown in FIG. 33 for 2 seconds each time the weapon is armed. See, for example, the 98% battery capacity read-out depicted in the FIG. 33 X26 system rear view.

[0091] FIG. 22 illustrates the electronic circuit located inside the X26 battery module 12. As illustrated in the FIG. 22 schematic diagram and in the FIG. 30 view of X26 system 10, the removable battery module 12 consists of two series-connected, 3-volt CR123 lithium batteries and a nonvolatile memory device. The nonvolatile memory device may take the form of a 24AA128 flash memory which contains 128 K bits of data storage. As shown in FIGS. 21 and 22, the electrical and data interface between the X26 system microprocessor and battery module 12 is established by a 6-pin jack JP1 and provides a 2-line I²C serial bus for data transmission purposes.

[0092] While the battery capacity monitoring apparatus and methodology has been described in connection with monitoring the remaining capacity of a battery energized power supply for a stun gun, this inventive feature could readily be applied to any battery powered electronic device which includes a microprocessor, such as cell phones, video camcorders, laptop computers, digital cameras, and PDA's. Each of these categories of electronic devices frequently shift among various different operating modes where each operating mode consumes a different level of battery power. For example, for a cell phone, the system selectively operates in the different power consumption modes described in Table 2.

[0093]

TABLE 2
Operating
Mode	Battery Consumption
1	power off/microprocessor clock on
2	power on standby/receive mode
3	receiving an incoming telephone call and
	amplifying the received audio input signal
4	transmit mode generating an RF power
	output of about 600 milliwatts
5	ring signal activated in response to an
	incoming call
6	backlight “on”

[0094] To implement the present invention in a cell phone embodiment, a battery module analogous to that illustrated in the FIG. 22 electrical schematic diagram would be provided. That module would include a memory storage device such as the element designated by reference number U1 in the FIG. 22 schematic diagram to receive and store a battery consumption table as illustrated in FIG. 27. The cell phone microprocessor can then be programmed to read out and display either at power up or in response to a user-selectable request the battery capacity remaining within the battery module or the percentage of used capacity.

[0095] Similar analysis and benefits apply to the application of the battery capacity monitor of the present invention to other applications such as a laptop computer which selectively switches between the different battery power consumption modes described in Table 3.

[0096]

TABLE 3
Operating
Mode	Battery Consumption
1	CPU “on,” but operating in a standby power
	conservation mode
2	CPU operating in a normal mode with the
	hard drive in the “on” configuration
3	CPU operating in a normal mode with the
	hard drive in the “off” configuration
4	CPU “on” and LCD screen also in the “on”
	fully illuminated mode
5	CPU operating normally with the LCD
	screen switched into the “off” power
	conservation configuration
6	modem on/modem off modes
7	optical drives such as DVD or CD ROM
	drives operating in the playback mode
8	optical drives such as DVD or CD ROM
	drives operating in the record or write mode
9	laptop audio system generating an audible
	output as opposed to operating without an
	audio output signal

[0097] In each of the cases addressed above, the battery capacity table would be calibrated for each different power consumption mode based on the power consumption of each individual operating element. Battery capacity would also be quantified for a specified number of different ambient temperature operating ranges.

[0098] Tracking the time remaining on the manufacturer's warranty as well as updating and extending the expiration date represents a capability which can also be implemented by the present invention.

[0099] An X26 system embodiment of the present invention is shipped from the factory with an internal battery module 12 (DPM) having sufficient battery capacity to energize the internal clock for much longer than 10 years. The internal clock is set at the factory to the GMT time zone. The internal X26 system electronic warranty tracker begins to count down the factory preset warranty period or duration beginning with the first trigger pull occurring 24 hours or more after the X26 system has been packaged for shipment by the factory.

[0100] Whenever the battery module 12 is removed from the X26 system and replaced 1 or more seconds later, the X26 system will implement an initialization procedure. During that procedure, the 2-digit LED Central Information Display (CID) designated by reference number 14 in FIG. 33, will sequentially read out a series of 2-digit numbers which represent the data described in Table 4.

[0101]

TABLE 4
Series
Position	Data
1, 2, 3	The first 3 sets of 2-digit numbers represent
	the warranty expiration date. The format is
	YY/MM/DD.
4, 5, 6	The current time is displayed: YY/MM/DD.
7	The internal temperature in degrees
	Centigrade is displayed: XX (negative
	numbers are represented by blinking the
	number).
8	The software revision is displayed: XX.

[0102] The system warranty can be extended by different techniques including by Internet and by extended warranty battery module. For extending by Internet, the X26 system includes a USB data interface module accessory which is physically compatible with the shape of the X26 system receptacle for battery module 12. The USB data module can be inserted within the X26 system battery module receptacle and includes a set of electrical contacts compatible with jack JP1 located inside the X26 system battery module housing as illustrated in FIG. 32. The USB interface module may be electrically connected to a computer USB port which supplies power via jack JP1 to the X26 system. While the USB interface is normally used to download firing data from the X26 system, it can also be used to extend the warranty period or to download new software into the X26 microprocessor system. To update the warranty, the user removes the X26 battery module 12, inserts the USB module, connects a USB cable to an Internet enabled computer, goes to the www.taser.com website, follows the download X26 system warranty extension instructions, and pays for the desired extended warranty period by credit card.

[0103] For extending by Extended Warranty Battery Module, the system warranty can also be extended by purchasing from the factory a specially programmed battery module 12 having the software and data required to reprogram the warranty expiration data stored in the X26 microprocessor. The warranty extension battery module is inserted into the X26 system battery receptacle. If the X26 system warranty period has not yet expired, the data transferred to the X26 microprocessor will extend the current warranty expiration date by the period pre-programmed into the extended warranty battery module. Once the extended warranty expiration date has been stored within the X26 system, the microprocessor will initiate a battery insertion initialization sequence and will then display the new warranty expiration date. Various different warranty extension modules can be provided to either extend the warranty of only a single X26 system or to provide warranty extensions for multiple system as might be required to extend the warranty for X26 systems used by an entire police department. If the warranty extension module contains only one warranty extension, the X26 microprocessor will reset the warranty update data in the module to zero. The module can function either before or after the warranty extension operation as a standard battery module. An X26 system may be programmed to accept one warranty extension, for example a 1-year extension, each time that the warranty extension module is inserted into the weapon.

[0104] The warranty configuration/warranty extension feature of the present invention could also readily be adapted for use with any microprocessor-based electronic device or system having a removable battery. For example, as applied to a cell phone having a removable battery module, a circuit similar to that illustrated in the FIG. 22 electrical schematic diagram could be provided in the cell phone battery module to interface with the cellular phone microprocessor system. As was the case with the X26 system of the present invention, the cell phone would be originally programmed at the factory to reflect a device warranty of predetermined duration at the initial time that the cell phone was powered up by the ultimate user/customer. By purchasing a specially configured cell phone replacement battery including data suitable for reprogramming the warranty expiration date within the cell phone microprocessor, a customer could readily replace the cell phone battery while simultaneously updating the system warranty.

[0105] Alternatively, a purchaser of an electronic device incorporating the warranty extension feature of the present invention could return to a retail outlet, such as Best Buy or Circuit City, purchase a warranty extension and have the on-board system warranty extended by a representative at that retail vendor. This warranty extension could be implemented by temporarily inserting a master battery module incorporating a specified number of warranty extensions purchased by the retail vendor from the OEM manufacturer. Alternatively, the retail vendor could attach a USB interface module to the customer's cell phone and either provide a warranty extension directly from the vendor's computer system or by means of data supplied by the OEM manufacturer's website.

[0106] For electronic devices utilizing rechargeable battery power supplies such as is the case with cell phones and video camcorders, battery depletion occurs less frequently than with the system described above which typically utilizes non-rechargeable battery modules. For such rechargeable battery applications, the end user/customer could purchase a replacement rechargeable battery module including warranty update data and could simultaneously trade in the customer's original rechargeable battery.

[0107] For an even broader application of the warranty extension feature of the present invention, that feature could be provided to extend the warranty of other devices such as desktop computer systems, computer monitors or even an automobile. For such applications, either the OEM manufacturer or a retail vendor could supply to the customer's desktop computer, monitor or automobile with appropriate warranty extension data in exchange for an appropriate fee. Such data could be provided to the warranted product via direct interface with the customer's product by means of an infrared data communication port, by a hard-wired USB data link, by an IEEE 1394 data interface port, by a wireless protocol such as Bluetooth or by any other means of exchanging warranty extension data between a product and a source of warranty extension data.

[0108] Another benefit of providing an “intelligent” battery module is that the X26 system can be supplied with firmware updates by the battery module. When a battery module with new firmware is inserted into the X26 system, the X26 system microcontroller will read several identification bytes of data from the battery module. After reading the software configuration and hardware compatibility table bytes of the new program stored in the nonvolatile memory within the battery module to evaluate hardware/software compatibility and software version number, a system software update will take place when appropriate. The system firmware update process is implemented by having the microprocessor (see FIG. 21) in the X26 system read the bytes in the battery module memory program section and programming the appropriate software into the X26 system nonvolatile program memory.

[0109] The X26 system can also receive program updates through a USB interface module by connecting the USB module to a computer to download the new program to a nonvolatile memory provided within the USB module. The USB module is next inserted into the X26 system battery receptacle. The X26 system will recognize the USB module as providing a USB reprogramming function and will implement the same sequence as described above in connection with X26 system reprogramming via battery module.

[0110] The High Voltage Assembly (HVA) schematically illustrated in FIGS. 23 and 24 converts a 3 to 6 Volt battery level to powerful 50 KV pulses having the capability of instantly incapacitating a subject. To provide maximum safety, to avoid false triggering, and to minimize the risk that the X26 system could activate or stay activated if the microprocessor malfunctions or locks up, the ENABLE signal from the microprocessor (FIG. 22) to the HVA (FIGS. 23, 24) has been specially encoded.

[0111] To enable the HVA, the microprocessor must output a 500 Hz square wave with an amplitude of 2.5 to 6 volts and around a 50% duty cycle. The D6 series diode within the HVA power supply “rectifies” the ENABLE signal and uses it to charge up capacitor C6. The voltage across capacitor C6 is used to run pulse width modulation (PWM) controller U1 in the HVA.

[0112] If the ENABLE signal goes low for more than around 1 millisecond, several functions operate to turn the PWM controller off. First, the voltage across capacitor C6 will drop to a level where the PWM can no longer run causing the HVA to turn off. Second, the input to the U1 “RUN” pin must be above a threshold level. The voltage level at that point represents a time average of the ENABLE waveform (due to R1 and C7). If the ENABLE signal goes low, capacitor C7 will discharge and disable the controller after just over 1 millisecond.

[0113] As the ENABLE signal goes high, resistor R3 charges capacitor C8. If the charge level on C8 goes above 1.23 Volts, the PWM will shut down—stopping delivery of 50 KV output pulses. Every time the ENABLE signal goes low, capacitor C8 is discharged, making sure the PWM can stay “on” as the ENABLE signal goes back high and starts charging C8 again. Any time the ENABLE signal remains high for more than 1 millisecond, the PWM controller will be shut down.

[0114] The encoded ENABLE signal requirements dictate that the ENABLE signal must be pulsed at a frequency of around 500 Hz (1 millisecond high, 1 millisecond low) to activate the HVA. If the ENABLE signal sticks at a high or low level, the PWM controller will shut down, stopping the delivery of the 50 KV output pulses.

[0115] The configuration of the X26 system high voltage output circuit represents a key distinction between the X26 system and conventional prior art stun guns. Referring now to FIGS. 23 and 24, the structure and function of the X26 system high voltage “shaped pulse” assembly will be explained. The switch mode power supply will charge up capacitors C1, C2, and C3 through diodes D1, D2, and D3. Note that diodes D1 and D2 can be connected to the same or to different windings of T1 to modify the output waveform. The ratios of the T1 primary and secondary windings and the spark gap voltages on GAP1, GAP2, and GAP3 are configured so that GAP1 will always breakover and fire first. When GAP1 fires, 2 KV is applied across the primary windings of spark coil transformer T2 from pin 6 to pin 5. The secondary voltage on spark coil transformer T2 from pins 1 to 2 and from pins 3 to 4 will approximate 25 KV, depending on the air gap spacing between the two output electrodes E1 and E2. The smaller the air gap, the smaller the output voltage before the air gap across output terminals E1 to E2 breaks down, effectively clamping the output voltage level.

[0116] The voltage induced in the secondary current path by the discharge of Cl through GAP1 and T2 sets up a voltage across C2, GAP2, E1 to E2, GAP3, C3 and C1. When the cumulative voltage across the air gaps (GAP2, E1 to E2, and GAP3) is high enough to cause them to break down, current will start flowing in the circuit, from C2 through GAP2, through the output electrodes E1 to E2, through GAP3, and through C3 in series with C1 back to ground. As long as C1 is driving the output current through GAP1 and T2, the output cuffent as described will remain negative in polarity. As a result, the charge level stored in both C2 and C3 will increase. Once C1 has become somewhat discharged, transformer T2 will not be able to maintain the output voltage across the output windings. At that time, the output cuffent will reverse and begin flowing in a positive direction and will begin depleting the charge on C2 and C3. The discharge of C1 is known as the “arc” phase. The discharge of C2 and C3 is known as the muscle “stimulation” phase.

[0117] Since the high voltage output coil T2 as illustrated in FIG. 24 consists of two separate secondary windings that create a negative polarity spark voltage on E1 followed by a positive polarity spark voltage on E2, the peak voltage measured from either electrode E1 or E2 to primary weapon ground will not exceed 25 KV, yet the peak voltage measured across power supply output terminals E1 and E2 will reach 50 KV. If the output coil T2 had utilized only a single secondary winding as is the case with all prior art stun guns and in other embodiments of the present invention, the maximum voltage from one output electrode (E1 or E2) referenced to primary weapon ground would reach 50 KV. Since a 25 KV output can establish an arc across a gap less than half the size of a gap that can establish an arc with a 50 KV output, reducing the peak output terminal to ground voltage by 50% from 50 KV to 25 KV reduces by more than a 2:1 ratio the risk that the user of this version of the X26 system will be shocked by the high voltage output pulses. This represents a significant safety enhancement for a handheld stun gun weapon.

[0118] Referring now to the FIGS. 23 and 24 schematic diagrams, a feedback signal from the primary side of the HVA (T1 pin 8) provides a mechanism for the FIG. 21 microprocessor to indirectly determine the voltage on capacitor C1, and hence where the X26 system power supply is operating within its pulse firing sequence. This feedback signal is used by the microprocessor to control the output pulse repetition rate.

[0119] The system pulse rate can be controlled to create either a constant or a time-varying pulse rate by having the microcontroller stop toggling the ENABLE signal for short time periods, thereby holding back the pulse rate to reach a preset, lower value. The preset values can changed based on the length of the pulse train. For example, in a police model, the system could be preprogrammed such that a single trigger pull will produce a 5-second long power supply activation period. For the first 2 seconds of that 5-second actuation period the microprocessor could be programmed to control (pull back) the pulse rate to 19 pulses per second (pps), while for the last 3 seconds of the 5-second activation period the pulse rate could be programmed to be reduced to 15 pps. If the operator continues to hold the trigger down, after the 5-second cycle has been completed, the X26 system could be programmed to continue discharging at 15 pps for as long as the trigger is held down. The X26 system could alternatively be programmed to produce various different pulse repetition rate configurations as described, for example, in Table 5.

[0120]

	TABLE 5
	Operating
	Duration	Pulse Repetition Rate
	(Seconds)	(Pulses Per Second)
	0-2	17
	2-5	12
	5-6	0.1
	6-12	11
	12-13	0.1
	13-18	10
	18-19	0.1
	19-23	9

[0121] Such alternative pulse repetition rate configurations could be applied to a civilian version of the X26 system where longer activation periods are desirable. In addition, lowering the pulse rate will reduce battery power consumption, extend battery life, and potentially enhance the medical safety factor.

[0122] To explain the operation of the X26 system illustrated in FIGS. 21-24 in more detail, the operating cycle of the HVA can be divided into the following four time periods as illustrated in FIG. 26.

[0123] For the first time period, T0 to T1, capacitors C1, C2 and C3 are charged by one, two or three power supplies to the breakdown voltage of spark gap GAP1.

[0124] For the second time period, T1 to T2, GAP1 has switched ON, allowing C1 to pass a current through the primary winding of the high voltage spark transformer T2 which causes the secondary voltage (across E1 to E2) to increase rapidly. At a certain point, the high output voltage caused by the discharge of C1 through the primary transformer winding will cause voltage breakdown across GAP2, across E1 to E2, and across GAP3. This voltage breakdown completes the secondary circuit current path, allowing output current to flow. During the T1 to T2 time interval, capacitor C1 is still passing current through the primary winding of the spark transformer T2. As C1 is discharging, it drives a charging current into both C2 and C3.

[0125] For the third time period, T2 to T3, capacitor C1 is now mostly discharged. The load current is being supplied by C2 and C3. The magnitude of the output current during the T2 to T3 time interval will be much lower than the much higher output current produced by the discharge of C1 through spark transformer T2 during the initial T1 to T2 current output time interval. The duration of this significantly reduced magnitude output current during time interval T2 to T3 may readily be tuned by appropriate component parameter adjustments to achieve the desired muscle response from the target subject.

[0126] Finally, during the time period T0 through T3, the microprocessor measured the time required to generate a single shaped waveform output pulse. The desired pulse repetition rate was pre-programmed into the microprocessor. During the fourth time period, the T3 to T4 time interval, the microprocessor will temporarily shut down the power supply for a period required to achieve the preset pulse repetition rate. Because the microprocessor is inserting a variable length T3 to T4 shut-off period, the system pulse repetition rate will remain constant independent of battery voltage and circuit component variations (tolerance). The microprocessor-controlled pulse rate methodology allows the pulse rate to be software controlled to meet different customer requirements.

[0127] The FIG. 26 timing diagram shows an initial fixed timing cycle TA followed by a subsequent, longer duration timing cycle TB. The shorter timing cycle followed by the longer timing cycle reflects a reduction in the pulse rate. Hence, it is understood that the X26 system can vary the pulse rate digitally during a fixed duration operating cycle. As an example, a 19 pps pulse rate can be achieved during the first 2 seconds of operation and then reduced to 15 pps for 3 seconds, to 0.1 pps for 1 second, and then increased to 14 pps for 5 seconds, etc.

[0128] The embodiment illustrated in FIGS. 23 and 24 utilizes 3 spark gaps. Only GAP1 requires a precise break-over voltage rating, in this case 2000 volts. GAP2 and GAP3 only require a break-over voltage rating significantly higher than the voltage stress induced on them during the time interval before GAP1 breaks down. GAP2 and GAP3 have been provided solely to ensure that if a significant target skin resistance is encountered during the initial current discharge into the target that the muscle activation capacitors C2 and C3 will not discharge before GAP1 breaks down. To perform this optional, enhanced function, only one of these secondary spark gaps (either GAP2 or GAP3) need be provided.

[0129] FIG. 25 illustrates a high voltage section with significantly improved efficiency. Instead of rectifying the T1 high voltage transformer outputs through diodes directly to very high voltages, as is the case with the FIG. 24 circuit, transformer T1 has been reconfigured to provide three series-connected secondary windings (windings 6-7, 8-9 and 9-10) where the design output voltage of each winding has been limited to about 1000 volts.

[0130] In the FIG. 24 circuit, capacitor C1 is charged directly up to 2000 volts by transformer winding 3-4 and diode D1. In the FIG. 25 circuit, C1 is charged by combining the voltages across C5 and C6. Each T1 transformer winding coupled to charge C5 and C6 is designed to charge each capacitor to 1000 volts, rather than to 2000 volts as in the FIG. 24 circuit.

[0131] Since the losses due to parasitic circuit capacitances are a function of the transformer AC output voltage squared, the losses due to parasitic circuit capacitances with the FIG. 25 1000 volt output voltage compared to the FIG. 24 2000 volt transformer output voltage are reduced by a factor of 4. Furthermore, in the FIG. 25 embodiment, the current required to charge C2 is derived in part from capacitor C6, the positive side of which is charged to 2 KV. Hence, to charge C2 to 3 KV, the voltage across transformer winding pins 6 to 7 is reduced to only 1 KV in comparison to the 3 KV level produced across transformer T1 winding 1-2 in the FIG. 24 circuit.

[0132] Another benefit of the novel FIG. 24 and FIG. 25 circuit designs relates to the interaction of C1 to C3. Just before GAP1 breaks down, the charge on C1 is 2 KV while the charge on C3 is 3 KV. After C1 has discharged and the output current is being supported by C2 and C3, the voltage across C3 remains at 3 KV. However, since the positive side of C3 is now at ground level, the negative terminal of C3 will be at −3 KV. Hence a differential voltage of 6 KV has been created between the positive terminal of C2 and the negative terminal of C3. During the time interval when C2 and C3 discharge after C1 has been discharged, the T2 output windings merely act as conductors.

[0133] The X26 system trigger position is read by the microprocessor which may be programmed to extend the duration of the operating cycle in response to additional trigger pulls. Each time the trigger is pulled, the microprocessor senses that event and activates a fixed time period operating cycle. After the gun has been activated, the Central Information Display (CID) 14 on the back of the X26 handle indicates how much longer the X26 system will remain activated. The X26 system activation period may be preset to yield a fixed operating time, for example 5 seconds. Alternatively, the activation period may be programmed to be extended in increments in response to additional, sequential trigger pulls. Each time the trigger is pulled, the CID readout 14 will update the countdown timer to the new, longer timeout. The incrementing trigger feature will allow a civilian who uses the X26 system on an aggressive attacker to initiate multiple trigger pulls to activate the gun for a prolonged period, enabling the user to lay the gun down on the ground and get away.

[0134] To protect police officers against allegations of stun gun misuse, the X26 system may provide an internal non-volatile memory set aside for logging the time, duration of discharge, internal temperature and battery level each time the weapon is fired.

[0135] The stun gun clock time always remains set to GMT. When downloading system data to a computer using the USB interface module, a translation from GMT to local time may be provided. On the displayed data log, both GMT and local time may be shown. Whenever the system clock is reset or reprogrammed, a separate entry may be made in the system log to record such changes.

[0136] It will be apparent to those skilled in the art that the disclosed electronic disabling device for generating a time-sequenced, shaped voltage output waveform may be modified in numerous ways and may assume many embodiments other than the preferred forms specifically set out and described above. Accordingly, it is intended by the appended claims to cover all such modifications of the invention which fall within the true spirit and scope of the invention.