Low voltage amplifier having a class-AB control circuit

[0001] This application claims priority under 35 U.S.C. § 119(e) of provisional application Ser. No. 60/586,013 filed Jul. 7, 2004.

[0002] The present invention relates to the field of differential amplifiers and operational amplifiers and more particularly to an amplifier of this kind fabricated in integrated circuit technology having a rail-to-rail common mode range at the amplifier output.

[0003] With the reduction of feature size in complementary metal oxide semiconductor (CMOS) processes, supply voltage requirements decrease. This higher integration density provides that each digital and analog cell in a mixed signal very-large-scale-integration (VLSI) circuit consumes less power. There is a benefit to the digital cells in that performance improves; however, analog cells suffer from the reduction in size since minimum size transistors cannot be used due to noise and offset requirements. Low supply voltage requirements dictate a need for complex circuit solutions, which in some cases result in performance degradation. Thereby efficient topologies that combine low-voltage operation with high power efficiency and small die area is the solution for analog cells.

[0004] An improved class AB control circuit is presented in the following publication: “Compact Low-Voltage Power-Efficient Operational Amplifier Cells for VLSI” by Klaas-Jan de Langen and Hohan H. Huijsing-IEEE Journal of Solid-State Circuits, Vol. 33, No. 10, October 1998 pages 1482-1496 which is incorporated by reference herein. A suitable class AB control circuit having a low number of components and low current consumption is shown in FIG. 1 and is similar to the control circuit shown in FIG. 7 of the foregoing reference. The op-amp has a PMOS input stage MP_INM1 and MP_INP1, allowing common-mode input voltage down to the negative rail. The input stage drives a cascoding and summing circuit including a current mirror MN₈ and MN₉, and cascodes MN₁₀, MN₁₁, MN₁₂, MP₅, MP₆, and MP₇. The summing circuit drives the output stage MP₈ and MN₁₅ and the class-AB control circuit MP₉, MP₁₀, MP₁₁, and MN₁₄. The class AB control uses the simple minimum selector MP₉, MP₁₀, and MP₁₁. In operation, the current of transistor MN₁₅ is measured by transistor MN₁₄. The drain current of transistor MN₁₄ flows through mirror MP₉ and MP₁₀, which is a part of the minimum selector circuit MP₉, MP₁₀, and MP₁₁. Transistor MP₁₀ of the minimum selector circuit operates mainly in the linear region. Only when transistor MP₈ handles large output current does transistor MP₁₀ operate in saturation. Transistor MP₁₁ measures the current of transistor MP₈ and is also part of the minimum selector. The drain current of transistor MP₁₁, which is the output of the minimum selector, flows through transistor MN₇ and steers the class-AB amplifier cascode MN₁₀, MN₁₁, MN₁₂, MP₅, MP₆, and MP₇.

[0005] This architecture is capable of having a supply voltage V_CC as low as the sum of twice the drain voltage at saturation (2V_dsat) and the gate-to-source voltage V_gs, wherein V_CC=V_gs+2V_dsat. This architecture is one of the several known architectures that have the capability of having a lower supply voltage and having a double cascode high impedance node. Moreover, the disclosed class AB control circuit does not introduce additional non-dominant poles as with other known implementations. As a result, this architecture has the best AC performance. Accordingly, the required power for a given bandwidth and phase margin is minimized.

[0006] The class-AB control circuit within the amplifier shown in FIG. 1, however, has a significant disadvantage. Specifically, when the output of the amplifier clips to ground or the negative rail, a surge of current is pulled from the power supply. Accordingly, NMOS device, transistor MN₁₅ goes into the linear triode region because its drain-to-source voltage V_ds is lower than what is necessary for it to stay in saturation. Accordingly, sensing NMOS device, MN₁₄, remains in saturation. Accordingly, when transistor MN₁₅ goes into the linear region for a given amount of drain current, the gate-to-source voltage V_gs of transistor MN₁₅ is much higher in the linear region. Thus, since transistor MN₁₄ is still in the saturation region, it has been conditioned to expect that transistor MN₁₅ will send substantially less current than when it is does during output clipping. Thereby, the gate-to-source voltage V_gs of transistor MN₁₅ causes a significantly larger amount of current to go through transistor MN₁₄. Accordingly, the ratio of the currents supplied to transistors, MN₁₄ and MN₁₅, prior to the output of the amplifier being pulled low is 10/1. Yet, as a result of the output clipping, this same ratio changes to approximately 10/0.01. The drain of transistor MN₁₄ connects to the diode connected transistor MP₉. Transistor MP₉ provides as much current as it can to transistor MN₁₄. The only limitation of current supplied by transistor MP₉ is when the gate-to-source voltage V_gs of transistor MP₉ is too large, wherein there is little voltage remaining if the gate-to-source voltage V_gs is subtracted from the supply voltage. There, however, must be enough gate-to-source voltage V_gs to keep transistor MN₁₄ ‘on’. Thereby, there exists a supply dependency, wherein when the supply voltage is low, there will be less current surge but it will be on the order of 4 microns to 1.8-2.7 milliamps. This poses a significant problem. Furthermore, when the output voltage is below the drain voltage V_dsat for transistor MN₁₅ at saturation, the gain of the summing circuit is dramatically reduced causing the voltage at the gate of transistor MN₁₅ or node N₁₇ to increase significantly. As a result, transistor MN₁₄ is biased at a DC point where a significant amount of additional current flows through transistor MN₁₄ on the order of several milliamps. The exact amount of additional current depends upon the power supply wherein a higher supply will generate a higher current.

[0007] A known approach to minimization of surge current produced by the class-AB control circuit within the amplifier has been to add a resistor between the drain of transistor MN₁₄ and the drain of transistor MP₉ to cause a voltage drop which will lower the amount of surge current. This approach, however, is not an adequate solution, in that it minimally limits the surge current; yet, a substantial amount of surge current exists. Additionally, while this approach reduces surge current, the value of the resistor varies with process, voltage, and temperature. Moreover, the minimized surge current remains power supply dependent.

[0008] Thus, there exists a need for a low voltage amplifier having a class-AB control circuit which generates minimal or no surge current when the output of the amplifier is driven low, in the case where the amplifier is used as a split supply.

[0009] The present invention is directed to overcoming, or at least reducing the effects of one or more of the problems set forth above.

[0010] The following presents a simplified summary in order to provide a basic understanding of one or more aspects of the invention. This summary is not an extensive overview of the invention, and is neither intended to identify key or critical elements of the invention, nor to delineate the scope thereof. Rather, the primary purpose of the summary is to present some concepts of the invention in a simplified form as a prelude to the more detailed description that is presented later.

[0011] To address the above-discussed deficiencies of low voltage amplifiers, the present invention teaches a low voltage amplifier having a class-AB control circuit which generates minimal or no surge current when the output of the amplifier clips to ground or the negative rail. The amplifier includes an input stage, a summing circuit, a class-AB control circuit and an output stage. The input stage connects to a first current source and couples to receive a pair of differential input signals to generate intermediate signals that are summed together using the summing circuit. A second current source supplies the summing circuit with current. The class-AB control circuit receives the summed signals from the summing circuit to provide control for the output stage which generates the amplifier result at the output of the amplifier which delivers current to drive an external load.

[0012] Specifically, the class-AB control circuit includes a first and second p-channel transistor coupled such that the gates of the first and second p-channel transistor coupled together and the source of the first and second p-channel transistor coupled to the first power supply rail. A third p-channel transistor includes a source connected to the drain of the first transistor and a gate the couples to form a positive input to the class-AB control circuit. A fourth p-channel transistor includes a source that connects to the drain of the second transistor and a gate that connects to the gate of the third p-channel transistor to form a first terminal of the differential output. A first n-channel transistor includes a drain connected to the drain of the fourth p-channel transistor, a source connected to the second power supply rail, and a gate connected to the negative input to form a second terminal of the differential output for the class-AB control circuit. A second n-channel transistor includes a gate coupled to it's drain, wherein the drain connects to the drain of the third p-channel transistor. The source of the second n-channel transistor connects to the second power supply rail.

[0013] In operation, when the output of the amplifier clips to ground or the negative rail, a substantially smaller amount of surge current is pulled from the power supply in the form of an initial spike wherein a miniscule amount of current within the range of 50 to 150 microamps follows.

[0014] The advantages include but are not limited to a low voltage amplifier having a class-AB control circuit which generates minimal or no surge current when the output of the amplifier clips to ground or the negative rail. This amplifier is power supply independent. Moreover, the improved amplifier is less dependent on process variation. This solution provided by the improved amplifier completely eliminates the problems that arise as a result of surge current; whereas other solutions only diminish the effects of surge current.

[0015] The following description and annexed drawings set forth in detail certain illustrative aspects and implementations of the invention. These are indicative of but a few of the various ways in which the principles of the invention may be employed.

[0023] One or more exemplary implementations of the present invention will now be described with reference to the attached drawings, wherein like reference numerals are used to refer to like elements throughout. The various aspects of the invention are illustrated below in a low voltage amplifier having an input stage, a summing circuit, a class-AB control circuit, and an output stage, although the invention and the appended claims are not limited to the illustrated examples.

[0024] One embodiment of the low voltage amplifier having a class-AB control circuit configured as a folded cascode amplifier which generates minimal or no surge current when the output of the amplifier clips to ground or the negative rail is illustrated in FIG. 2. As shown, an input stage 205 connects to receive a positive and a negative input, INP₂ and INM₂, to receive a differential input signal. A first and second current source, I₁ and I₂, connect to the input stage 205 and the summing circuit 210, respectively. Class-AB control circuit 215 connects between summing circuit 210 and output stage 220.

[0025] Specifically, input stage 205 includes a differential pair, wherein a first p-channel transistor MP_INM2 connected to a second p-channel transistor MP_INP2. The gate of the first p-channel transistor MP_INM2 connects to the negative input INM₂, while the gate of the second p-channel transistor MP_INP2 connects to the positive input INP₂. Both sources of the first and second transistors, MP_INM2 and MP_INP2, connect to the first current source I₁ which supplies bias current to the differential pair, MP_INM2 and MP_INP2. Summing circuit 210 connects to the input stage 205, wherein the drain of both transistors, MP_INM2 and MP_INP2, connect to the source on a first n-channel transistor MN₃₄ and source of a second n-channel transistor MN₃₅, respectively. A third current source I₃ connects between the second power supply rail AV_ss and the source of transistor MN₃₄. The first transistor MN₃₄ connects with the third n-channel transistor MN₃₃ to form a current mirror, wherein the transistor MN₃₃ is diode coupled and the gates of transistors MN₃₃ and MN₃₄ connect together. The drain of transistor MN₃₃ connects to receive current from the second current source I₂. The source of n-channel transistor MN₃₆ connects to the source of transistor MN₃₅, wherein a fourth current source I₄ connects between the source and the second power supply rail AV_SS. A pair of p-channel transistors, MP₃₅ and MP₃₇, couple to form another current source, wherein transistor MP₃₅ is the diode connected transistor that couples its gate to the gate of MP₃₇. P-channel transistor MP₃₉ connects to transistors MN₃₆ and MP₃₇, wherein the drain of transistor MP₃₉ couples to the drain of transistor MN₃₆ and the gate of transistor MP₃₉ connects to the gate of transistor MP₃₇. The drain of transistor MP₃₇ connects to the drain of MN₃₅. The drain of MP₃₅ connects to the drain of MN₃₄. The sources of transistors, MP₃₇ and MP₃₉, connect to a drain of p-channel transistor MP₃₈. The gate of transistor MP₃₈ couples to the gate of transistor MP₃₄, wherein the sources of both transistors MP₃₄ and MP₃₈ coupled to the first power supply rail AV_DD. The drain of transistor MP₃₄ connects to the source of transistor MP₃₅.

[0026] The class-AB control circuit 215 includes p-channel transistor MP₄₂ having a gate which couples to form a first input of the control circuit 215. This first input of control circuit 215 couples to the drain of transistor MP₃₇ of the summing circuit 210. A n-channel transistor MN₃₇ couples to form the second input of control circuit 215, wherein the gate of transistor MN₃₇ forms the second input and couples to the drain of the transistor MN₃₆ of the summing circuit 210. A diode connected n-channel transistor MN₃₁ couples between the second power supply rail AV_SS and the drain of the transistor MP₄₂. A p-channel transistor MP₄₄ includes a drain the connects to the drain of transistor MN₃₇ and a gate that connects to the gate of transistor MP₄₂. P-channel transistors MP₄₀ and MP₄₃ have their gates connected together. Each source of transistors MP₄₀ and MP₄₃ connect to the first power supply rail AV_DD. Each drain of transistors MP₄₀ and MP₄₃ connects to a respective source of the transistor MP₄₂ and MP₄₄. The gate of transistor MP₄₃ connects to the drain of transistor MP₄₄.

[0027] Output stage 220 includes a p-channel transistor MP₄₆ and a n-channel transistor MN₃₈. The gate of transistor MP₄₆ couples to the first output of control circuit 220, wherein the gate of MP₄₄ connects to the gate of transistor MP₄₆. The gate of transistor MN₃₈ couples to the second output of control circuit 220, wherein the gate of MN₃₇ connects to the gate of transistor MN₃₈. The drains of transistors MP₄₆ and MN₃₈ connect to form the output Out₂ of amplifier 200. The source of transistors MP₄₆ and MN₃₈ connect to the first and second power supply rail, AV_DD and AV_SS, respectively. In addition, the output stage 220 includes first and second AC compensation circuit R₃, C₃, R₄, and C₄. The first AC compensation circuit, R₃ and C₃, connects between the gate of transistor MP₄₄ and the output Out₂.

[0028] In operation, input stage 205 transistors MP_INM2 and MP_INP2 are connected to common gate transistors MN₃₃, MN₃₄, MN₃₅, and MN₃₆, wherein transistors MN₃₄, MN₃₅ and MN₃₆ make up the cascoding nodes. The conversion of the signal happens using transistors MP₃₄ and MP₃₅. A first pair of drains of transistors, MP₃₇ and MN₃₅, and a second pair of drains of transistors, MP₃₉ and MN₃₆ form a first and second summing node, N₂₆ and N₂₇, respectively. It is at these nodes, N₂₆ and N₂₇, where the summing and gain occurs for the input signals. The signal goes into the source of transistors, MN₃₅ and MN₃₆, and goes through the drain of both transistors, MN₃₅ and transistor MN₃₆, respectively. The signal also goes into the source of transistor MN₃₄ and exits through the drain of transistor MN₃₄. It travels further to the gate and drain of transistor MP₃₅. The signal is applied to the gate of MP₃₈ and exits through the drain of transistor MP₃₈. Following, the signal goes into the source of transistors MP₃₇ and MP₃₉.

[0029] From the first power supply rail AV_DD to the gate of transistor MP₄₆, the voltage level is decremented by the gate-to-source voltage V_gs of transistor MP₄₆ which is the threshold voltage for transistor MP₄₆ [V_gst=(V_gs−V_t)≈100 mv]. Transistor MP₄₃ must stay in saturation to keep amplifier 200 functional. As a result, transistor MP₄₄ must be a low threshold voltage transistor. Specifically, the voltage applied to the drain of transistor MP₄₃ is the value of the first power supply voltage AV_DD decremented by the gate-to-source voltage V_gs of transistor MP₄₆ and incremented by the gate-to-source voltage V_gs of transistor MP₄₄. Since transistor MP₄₄ is a low threshold voltage transistor, however, the threshold voltage V_t will be approximately 0.4 v as opposed to the threshold voltage V_t of transistor MP₄₆ which is approximately 0.8 v. Thereby, approximately 0.4 v will be applied across the source to drain of transistor MP₄₃, in an effort to maintain transistor MP₄₃ in saturation.

[0030] Nodes, N₂₆ and N₂₇, have voltages that in phase, wherein when one node increases in voltage, the other node increases accordingly. When the output of the amplifier Out₂ is driven low, nodes, N₂₆ and N₂₇, transitioned high. As a result, the voltage at the gate of transistor MP₄₆ is much higher than in the known implementation of the low voltage amplifier. Therein, transistor MP₄₆ is almost turned off. Transistor MP₄₄, however, maintains its low threshold voltage V_t. In addition, since the voltage in node N₂₆ is higher than previously, there will not be enough voltage for transistor MP₄₃ to be on completely in saturation.

[0031] Further, the voltage at nodes N₂₆ and N₂₇ are in phase; thereby, as the voltage of node N₂₇ increases, the voltage of node N₂₆ increases. Accordingly, the DC current path is cut off by transistor MP₄₄. Since transistor MP₄₄ is a natural or low threshold voltage PMOS device, transistor MP₄₃ stays in the saturation region during normal operation. The threshold voltage V_t of transistors, MP₄₃ and MP₄₄, track through all process variation, if there is only one implant difference in the fabrication process. This guarantees a robust manufacturable solution. The end result is that class-AB control circuit 215 functions in the same way it did before having added transistor MP₄₄, with the exception of minimal to zero surge current.

[0032] In summary, this approach provides an additional PMOS device, transistor MP₄₄, that turns off the DC current path that is composed of diode connected PMOS transistor MP₄₃ and sensing NMOS transistor MN₃₇ during the output clipping condition to minimize the surge current of amplifier 200. The low voltage amplifier in accordance with the present invention is typically used in low voltage applications of 1.5V.

[0033] In the alternative, a second embodiment may include a transistor MP₄₄ that is implemented using a standard threshold voltage V_t device. In order for this implementation to be effective, transistors MP₄₃ and MP₄₄, would have to be properly sized, since transistor MP₄₃ will operate in the linear region given this alteration.

[0034] Another alternative embodiment of the low voltage amplifier 300 having an improved class-AB control circuit is shown in FIG. 3. This embodiment corresponds to the sample and hold circuit 400 that is illustrated in FIG. 4. The distinction between this embodiment of the amplifier 300 from the amplifier 200 in FIG. 2 is the addition of a first and a second shutdown circuit, 325 and 330. Specifically, amplifier 300 includes an input stage 305, a summing circuit 310, a class-AB control circuit 315, an output stage 320, and a first and a second shutdown circuit, 325 and 330; wherein, input stage 305, summing circuit 310, class-AB control circuit 315, and output stage 320 couple and are arranged similarly to amplifier 200 in FIG. 2.

[0035] The first shutdown circuit 325 includes p-channel transistors MP₃₆, MP₄₁, and MP₄₅. Each source of transistors MP₃₆, MP₄₁, and MP₄₅ connect to the first power supply rail AV_DD. The drain of transistors MP₃₆, MP₄₁, and MP₄₅ connects to the respective gates of transistor MP₃₈, MP₄₃, and MP₄₆. In operation, when the SHDN pin is high, transistors MP₃₆, MP₄₁ and MP₄₅ have a gate-to-source voltage V_gs of zero which means that the transistors are off. Similarly, in the second shutdown circuit 330, inverter 332 inverts the SHDN signal such that transistors, MN₂₈, MN₃₀ and MN₃₂, have a gate to source voltage of zero. Thereby, transistors, MN₂₈, MN₃₀ and MN₃₂, are off as well.

[0036] Specifically, transistors, MN₃₅ and MN₃₆, form a differential pair comparing the fixed voltage generated by the fixed current provided by the second current source I₂ and the voltage at node N₂₈ which depends on the sensing circuitry in output stage 320. The current supplied by the second current source I₂ to the gate of transistor MN₃₃ sets the gate-to-source voltage V_gs of transistor MN₃₃. The differential pair, transistors, MN₃₅ and MN₃₆, compares this voltage with the voltage at node N₂₈ and sets the voltage at nodes N₂₆ and N₂₇ so that the voltage at node N₂₈ is the same as the gate-to-source voltage V_gs of transistor MN₃₃. When the SHDN pin goes high, however, transistors MN₂₈, MN₃₀ and MN₃₂ in the second shutdown circuit 330 turn on and clamp the class AB control circuit 315 and clamp the output stage 320. In addition, when the SHDN pin goes high, transistors MP₃₆, MP₄₁ and MP₄₅ turn on. Specifically, transistors MP₃₆, MP₄₁ and MP₄₅ short circuit the gate-to-source voltage of transistors MP₃₈ and MP₄₆ in the summing circuit 310 and the output stage 320, accordingly. Effectively, transistors, MP₃₈ and MP₄₆, are turned off, thereby turning off amplifier 300.

[0037] In FIG. 4, we have an illustration of a known sample and hold circuit 400 application from National Semiconductor's LMV 341 datasheet where the effects of the surge current problem may be observed. Specifically, sample and hold circuit 400 includes a first amplifier 410 connected to a second amplifier 420. Further, capacitor C₁ couples to charge up to the voltage of an input signal supplied to the first amplifier 410. Voltage source V₄ provides the power supplied for both amplifiers 410 and 420. The output of each amplifier feeds back to the negative input of each amplifier, 410 and 420, respectively. Voltage source V₁ provides input voltage to the positive input of amplifier 410, while voltage supplies, V₂ and V₃, provide voltage for the shut down signal SHDN.

[0038] In FIG. 5, the operation of this sample and hold circuit 400 is shown, wherein a timing diagram is displayed to illustrate the exemplary operation of the sample and hold circuit application 400 using known amplifiers. First amplifier 400 is on when the shutdown signal SHDN is high. Accordingly, the first amplifier 410 turns on and off, tracking the SHDN signal clock frequency. Second amplifier 420 remains on. It can be noticed that the DC current through MP₄₃ jumps into the 10 mA range when the output of the op-amp is clipped to ground for both the first and second amplifier, 410 and 420. In the first amplifier 410, the surge current appears when the output is being clipped to ground and the shutdown signal SHDN is high. In the second amplifier 420, the surge current appears simply when the output is clipped to ground. When the output Out clips to ground and the shutdown pin SHDN goes high, each amplifier, 410 and 420, are turned on and they generate a surge of current. As shown and noted using points A, B, C and D, each surge of current for the first amplifier 410 corresponds to the shutdown pin SHDN going high during the time when the input and output signals are low. The surge for the second amplifier 420, however, always stays high as shown at points E and F, because the second amplifier 420 is always on. As a result, the surge current presents a problem where the RMS power burned in this circuit 400 is much higher than necessary.

[0039] Referencing FIG. 6, the operation of this sample and hold circuit 400 is shown, wherein a timing diagram is displayed to illustrate the exemplary operation of the sample and hold circuit application 400 using amplifiers in accordance with the present invention from FIG. 3. When the shutdown signal SHDN is high, the first amplifier 410 turns on. Accordingly, the first amplifier 410 turns on and off, tracking the SHDN signal clock frequency. The second amplifier 420, however, remains on. It can be noticed at points, G and I, that the DC current through MP₄₃ spikes to approximately 600-800 microamps when the output of the op-amp is clipped to ground for both the first and second amplifier, 410 and 420. In the first amplifier 410, the surge current appears when the output is being clipped to ground and the shutdown signal SHDN is high. In the second amplifier 420, the surge current appears simply when the output is clipped to ground. Specifically, when the input and output of the sample and hold circuit 400 having amplifiers in accordance with the present invention goes low, the surge of current is spikes at the onset of the shutdown signal SHDN going high. Following, the spike of surge current, there remains a insubstantial amount of current ranging from approximately 25 to 68 microamps as shown at points, H and J. Therein, the surge current is diminished utilizing amplifiers in accordance with the present invention. This application is just one of the many illustrative examples where the current surge problem may be minimized.

[0040] Those of skill in the art will recognize that the physical location of the elements illustrated in FIGS. 2 and 3 can be moved or relocated while retaining the function described above. For example, another embodiment may use a transistor having a standard threshold voltage for transistor MP₄₄ and use of a resized transistor MP₄₃ double in size as express previously.

[0041] Advantages of this design include but are not limited to a low voltage amplifier having a class-AB control circuit that minimizes surge current, wherein there is a high performance, simple, and cost effective design.

[0042] The reader's attention is directed to all papers and documents which are filed concurrently with this specification and which are open to public inspection with this specification, and the contents of all such papers and documents are incorporated herein by reference.

[0043] All the features disclosed in this specification (including any accompany claims, abstract and drawings) may be replaced by alternative features serving the same, equivalent or similar purpose, unless expressly stated otherwise. Thus, unless expressly stated otherwise, each feature disclosed is one example only of a generic series of equivalent or similar features.

[0044] The terms and expressions which have been employed in the foregoing specification are used therein as terms of description and not of limitation, and there is no intention in the use of such terms and expressions of excluding equivalents of the features shown and described or portions thereof, it being recognized that the scope of the invention is defined and limited only by the claims which follow.