Constant-on-time power-supply controller and related system and method

[0001] This application claims priority to U.S. Provisional Application Ser. No. 60/687,165, filed on Jun. 3, 2005, which is incorporated by reference.

[0002] A traditional constant-on-time buck-converter regulates an output voltage by using the ripple on the output voltage as a PWM ramp signal to control the turn-on instant of the transistor that couples the input voltage to the filter inductor.

[0003] A constant-on-time buck converter may have a number of advantages over other types of power supplies. For example, a constant-on-time buck converter typically operates at a constant-frequency for steady-state loads, has high efficiency over a wide load range, requires few of any additional compensation components and responds quickly to changes in the load. Furthermore, such buck converter may transition relatively seamlessly between a pulse-width modulation mode (normal load conditions where switching frequency relatively constant) and a pulse-frequency modulation mode (heavy or light load conditions where the switching frequency increases or decreases, respectively).

[0004] FIG. 1 is a schematic diagram of a traditional constant-on time buck converter, i.e., power supply 10, and FIGS. 2A and 2B are respective plots of the voltages VOUT, Vref, and Q1_drive of FIG. 1.

[0005] Referring to FIGS. 1-2B, the operation of the power supply 10 is described.

[0006] During a discharge time Toff, the transistor Q1 is deactivated and the transistor Q2 is activated such that the decaying current I_L flowing through the inductor L also flows through the closed transistor Q2. As I_L decays, VOUT ramps downward toward Vref as shown in FIG. 2A.

[0007] When VOUT ramps below Vref, a comparator 12 activates a one shot 14, which activates Q1 and deactivates Q2 for a predetermined “constant-on” or charge time Ton. During the charge time Ton, an increasing current I_L flows from the input voltage Vin, through the transistor Q1 and the inductor L, to the filter capacitor Co and load Ro. As I_L increases, VOUT ramps upward as shown in FIG. 2.

[0008] After the elapse of the predetermined charge time Ton, the one shot 14 deactivates Q1 and activates Q2 and the above-described cycle repeats.

[0009] There are two components to the ripple on VOUT.

[0010] The first component is the in-phase component, which is the voltage generated by current flowing through the equivalent series resistance (ESR) of the output filter capacitor Co. The in-phase component is in phase with the inductor current I_L, because the voltage across a resistor is in phase with the current through a resistor.

[0011] The second component is the out-of-phase component, which is generated by the charging and discharging of the output filter capacitor Co. The out-of-phase component is out of phase with the inductor current I_L, because the phase of the voltage across a capacitor lags the phase of the current through the capacitor.

[0012] Therefore, as discussed below, the value of the ESR affects the stability of the feedback loop of the power supply 10.

[0013] Generally, the loop is stable where f_ESR≦f_SW/π, where f_SW=1/(Ton+Toff)(the switching frequency), and f_ESR=1/(2π·ESR·Co).

[0014] Consequently, as long as both the ESR and output filter capacitor Co are relatively large (e.g. ESR≧40 milliohms (mΩ)) then the in-phase component of the ripple on VOUT is the dominant component, and thus the phase shift of the ripple relative to the inductor current I_L is relatively small. That is, the in-phase component of the ripple caused by the portion of I_L that flows through the ESR “swamps out” the out-of-phase component of the ripple.

[0015] Therefore, a traditional constant-on-time power supply includes an output filter capacitor Co having an ESR that is large enough to provide a stable feedback loop.

[0016] Recently, filter capacitors having ESR values of 5 mΩ or less have become available; it is sometimes desirable to use such a low-ESR filter capacitor in a buck-converter power supply with a relatively high steady-state switching frequency to reduce the size and cost of the converter.

[0017] Unfortunately, using such a low-ESR capacitor may render a traditional constant-on-time power supply unstable. An unstable power supply may have too large of a voltage tolerance VT as described below in conjunction with FIG. 3, or may oscillate.

[0018] FIG. 3 is a plot of VOUT of FIG. 1 simulated for a 13 Ampere (A) step increase in the load current lo followed by a 13 A step decrease in lo, where VIN=10 Volts (V), Vref=1.76 V, L=1 microhenry (μH), Co=800 microfarads (μF), ESR=0, and f_SW≈300 kilohertz (kH). The low value of ESR decreases the stability margin of the feedback loop, and thus causes transient “ringing” on VOUT in response to the step change in lo. Typically, a customer specifies the maximum voltage tolerance VT, which is the difference between the maximum droop of VOUT in response to a step increase in the load current lo and the maximum peak of VOUT in response to a step decrease in lo. In this example, the transient “ringing” on VOUT results in a VT of approximately 70 millivolts (mV). Because VT is inversely proportional to the value of the output filter capacitor Co, one typically chooses a value of Co large enough to provide the specified value of VT.

[0019] Some integrated-circuit (IC) manufacturers have developed constant-on-time topologies that allow the use of a low-ESR filter capacitor. But unfortunately, these topologies may require additional feedback and compensation circuitry and that the power-supply controller chip have an additional pin, and may yield a relatively poor regulation of VOUT.

[0020] An embodiment of the invention is a power-supply controller that includes an adder and a control circuit. The adder generates a sum of a sense voltage and a regulated output voltage generated by a filter inductor. The sense voltage is generated by a sense circuit that sources a current to the filter inductor while the inductor is uncoupled from an input voltage, and the sense voltage is related to the sourced current. The control circuit couples the filter inductor to the input voltage for a predetermined time in response to the sum having a predetermined relationship to a reference voltage.

[0021] Such a power-supply controller may yield a relatively tight and stable regulation of the output voltage with a low-ESR filter capacitor, with no additional compensation components, and with no additional pin on the power-supply-controller chip.

[0032] In the following detailed description of exemplary embodiments of the invention, reference is made to the accompanying drawings, which form a part hereof. The detailed description and the drawings illustrate specific exemplary embodiments by which the invention may be practiced. These embodiments are described in sufficient detail to enable those skilled in the art to practice the invention. It is understood that other embodiments may be utilized, and other changes may be made, without departing from the spirit or scope of the invention. The following detailed description is therefore not to be taken in a limiting sense.

[0033] FIG. 4 is a schematic diagram of a constant-on-time power supply 40 according to an embodiment of the invention. As further described below, the power supply 40 allows the use of a filter capacitor Co having an ESR of less than 40 mΩ, for example from approximately 0 mΩ-5 mΩ. Furthermore, the power supply 40 requires no additional pin on the power-supply controller integrated circuit (IC), requires no additional compensation circuitry, and yields a relatively tight regulation of VOUT as compared to prior constant-on-time supplies such as the supply 10 of FIG. 1. Moreover, for a given VT, the supply 40 may allow one to use a smaller filter capacitor Co than prior supplies such as the supply 10.

[0034] The supply 40 includes a power-supply controller IC 42 and a filter circuit 44, and provides a regulated output voltage VOUT to a load, which is represented by a resistor Ro.

[0035] In addition to the transistors Q1 and Q2, the comparator 12, and the one shot 14, the IC 42 includes an adder 46, which adds an inverse of a sense voltage V_sense to VOUT, and which provides the result V_sum to an inverting input node of the comparator 12. V_sense effectively enhances or replaces the contribution to the feedback signal of the in-phase component of the VOUT ripple generated by ESR, and thus allows the power supply 40 to maintain a tight and stable regulation of VOUT even where ESR is relatively low or even near 0Ω. Furthermore, because the on resistance Rds(on) of the transistor Q2 generates V_sense at the source of Q2, the adder 46 is the only additional circuitry that the IC 42 includes as compared to the power supply 10 of FIG. 1. In addition, because Q2 generates V_sense internally, the IC 42 needs no additional pin to receive V_sense. Moreover, in an alternate embodiment where Q1 and Q2 are disposed external to the IC 42, a current-sense pin (not shown in FIG. 4) of the IC is typically connected to the source of Q2 for current-limiting purposes. Consequently, because the current-sense pin inherently receives V_sense, the IC 42 needs no additional pin to receive V_sense.

[0036] The filter circuit 44 includes the filter inductor L having an equivalent series resistance Rcs, and includes the output filter capacitor Co having a relatively low ESR.

[0037] FIGS. 5A-5C are respective plots of the voltages V_sum and Vref, Q1_drive, and Q2_drive of FIG. 4 according to an embodiment of the invention. Although these voltages are shown having an approximately 50% duty cycle, they may have other duty cycles as discussed below.

[0038] Referring to FIGS. 4-5C, the operation of the power supply 40 is described according to an embodiment of the invention.

[0039] During a discharge time Toff, the transistor Q1 is deactivated and the transistor Q2 is activated such that the decaying current I_L flowing through the inductor L also flows through the transistor Q2. As I_L decays, V_sum ramps downward toward Vref as shown in FIG. 5A.

[0040] When V_sum ramps below Vref, the comparator 12 activates the one shot 14, which activates Q1 and deactivates Q2 for a predetermined “constant-on” charge time Ton, the value of which for a steady-state load is Ro given by the following equation

[0041] $\begin{array}{cc}\mathrm{Ton}=\frac{K\left(\mathrm{VOUT}\right)+I_L·Q{2}_{\mathrm{Rds}\left(\mathrm{on}\right)})}{\mathrm{Vin}}& \left(1\right)\end{array}$
where K is a constant that depends on the circuit topology and component values. The dependence of Ton on I_L·Q2_Rds(on) allows the switching frequency f_sw to be substantially independent of the load current lo within a predetermined range of lo. That is, by changing Ton, and thus the switching duty cycle, the power supply 40 maintains f_sw at a substantially constant frequency for a steady-state load Ro. Furthermore, because at the beginning of Ton the voltage V_sense changes from a negative to a positive voltage with respect to ground, V_sum “jumps” downward by a DC offset voltage V_offsetdown. The adder 46 may be designed to effectively “ignore” a positive V_sense by assigning a predetermined value such as 0V to V_sense when V_sense is positive. For example, one may connect a diode (not shown) between the source of Q2 (cathode) and the inverting input of the adder 46 (anode) so that the diode blocks V_sense where V_sense is positive. Other current topologies for implementing this function are known, and therefore, are not described in detail.

[0042] During Ton, an increasing current I_L flows from the input voltage Vin, through the transistor Q1 and the inductor L, and to the filter capacitor Co and the load Ro. As I_L increases, VOUT, and thus V_sum, ramps upward.

[0043] After the elapse of the predetermined on time Ton, the one shot 14 deactivates the transistor Q1 and activates the transistor Q2 and the above-described cycle repeats. Because at the end of Ton V_sense changes from a positive to a negative voltage, V_sum “jumps” upward by a DC offset voltage V_offsetup. Furthermore, because I_L is at a minimum at the beginning of Ton and at a maximum at the end of Ton, V_offsetdown is typically less than V_offsetup, assuming that the adder 46 ignores V_sense when V_sense is positive as discussed above.

[0044] Referring to FIG. 4, although the adder 46 is shown inverting V_sense, the adder may have two noninverting inputs and the supply 40 may include an inverter or other circuitry to invert V_sense.

[0045] FIG. 6 is a plot of VOUT of FIG. 4 simulated for a 13 Ampere (A) step increase in the load current lo followed by a 13 A step decrease in lo where Vin=10V, Vref=1.76V, L=1 microhenry (μH), Co=800 microfarads (μF), and ESR=0; these are the same parameters as plotted in FIG. 3 for the power supply 10 of FIG. 1. As compared to the supply 10, the modified topology of the power supply 40 stabilizes the supply so as to virtually eliminate transient “ringing” on VOUT in response to the step changes in the load current, and lowers VT from about 70 mV to about 35 mV. Because for the same parameters, the VT of the supply 40 is approximately ½ the VT of the supply 10, if one specifies a VT of 70 mV, then he can reduce the value of Co of the supply 40 to 400 μF, which is half the 800 μF value of Co of the supply 10. That is, using a Co=400 μF in the power supply 10 yields a VT of approximately 70 mV. This reduction in the value of Co may reduce the size and cost of Co as compared to the Co of the supply 10, and thus may reduce the size and cost of the supply 40 as compared to the supply 10.

[0046] FIG. 7 is a schematic diagram of a sensing circuit 50 and the portion of the power supply 40 of FIG. 1 to which the sensing circuit is connected according to an embodiment of the invention. The sensing circuit 50 can be included in the power supply 40 of FIG. 4 to provide a temperature-stable value for V_sense. That is, the circuit 50 reduces the sensitivity of V_sense to temperature as compared to V_sense being measured directly across Q2 as shown in FIG. 4. The circuit 50 may be entirely or partially disposed on the IC 42, or may be totally disposed external to the IC 42. However, where the sensing circuit 50 is disposed entirely on the IC 42, the components of the sensing circuit may be more closely matched to each other; therefore, the sensing circuit may provide V_sense with a lower temperature sensitivity than where some or all of the components of the sensing circuit are disposed off chip.

[0047] In addition to the transistor Q2, the sensing circuit 50 includes transistors Q3 and Q4, a differential amplifier 52, and a resistive sense element 54, here a sense resistor. The transistors Q2 and Q3 have similar dimensions, and the gain G of the amplifier 52 and the value of the sense amplifier 54 are selected such that I_sense/I_Q2 is small, for example 1/1000. Also, the thermal response of Q2 closely matches that of Q3; a sufficient matching of the thermal response typically occurs where Q2 and Q3 are disposed on the same IC.

[0048] The operation of the sensing circuit 50 is now described in conjunction with FIGS. 4 and 7 according to an embodiment of the invention. For purposes of example, the temperature sensitivity of the resistor 54 is ignored, and the gain G of the amplifier and the value of the sense resistor 54 are selected such that I_sense/I_Q2≈1/1000.

[0049] When the one shot 14 (FIG. 4) deactivates the transistor Q1 and activates the transistor Q2 during the discharge time T_off (FIG. 5B), the one shot also activates the transistor Q3.

[0050] The amplifier 52 generates an output signal Q4_drive that equals G(Vs_Q3−Vs_Q2), where Vs_Q3 is the voltage at the source of the transistor Q3, and Vs_Q2 is the voltage at the source of the transistor Q2. Because Q2 and Q3 have similar dimensions and I_sense<<I_Q2, Vs_Q2<Vs_Q3 for I_Q2>0.

[0051] The signal Q4_drive causes the transistor Q4 to source to the element 54 the current I_sense, which is proportional to Vs_Q3−Vs_Q2; consequently, V_sense is also proportional to Vs_Q3−Vs_Q2. More specifically, because the transistor Q4 acts as a voltage follower, I_sense≈[G(Vs_Q3−Vs_Q2)−Vt]/(sense element 54), where Vt is the threshold voltage of the transistor Q4. Furthermore, because I_sense is relatively small, Vs_Q3 remains relatively constant as compared to Vs_Q2 such that I_sense varies substantially linearly with Vs_Q2. And because Vs_Q2 is the voltage across the Rds(on) of Q2, and thus varies substantially linearly with I_Q2, I_sense varies substantially linearly with I_Q2. Moreover, because I_sense<<I_Q2, I_Q2 approximately equals I_L (i.e., I_Q2≈I_L); consequently, I_sense varies substantially linearly with I_L, as is desired in this embodiment.

[0052] Because Q2 and Q3 are thermally matched, Vs_Q3 and Vs_Q2 shift by substantially the same amount in response to a change in temperature. But even with this temperature-induced shift, the difference between Vs_Q3 and Vs_Q2, and thus Q4_drive, I_sense, and V_sense, remain substantially unchanged with temperature for a given I_L. Put another way, the similar temperature-induced shifts in Vs_Q3 and Vs_Q2 compose a common-mode signal at the input nodes of the differential amplifier 52, which rejects this common-mode signal, thus rendering V_sense less sensitive to temperature than if V_sense were taken directly across the transistor Q2 as shown in FIG. 4.

[0053] FIG. 8 is a block diagram of a computer system 60, which can incorporate the power supply 40 of FIG. 4 according to an embodiment of the invention. The supply 40 may provide regulated voltage VOUT to a processor 62, a memory 64, or another component (not shown) of the computer system 60. Alternatively, the power supply 40 may be installed in any other type of electronic system.

[0054] FIG. 9 is a schematic diagram of an alternate embodiment of the supply 40, where the transistor Q2 is replaced with a diode D1 and a resistor or other resistive element R1. The voltage V_sense may be taken across the R1 as shown, or may be taken across the combination of R1 and D1.

[0055] From the foregoing it will be appreciated that, although specific embodiments of the invention have been described herein for purposes of illustration, various modifications may be made without deviating from the spirit and scope of the invention.