Switching controller for resonant power converter

[0001] 1. Field of Invention

[0002] The present invention relates to resonant power converters, and more particularly to the control circuit of the resonant power converter.

[0003] 2. Description of Related Art

[0004] Power converters are used to convert an unregulated power source to a constant power source. A power converter normally includes a transformer having primary and secondary winding to provide the isolation. Switching devices connected in the primary winding control energy transferring from the primary winding to the secondary winding. The power converters operate at high frequencies allows a size and weight reduction. However, the switching losses, component stresses, and electric-magnetic-interference (EMI) are inherent problems. In the last decades, many resonant power converters have been proposed for high frequency power conversion to reduce the switching losses. Among them, the resonant switching techniques are described in following prior arts: “Regulated switched power circuit with resonant load” by Tamas S. Szepesi, U.S. Pat. No. 4,535,399; “Frequency controlled resonant regulator” by Frank S. Wendt, U.S. Pat. No. 4,631,652.

[0005] For the power saving at no load, a burst switching schemes is developed for resonant power control, “Full load to no-load control for a voltage fed resonant inverter” by John N. Park and Robert L. Steigerwald, U.S. Pat. No. 4,541,041. In various resonant power converters, the leakage inductance of the transformer or additional magnetic components is employed as a resonant inductor to generate the circulating current for achieving the soft switching.

[0006] However, there still exist several drawbacks in conventional technologies. The resonant inductor associated with a resonant capacitor forms a resonant circuit for the power converter to deliver the maximum power to the load. Although the switching frequency of the power converter is controlled to higher or lower than the resonant frequency in response to the change of the load, the change of the switching frequency must be restricted to ensure the linear operation and achieve the soft switching of the resonant power converter. Furthermore, the poor audio-susceptibility of the power converter and the acoustic noise of the burst switching are disadvantages. These shortcomings are the main object of the present invention to overcome.

[0007] One of the objects of the invention is to provide an integrated controller for a resonant power converter. The range of the switching frequency can be determined to achieve the linear operation of the power converter. An independent feedback terminal is coupled to the output of the power converter to receive a feedback signal through an optical-coupler. The switching frequency is thus modulated in accordance with the feedback signal to regulate the output of the power converter. A minimum-frequency programming circuit is connected to a first resistor to program a minimum switching frequency of the power converter. A feedback circuit is coupled to the feedback terminal to receive the feedback signal for generating an adjustment signal. A maximum-frequency programming circuit is connected to a second resistor to determine a maximum switching frequency in response to the adjustment signal. An oscillator is coupled to the minimum-frequency programming circuit and the maximum-frequency programming circuit to generate an oscillation signal for regulating the switching frequency of the power converter. A feed-forward circuit is connected to a feed-forward terminal to receive a feed-forward signal that represents the input voltage of the power converter. The feed-forward signal is applied to modulate the oscillation signal to improve the audio-susceptibility of the power converter without going through the feedback loop of the power converter. The switching frequency is increased in response to decrease of the feedback signal, and the switching frequency is increased in response to the increase of the feed-forward signal. Furthermore, a power management circuit is developed to generate a burst signal to on/off the switching signal of the power converter for light load conditions. The burst signal is generated once the feedback signal is lower than a light-load threshold. The light-load threshold is determined by a power-level signal. The duty cycle of the burst signal is linearly modulated in response to the change of the feedback signal for power saving. Additionally, the burst signal is disabled if the frequency of the burst signal will be operated in audio band, which reduce the acoustic noise of the power converter.

[0020] Reference will now be made in detail to the present preferred embodiments of the invention, examples of which are illustrated in the accompanying drawings. Wherever possible, the same reference numbers are used in the drawings and the description to refer to the same or like parts.

[0021] FIG. 1 shows a resonant power converter including a switching controller 100 to control power switches 10 and 20. The power switch 10 is connected to the input voltage V_IN of the power converter. The power switch 20 is connected to the ground. Power switches 10 and 20 form a half bridge circuit to switch the primary winding N_P of a transformer 50 through a series resonant circuit. An inductance device 30 and a capacitor 40 develop the resonant circuit. A maximum power is delivered to the secondary windings N_S1, N_S2 of the transformer 50 when the switching frequency of power switches 10, 20 is operated at the resonant frequency f₀ of the resonant circuit.

[0022] $\begin{array}{cc}{f}_0=\frac{1}{2\pi \sqrt{L_{30}\times C_{40}}}& \left(1\right)\end{array}$

[0023] The secondary windings N_S1, N_S2 are further coupled to generate the output voltage V_O of the power converter through rectifiers 60, 65 and capacitor 70. An optical coupler 75 is coupled to the output voltage V_O via a resistor 71, a regulator circuit 72 to generate a feedback signal. A feedback terminal FB of the switching controller 100 is connected to the optical coupler 75 to receive the feedback signal for regulating the output voltage V_O. A diode 85 and a capacitor 86 form a charge-pump circuit connected to a bootstrap terminal BS and a bootstrap-ground terminal HGND for providing a power source to a high-side driver circuit of the switching controller 100. The high-side driver circuit is used to drive the power switch 10 via a HD terminal. A low-side driver circuit is utilized to drive the power switch 20 through a LD terminal. A diode 80, a capacitor 83 and resistors 81, 82 develop a detection circuit to detect the voltage of the capacitor 40. A higher voltage will be generated at the capacitor 40 if the output of the power converter is short circuit. A voltage signal V_P detected by the detection circuit is connected to a protection input terminal PRT. When the voltage signal V_P is higher than an over-voltage threshold, the switching controller 100 will initial a delay time. A capacitor 90 connected to a time-delay programming terminal DLY to determine the period of the delay time. Power switches 10, 20 will be turned off to protect the power converter once the delay time is expired.

[0024] A resistor 91 is connected to a minimum-frequency programming terminal F_MIN to determine a minimum switching frequency of the power converter. A resistor 92 is connected to a maximum-frequency programming terminal F_MAX to determine a maximum switching frequency of the power converter. A capacitor 95 is connected to a soft-start terminal SS to generate a soft-start signal in response to the charge of the capacitor 95. The soft-start signal will adjust the switching frequency to limit the power transfer to the output of the power converter during the power on stage. A resistor 96 is connected to a power-level programming terminal VT to generate a power-level signal V_T. The power-level signal V_T is used to determine a light-load threshold for generating a burst signal in response to the feedback signal. The burst signal performs a burst switching for the power converter to save power. Besides, a resistor 97 is connected from the input voltage V_IN to a feed-forward terminal FW of the switching controller 100. A resistor 98 is connected from the feed-forward terminal FW to ground. Resistors 97, 98 form a voltage divider to generate a feed-forward signal that represents the input voltage V_IN. The feed-forward signal is coupled to modulate the switching frequency of the power converter. The switching frequency is increased in response to the increase of the feed-forward signal, which improves the audio-susceptibility of the power converter without going through the feedback of the power converter.

[0025] In the embodiment, the resonant power converter does not utilize leakage inductance of the transformer 50 or other magnetic components to achieve soft switching. Several embodiments will be described below.

[0026] FIG. 2 shows a preferred embodiment of a switching controller 100 according to the present invention. The switching controller 100 includes a voltage-control-oscillator (VCO) 200, a protection circuit 400, a power-management circuit (PM) 500, a control circuit 600 and drive circuits 110, 150. The voltage-control-oscillator 200 is connected to the feedback terminal FB, the soft-start terminal SS, the maximum-frequency programming terminal FMAX, the minimum-frequency programming terminal FMIN and the feedforward terminal FW to generate an oscillation signal PLS. The oscillation signal PLS is used to determine the switching frequency of the power converter. The protection circuit 400 is connected to the protection input terminal PRT and the time-delay programming terminal DLY to generate an enable-signal ENB. The power-management circuit 500 is coupled to the feedback terminal FB and the power-level programming terminal VT to generate the burst signal V_G. The control circuit 600 receives the oscillation signal PLS, the burst signal V_G and the enable-signal ENB for generating switching signals OL and OH. The switching signal OL is connected to the low-side driver circuit 150 to drive the power switch 20. The switching signal OH is connected to the high-side driver circuit 110 drive the power switch 10 through the HD terminal.

[0027] FIG. 3 shows the block diagram of the voltage-control oscillator 200. It includes a minimum-current generation circuit (IN1) 210 connected to the minimum-frequency programming terminal F_MIN and the feed-forward terminal FW to generate a minimum current signal I_MIN. A maximum-current generation circuit (IN2) 250 is connected to the maximum-frequency programming terminal F_MAX, the soft-start terminal SS and the feedback terminal FB to generate a maximum current signal I_MAX. An oscillation circuit (OSC) 300 is coupled to the minimum-current generation circuit 210 and the maximum-current generation circuit 250 to receive the minimum current signal I_MIN and the maximum current signal I_MAX for generating the oscillation signal PLS.

[0028] FIG. 4 shows the minimum-current generation circuit 210. It includes a minimum-frequency programming circuit 220 and a feed-forward circuit 230. The feed-forward circuit 230 includes an operational amplifier 231, a resistor 233, a current source 237 and transistors 232, 234, 235. The operational amplifier 231 associates with the transistor 232 and the resistor 233 forms a voltage-to-current converter to generate a current I₂₃₂ in response to the feed-forward signal. The current I₂₃₂ and the current source 237 are connected to transistors 234 and 235. Transistors 234 and 235 develop a current mirror to generate a current I₂₃₅ in response to the current I₂₃₂ and the current of the current source 237. The current source 237 provides a threshold to produce the current I₂₃₅. The current I₂₃₅ is generated once the current I₂₃₂ is higher than the current of the current source 237.

[0029] The minimum-frequency programming circuit 220 includes a current source 221, a resistor 223, an operational amplifier 224 and transistors 225, 226, 227. The current I₂₃₅ of the feed-forward circuit 230 is connected to the current source 221 to generate a reference signal V_R1 at the resistor 223. The reference signal V_R1 is connected to the operational amplifier 224 to generate a current signal I₂₂₅ at the transistors 225 in accordance with the resistance of the resistor 91. Transistors 226, 227 form a current mirror to generate the minimum current signal I_MIN in response to the current I₂₂₅.

[0030] FIG. 5 shows a maximum-current generation circuit 250. It includes a soft-start circuit 260, a feedback circuit 280 and a maximum-frequency programming circuit 270. A current source 265, an operational amplifier 268, an inverter 267 and transistors 266, 269 form the soft-start circuit 260. The current source 265 is connected to the soft-start terminal SS to charge the capacitor 95 and generate the soft-start signal. The transistor 266 is connected to the soft-start terminal SS to discharge the capacitor 95 once the enable-signal ENB is disabled. The transistor 266 is controlled by the enable-signal ENB through the inverter 267. The operational amplifier 268 and the transistor 269 are connected as an open-drain buffer to receive the soft-start signal.

[0031] A transistor 282, an operational amplifier 286 and resistors 281, 283, 284, 285, 287 develop the feedback circuit 280. The resistor 281 is connected to the feedback terminal FB to provide a power source to the optical-coupler 75 for generating the feedback signal. The transistor 282 is connected to the feedback terminal FB to generate a level-shift signal V_B in accordance with the feedback signal. Resistors 283, 284 and 285 are connected in series to provide attenuation for the level-shift signal V_B. The resistor 283 is connected to the level-shift signal V_B. The resistor 285 is connected to ground. The join of resistors 283, 284 is coupled to the output of the open-drain buffer. The join of resistors 284, 285 is coupled to the negative input of the operational amplifier 286. A reference voltage V_R2 supplies the positive input of the operational amplifier 286. The resistor 287 is connected to build the operational amplifier 286 as inverse amplifier. The output of the operational amplifier 286 produces an adjustment signal V_A. The adjustment signal V_A is thus generated in response to the feedback signal and the soft-start signal.

[0032] The maximum-frequency programming circuit 270 includes a current source 275, an operational amplifier 271 and transistors 272, 273, 274. The adjustment signal V_A is connected to the operational amplifier 271 to generate a current signal I₂₇₂ at the transistors 272 in accordance with the resistance of the resistor 92. The current I₂₇₂ and the current source 275 are connected to transistors 273 and 274. Transistors 273 and 274 develop a current mirror to generate the maximum current signal I_MAX in response to the current I₂₇₂ and the current of the current source 275. The current source 275 provides a threshold. Therefore, the maximum current signal I_MAX is generated once the current I₂₇₂ is higher than the current of the current source 275.

[0033] FIG. 6 shows the oscillation circuit 300. Transistors 310, 311 and 312 are connected as a current mirror to generate a current I₃₁₁ and a current I₃₁₂ in accordance with the minimum current signal I_MIN and the maximum current signal I_MAX. Transistors 314 and 315 are connected as another current mirror to generate a current I₃₁₅ proportional to the current I₃₁₁. The current I₃₁₅ and the current I₃₁₂ are coupled to charge a capacitor 320 through a switch 325 and a switch 326 respectively. The capacitor 320 is connected to the negative input of a comparator 351 and the positive input of a comparator 352. The positive input of the comparator 351 and the negative input of the comparator 352 are connected to trip-point voltages V_H and V_L respectively. The output of the comparator 351 and the output of the comparator 352 are connected to NAND gates 353 and 354. NAND gates 353 and 354 are connected as a latch circuit. The output of the NAND 353 is used to control the on/off of the switch 326. The output of the NAND 353 is utilized to control the on/off of the switch 325 via an inverter 356. The output of the inverter 356 is further connected to an inverter 357. The output of the inverter 357 generates the oscillation signal PLS to determine the switching frequency of the power converter. The switching frequency is increased in response to decrease of the feedback signal. The switching frequency is increased in response to the increase of the feed-forward signal.

[0034] FIG. 7 shows the protection circuit 400. A comparator 405 and a threshold voltage V_R3 form a monitoring circuit coupled to the protection input terminal PRT to generate an over-voltage signal when the voltage of the protection input terminal PRT is higher than the threshold voltage V_R3. The over-voltage signal is coupled to a time-delay circuit 410 to initiate a charge time delay in response to the over-voltage signal. The time-delay circuit 410 will disable the enable-signal ENB to inhibit switching signals OH and OL once the charge time delay is expired. The charge time delay provides a debounce function for detecting of the over-voltage signal. The over-voltage signal is connected to an input of an AND gate 416. Another input of the AND gate 416 is connected to the enable-signal ENB. A current source 412 is coupled to charge the capacitor 90 through a transistor 420 and the time-delay programming terminal DLY. A current source 415 is coupled to discharge the capacitor 90 via a transistor 425 and the time-delay programming terminal DLY. The current I₄₁₅ of the current source 415 is higher than the current I₄₁₂ of the current source 412. For example, the current I₄₁₂ equals to 100 uA and the I₄₁₅ is 200 uA. Therefore, the capacitor 90 will be discharged by the current source 415 when the enable-signal ENB is enabled (logic-high) and the over-voltage signal is disabled. The capacitor 90 is further connected to a transistor 429. The transistor 429 is controlled by a power-on reset signal RST through an inverter 453. The capacitor 90 is therefore discharged during the power on stage of the power converter. Comparators 450 and 451 are coupled to detect the voltage of the capacitor 90 via the time-delay programming terminal DLY. The comparator 450 and the comparator 451 include trip-point voltage V_R4 and V_R5 respectively. The output of the comparator 450 is connected to activate a flip-flop 460. The /Q output of the flip-flop 460 is connected to an AND gate 461 to disable the enable-signal ENB when the voltage of the capacitor 90 is higher than the trip-point voltage V_R4.

[0035] The time delay circuit 410 will initiate a discharge time delay once the enable-signal ENB is disabled (logic-low). The enable-signal ENB will be automatically enabled after the discharge time delay is expired. A current source 417 is coupled to discharge the capacitor 90 through a transistor 427 and the time-delay programming terminal DLY. The enable-signal ENB is coupled to control the transistor 427 via an inverter 418. The capacitor 90 is thus discharged by the current source 417 once the enable-signal ENB is disabled. The current I₄₁₇ of the current source 417, such as 10 uA is lower than the current I₄₁₂. The capacitor 90 will be discharged by the current I₄₁₇ until the voltage of the capacitor 90 is lower than the trip-point voltage V_R5. The output of the comparator 451 is connected to an input of an NAND gate 452. Another input of the NAND gate 452 is connected to the enable-signal ENB. The output of the NAND gate 452 is connected to an AND gate 454. Another input of the AND 454 is coupled to the power-on reset signal RST. The output of the AND gate 454 is coupled to reset the flip-flop 460. The period of the discharge time delay is longer than the period of the charge time delay, which reduces the duty cycle of the power converter during the protection.

[0036] FIG. 8 shows the power management circuit 500. The power management circuit 500 is coupled to the power-level programming terminal VT to receive the power-level signal V_T for producing the light-load threshold and generating the burst signal V_G. A current source 531 is coupled to the power-level programming terminal VT. The current source 531 associated with the resistor 96 generate the power-level signal V_T that is connected to an input of an operational amplifier 532. The operational amplifier 532, a resistor 533 and a transistor 534 develop a voltage-to-current converter to generate a current I₅₃₄ at the transistor 534 in accordance with the power-level signal V_T. The current I₅₃₄ is coupled to transistors 535, 536. Transistors 535 and 536 form a current mirror to generate a current I₅₃₆ at the transistor 536 in accordance the current I₅₃₄. The level-shift signal V_B of the feedback circuit 280 is connected to the input of an operational amplifier 512. A transistor 516, a resistor 513 and the operational amplifier 512 develop another voltage-to-current converter to generate a current I₅₁₆ at the transistor 516 in response to the level-shift signal V_B. Transistors 517 and 518 develop a current mirror to generate a current I_W at the transistor 518 and a current I₅₁₉ at the transistor 519 in response to the current I₅₃₆ and the current I₅₁₆. The current I_W and the current I₅₁₉ can be expressed as,

[0037] $\begin{array}{cc}{I}_W=k2\times \left[\frac{V_B}{R_{513}}-\left(k1\times \frac{V_T}{R_{533}}\right)\right]& \left(2\right)\end{array}$

[0038] $\begin{array}{cc}{I}_{519}=k3\times \left[\frac{V_B}{R_{513}}-\left(k1\times \frac{V_T}{R_{533}}\right)\right]& \left(3\right)\end{array}$

[0039] where k1 is the ratio of mirror transistors 535, 536; k2 is the ratio of mirror transistors 517, 518; k3 is the ratio of mirror transistors 517, 519; R₅₁₃ is the resistance of the resistor 513; R₅₃₃ is the resistance of the resistor 533.

[0040] The current I₅₁₉ and a current source 510 are coupled to charge a capacitor 530 via a switch 541. The current source 510 provides a minimum current to charge the capacitor 530. A current source 515 is further coupled to discharge the capacitor 530 through a switch 542. A ramp signal V_W is therefore generated at the capacitor 530. The capacitor 530 is connected to the negative input of a comparator 560 and the positive input of a comparator 561. The positive input of the comparator 560 and the negative input of the comparator 561 are connected to trip-point voltages V₁ and V₂ respectively. The output of the comparator 560 and the output of the comparator 561 are connected to NAND gates 562 and 563. NAND gates 562 and 563 are connected as a latch circuit. The output of the NAND 562 is used to control the on/off of the switch 542. The output of the NAND 562 is utilized to control the on/off of the switch 541 via an inverter 570. The output of the inverter 570 is further connected to an NAND gate 571 to generate the burst signal V_G at the output of the NAND gate 571. The enable time of the burst signal V_G is fixed, which is determined by the current of the current source 515. The current I₅₁₉ and the current of the current source 510 determine the disable time of the burst signal V_G. Therefore, the disable time of the burst signal V_G is modulated in response to the change of the feedback signal and the load of the power converter.

[0041] The current I_W is coupled to a current arbiter 520 to generate control signals V_W1 and V_W2. The control signal V_W1 will be set to logic-high if the switching frequency of the burst signal V_G will be fall into the audio band. The control signal V_W1 is coupled to disable the burst signal V_G by reset the capacitor 530 via a transistor 543. The control signal V_W2 will be set to logic-low when the feedback signal is higher than the light-load threshold. The control signal V_W2 is connected to another input of the NAND gate 571 to enable the burst signal V_G. The burst signal V_G is coupled to turn off switching signals OH and OL when the burst signal V_G is disabled (logic-low).

[0042] FIG. 9 shows the circuit schematic of the current arbiter 520. The current I_W flows into a transistor 521. Transistors 521, 522, 523 and 524 develop current mirror to generate currents I₅₂₂, I₅₂₃ and I₅₂₄ in response to the current I_W. A current source I₂ and an inverter 552 are connected to compare with the current I₅₂₂. A current source I₃ and an inverter 553 are connected to compare with the current I₅₂₃. A current source I₄ and an inverter 554 are connected to compare with the current I₅₂₄. An inverter 557 is connected to the output of the inverter 552 to generate the control signal V_W2. The output of the inverter 552 is further connected to an input of an OR 559. An input of an AND gate 558 is connected to the inverter 554. Another input of the AND gate 558 is connected to the inverter 553 through an inverter 556. The output of the AND gate 558 is connected to another input of the OR gate 559. The output of the OR gate 559 generates the control signal V_W1. When the feedback signal is higher than the light-load threshold, the current I₅₂₂ will be higher than the current of the current source I₂. Meanwhile, a logic-low will be generated at the input of the inverter 552. When the frequency of the burst signal V_G is lower than a first frequency, the current I₅₂₃ will be lower than the current of the current source I₃. A logic-high will be generated at the input of the inverter 553 When the frequency of the burst signal V_G is higher than a second frequency, the current I₅₂₄ will be higher than the current of the current source I₄. A logic-low will be generated at the input of the inverter 554.

[0043] Therefore, the burst signal V_G is generated once the feedback signal is lower than the light-load threshold. The light-load threshold is programmed by the power-level signal V_T. The burst signal V_G is coupled to on/off switching signals OL and OH. The burst signal V_G will be disabled to reduce the acoustic noise when the frequency of the burst signal V_G may operate in between the first frequency and the second frequency. The frequency between the first frequency and the second frequency is the audio band such as 500 Hz˜20 kHz. FIG. 10 shows waveforms of the ramp signal V_W and the burst signal V_G, in which the on time T_ON of the burst signal is fixed. The off time T_OFF of the burst signal V_G is modulated by the feedback signal.

[0044] FIG. 11 shows the control circuit 600. It includes a T flip-flop 610, D flip-flops 620, 650, AND gates 630, 640, 660 and inverters 611, 615. The oscillation signal PLS is connected to set the D flip-flop 620 and trigger the T flip-flop 610 via an inverter 611. Through an inverter 615, the oscillation signal PLS is further connected to AND gates 630 and 640 to limit the maximum on time of switching signals OH and OL. The T flip-flop 610 alternately generates an output signal S_H or an output signal S_L in response to the oscillation signal PLS. The output signal S_H and the output signal S_L are connected to AND gates 630 and 640 respectively. The AND gate 660 is coupled to receive the oscillation signal PLS and the output signal S_H. The output of the AND gate 660 is connected to set the D flip-flop 650. The D-input of the D flip-flop 650 is connected to the burst signal V_G. The output of the D flip-flop 650 is connected to the D-input of the D flip-flop 620. The output of the D flip-flop 620 is connected to AND gates 630 and 640. The output of the AND gate 630 generates the switching signal OH. The output of the AND gate 640 generates the switching signal OL. The enable signal ENB is coupled to reset the D flip-flop 620 for turning off switching signals OH and OL when the enable signal ENB is disabled. The burst signal V_G is utilized to enable or disable of switching signals OH, OL, which is synchronized with the oscillation signal PLS.

[0045] It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the present invention without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the present invention cover modifications and variations of this invention provided they fall within the scope of the following claims and their equivalents.