Non-isolated power conversion system having multiple switching power converters

[0001] 1. Field of the Invention

[0002] This invention generally relates to power conversion systems, more particularly to non-isolated power conversion systems that use multiple switching power converters.

[0003] 2. Description of the Prior Art

[0004] Isolated and non-isolated power conversion systems are known. Isolated power supplies generally use a transformer for isolating an input power stage from an output power stage through primary and secondary windings. Non-isolated power conversion systems usually use cascaded switching power converter stages associated with one or more switching cycles.

[0005] Known non-isolated power converters have been used in AC-DC, AC-AC, DC-AC, and DC-DC applications. Examples of such converters include buck converters, boost converter and buck-boost converters that can be implemented using various switching power conversion topologies. In such topologies, the input/output conversion ratios are determined according to duty cycles associated with the switching cycles. For example, a boost converter is a step-up power converter having a voltage conversion ratio that is greater than 1. On the other hand, a buck converter is a step-down converter having a voltage conversion ratio that is less than 1. In other words, the input voltage of the boost converter is always less than or equal to the output voltage, whereas, the input voltage of the buck converter is always greater than or equal to the output voltage.

[0006] The buck converter topology has been extensively used in various DC-DC applications. In fact, the non-isolated voltage-regulation modules (VRM) of today's microprocessor power supplies are almost exclusively implemented with the buck topology. FIG. 1 shows a prior art buck converter that uses a single-inductor and a switching stage to provide an output voltage that is less than the input voltage. In this prior art converter, the output-to-input conversion ratio is equal to the duty cycle associated with the switching cycle.

[0007] This prior art buck converter and its known variations exhibit satisfactory performance in low-current applications. In high-current applications, however, it may be desirable to implement a multi-stage buck converter topology that comprises multiple switching stages and inductors. One such multi-stage buck converter used in high-current applications is shown in FIG. 2. Also, it is known to use multi-stage buck converters in low output voltages applications in order to improve conversion efficiency when conduction loss under a single-stage buck converter topology is severely degraded.

[0008] The buck converter of FIG. 2 uses buck topology in parallel. This topology is often used in high-current VRM applications to reduce current stress by operating more than one buck converter in parallel. The switching instances of each switch are interleaved, i.e., phase shifted, for 180 degrees. With such interleaving, the output current ripple is reduced and, consequently, the size of the output filter capacitor is minimized. Because the duty cycle of the conventional buck converter is proportional to the conversion ratio of input/output voltage in applications that use high switching frequencies for providing high conversion ratios, the turn-on periods of the switches are extremely short. Consequently, extremely narrow switch activation pulses are necessary for maintaining very short duty cycles. Generating very narrow turn-on switch activation control signals, however, is difficult because of parasitic components that are associated with the switching devices and the switch activation circuit.

[0009] In addition, a conventional buck converter in applications that require a high-voltage conversion ratio suffers from a serious efficiency degradation. This is because the blocking voltage of the switches in a conventional buck converter is equal to its input voltage. Thus, the voltage rating of the switching devices should be higher than the input voltage. Usually, high-voltage switching devices are more expensive and have greater conduction losses in comparison with low-voltage-rated switching devices. The efficiency of the conventional buck converter is further degraded by a severe switching loss. This is because the switching loss is approximately proportional to the square of the voltage across the switch during the instances when the switch is turned on and turned off.

[0010] Other known prior art approach to non-isolated power converters used in applications that require delivering high-voltage output from low-voltage input is a boost converter. Because the duty cycle of the conventional boost converter should be maximized to provide a very large conversion ratio of input/output voltage, the turn-on periods of the switches are extremely long. Consequently, extremely long switch conduction period increases conduction losses and lowers converter efficiency. Therefore, there exists a need for a power conversion system that includes multiple power converters to provide efficient power conversion, even at high conversion ratios.

[0011] The present invention discloses different embodiments of non-isolated multi-stage switching power converters that provide high conversion ratios in response to the duty cycle associated with a switching cycle of cascaded power converters. In one embodiment, the power converter is a step-down power converter that provides regulated low-voltage output at an output stage from high-voltage input at an input stage. The step-down power conversion system provides a power conversion ratio less than or equal to one. According to these embodiments, the step-down power conversion system of the invention operates with larger duty cycle that produces wider switch activation control signals applied to switches that lower voltage stresses and reduce switching losses.

[0012] In another embodiment, the power conversion system of the present invention is a step-up power conversion system that provides regulated high-voltage output at an output stage from a low-voltage input at an input stage. The step-up power conversion system provides a power conversion ratio greater than or equal to one. According to these embodiments, the step-up power conversion system of the invention operates with smaller duty cycle that reduces switch conduction losses.

[0013] Briefly, according to the present invention, a non-isolated power conversion system has an input stage and an output stage. A plurality of cascaded switching power converter stages are coupled between the input stage and the output stage. Each of the plurality of switching power converter stages has at least one switch that is activated in accordance with a duty cycle associated with a switching cycle. At least one energy storage device temporarily stores energy that is proportional to the duty cycle during the switching cycle for delivery to the output stage.

[0014] According to some of the more detailed features of the invention, the duty cycle is phase-shifted between corresponding switches of the plurality of switching power converter stages. According to another features of the invention, at least one energy storage device is coupled to adjacent switching power converter stages. Also, the energy storage device is a blocking capacitor.

[0015] According to other more detailed features of the invention, a junction point is between corresponding switches on adjacent switching power converter stages. Additionally, the junction point connects to the output stage through at least one energy storage device and an inductor. Also, each of the plurality of switching power converter stages have an inductor and a rectifier. The rectifier may be a diode rectifier or a synchronized rectifier switch.

[0016] According to additional features of the invention, at least one energy storage device is charged to provide the same volt-second product across the inductor for each of the plurality of switching power converter stages. Another feature of the present invention places a junction point that connects diode rectifiers from adjacent cascaded switching power converter stages, the junction point is connected to at least one inductor through the at least one energy storage device. Also, multiple pairs of switches, inductors, and energy storage devices are arranged in parallel for each of the plurality of switching power converter stages.

[0017] According to further additional features of the invention, activation of the at least one switch is at least one of turning-on or turning-off of the switch. A further embodiment of the present invention has the output device as a capacitor. Also, the energy storage device is charged to a fraction of an input voltage from the input stage.

[0018] According to yet another more detailed feature of the invention, A power conversion ratio of the plurality of switching power converter stages can be greater than one or alternatively less than or equal to one. A voltage of the energy storage device can be related to an input voltage from the input stage and to the duty cycle. Alternatively, a voltage of the energy storage device can be related to an output voltage of the output stage and to the duty cycle. Also, a plurality of energy storage devices may be arranged in parallel.

[0019] In yet further detailed features of the invention, the circuits of this invention can be implemented in a variety of ways. Specifically, multiple pairs of switches, diode rectifiers, and capacitors can be connected in parallel to reduce the current stresses in the power conversion system. Also, the power conversion system could be connected in parallel for high current applications.

[0052] The power conversion system of the present invention is described in terms of multiple switching power converter stages that provide conversion ratios in response to a duty cycle associated with a switching cycle. The multiple switching power converter stages are cascaded in accordance with the present invention.

[0053] In one embodiment, the power conversion system of the invention comprises a step-down power conversion system that provides regulated low-voltage output at an output stage from a high-voltage input at an input stage. The step-down power conversion system provides a power conversion ratio less than or equal to one. According to these embodiments, the power conversion system of the invention operates with larger duty cycle that produces wide switch activation control signals applied to switches that lower voltage stresses and reduce switching losses.

[0054] In another embodiment, the power conversion system of the present invention is a step-up power conversion system that provides regulated high-voltage output at an output stage from a low-voltage input at an input stage. The step-up power conversion system provides a power conversion ratio greater than or equal to one. According to these embodiments, the step-up power conversion system of the invention operates with smaller duty cycle that reduces switch conduction losses.

[0055] FIG. 3 shows the schematic diagram of a non-isolated step-down (or buck) 2-stage power conversion system 300 according to an exemplary embodiment of the present invention. The power conversion system 300 comprises switches S₁ and S₂, inductors L₁ and L₂, a energy storage device (or blocking capacitor) C_B, diode rectifiers D₁ and D₂, and output capacitor C_F connected across load R.

[0056] In this exemplary embodiment, a first switching power converter stage comprises switch S₁, output inductor L₁ and output diode rectifier D₁. As described later in detail, this switching power converter stage, during one switching cycle, temporarily provides a voltage across output inductor L₁ and output capacitor C_F that is equal to the blocking capacitor voltage V_CB. A second switching power converter stage comprises switch S₂, output inductor L₂ and output diode rectifier D₂. During another phase shifted switching cycle, this switching power converter stage temporarily provides a voltage across output inductor L₂ and output capacitor C_F that is equal to the input stage voltage V_IN minus the blocking capacitor voltage V_CB.

[0057] As shown in FIG. 3, switches S₁ and S₂ are on cascaded switching power converter stages that can be viewed as connected in series between the input stage and the output stage 310. The input stage comprises a voltage source V_IN, and the output stage comprises an output capacitor C_F. Between the series connected switches S₁ and S₂ is a junction point that connects to the output stage through blocking capacitor C_B and inductor L₂. Switching power converter stages that share a junction point are adjacent, and two or more switching power converter stages coupled together are cascaded.

[0058] As depicted in FIG. 3, the energy storage device (blocking capacitor) C_B is coupled to the first and second switching power converter stages. Blocking capacitor C_B acts as a voltage divider to divide the input voltage V_IN. In this arrangement, the blocking capacitor C_B temporarily stores energy during a switching cycle for delivery to the output stage such that the temporarily stored energy is proportional to the duty cycle. During steady state, the voltage across the blocking capacitor is the output voltage V_o divided by the duty cycle D.

[0059] More specifically, during the steady state operation, the energy storage device (blocking capacitor) C_B is temporarily charged with a fraction of the input voltage V_IN to provide the same volt-second product across each individual inductor L₁ and L₂ for each of the switching power converter stages. Consequently, the duty cycle of the power conversion system 300 can be increased, relative to conventional buck converters, in order to provide the same volt-second product. As a result, the step-down power converter of the invention can be designed to operate with relatively wide-pulse turn-on switch activation control signals, even at high switching frequencies. Moreover, because the blocking capacitor voltage V_CB across the switches S₁ and S₂ is only a fraction of the input voltage V_IN, more efficient low-voltage rated semiconductor devices may be employed to reduce conduction losses. A control circuit 320 generates switch activation control signals to control the switches S₁ and S₂ in a phase-shifted manner, as described below.

[0060] FIG. 4 shows a simplified circuit diagram of the 2-stage non-isolated power conversion system of FIG. 3. In the simplified circuit diagram, capacitor C_B has been idealized as a voltage source V_CB, and output capacitor C_F has been idealized as a voltage source V_o. The intersection of switch S₁, inductor L₁, and diode rectifier D₁ is interconnection A. The intersection of capacitor C_B, inductor L₂, and diode rectifier D₂ is interconnection B. In the simplified circuit, it is assumed that output capacitor C_F and blocking capacitor C_B are large enough so that the voltage ripples across them are small compared to their DC voltages. Moreover, in this analysis, it is also assumed that all semiconductor components are ideal, i.e., they represent zero impedances in the ‘on’ state and infinite impedances in the ‘off’ state. Finally, it is assumed that the conducting duty cycle periods of switches S₁ and S₂ are identical.

[0061] FIG. 5 shows the topological stages of the circuit diagram of FIG. 4 during a switching cycle, whereas FIG. 6 shows its key waveforms. The reference directions of currents and voltages plotted in FIG. 6 are shown in FIG. 4 to indicate the polarity of the waveforms. As can be seen from the timing diagrams of FIG. 6(a), the activation of switches S₁ and S₂ is phase-shifted, i.e., delayed, on adjacent switching power converter stages according to switch activation control signals. The turn-on moment of switch S₁ is phase-shifted 180 degrees from the turn-on moment of switch S₂. The switch activation control signals either close (turn-on) or-open (turn-off) the switch receiving the signal. The conducting time of each switch represents duty cycle D of the converter, and the switches operate with a switching cycle T_S, as illustrated in FIG. 6(a). It would be appreciated that the duty cycle D and the switching cycle T_S have a fractional relationship with each other, and the duration of the duty cycle is within the duration of the switching cycle. It should be noted that FIGS. 5 and 6 describe the power conversion system of the invention for operation with a duty cycle of less than 0.5, i.e., D<0.5.

[0062] FIGS. 6(a)–6(f) depict the activation, voltage, and current waveforms for the circuit components in the switching power converter stages according to the switch activation control signals for switches S₁ and S₂. FIG. 6(b) depicts the voltage waveform V_S2 across switch S₂, FIG. 6(c) depicts the voltage waveform V_S1 across switch S₁, FIG. 6(d) illustrates current waveform i_CB across capacitor C_B, FIG. 6(e) depicts the current waveforms i_S1 and i_S2 across switches S₁ and S₂, respectively, FIG. 6(f) illustrates the current waveforms i_L1 and i_L2 across inductors L₁ and L₂, respectively, FIG. 6(g) depicts the current waveform i_o across output capacitor C_F, FIG. 6(h) illustrates voltage waveform V_AB across interconnections A and B, FIG. 6(i) depicts the voltage waveforms V_A and V_B measured at interconnections A and B, respectively, FIG. 6(j) illustrates the current waveforms i_D1 and i_D2 across diode rectifiers D₁ and D₂, respectively.

[0063] It should be understood that the FIGS. 6(a)–(j) depict idealized waveforms, and that the actual waveforms would contain noise and other non-ideal component characteristics that would affect the appearance of the measured waveforms, as one of ordinary skill in the art would understand. Accordingly, the circuit components of the present invention behave similarly to the waveforms depicted, but include inherent non-idealities. The current and voltage waveforms are for illustrative purposes only, and are not intended to depict actual voltage and current waveforms measured from the circuit.

[0064] In steady state operation, with a duty cycle D of less than 0.5, the voltage across blocking capacitor C_B is one-half of input voltage V_IN, i.e., V_CB=V_IN/2. FIG. 5(a) illustrates the time interval when switch S₂ is on, i.e., during the time interval T₀–T₁ in FIG. 6. In this interval, the input current flows through blocking switch S₂, capacitor C_B, and inductor L₂ into output voltage source V_O, while the current in inductor L₁ flows through diode rectifier D₁ into output voltage source V_O. The input voltage V_IN is divided between the blocking capacitor C_B and the output stage. The voltage across output inductor L₂ and output capacitor C_F at the output stage is the difference between the input voltage V_IN and the voltage V_CB across the blocking capacitor C_B. During the time interval T₀–T₁, inductor current i_L2 is increasing at the rate

[0065] $\begin{array}{cc}\frac{d{i}_{L2}}{dt}=\frac{V_{\mathrm{IN}}-V_C-V_0}{L_2}=\frac{\frac{V_{\mathrm{IN}}}{2}-V_0}{L_2},& \left(1\right)\end{array}$
while inductor current i_L1 is decreasing at the rate

[0066] $\begin{array}{cc}\frac{d{i}_{L1}}{dt}=\frac{-V_0}{L_1}& \left(2\right)\end{array}$
When at t=T₁, switch S₂ is turned off, inductor current i_L2 is diverted from switch S₂ to diode rectifier D₂, as shown in FIG. 5(b), and the energy stored in inductor L₂ starts to discharge into voltage source V_O. During the time interval T₁–T₂, current i_L2 decreases at the rate

[0067] $\begin{array}{cc}\frac{d{i}_{L2}}{dt}=\frac{-V_0}{L_2},& \left(3\right)\end{array}$
while inductor current i_L1 continues to decrease at the rate given in Eq. (2).

[0068] When at t=T₂, switch S₁ is turned on, the circuit enters the topological stage shown in FIG. 5(c). During the time interval T₂–T₃ in FIG. 6, inductor current i_L1 flows through diode rectifier D₂, capacitor C_B, switch S₁ and inductor L₁ into the output while the current in inductor L₂ flows through diode rectifier D₂ into the output, as shown in FIG. 5(c). The voltage across output inductor L₁ and output capacitor C_F is equal to the voltage V_CB of the blocking capacitor C_B. During the time interval T₂–T₃, inductor current i_L1 is increasing at the rate

[0069] $\begin{array}{cc}\frac{d{i}_{L1}}{dt}=\frac{V_C-V_0}{L_1}=\frac{\frac{V_{\mathrm{IN}}}{2}-V_0}{L_1},& \left(4\right)\end{array}$
while inductor current i_L2 is decreasing at the rate given in Eq. (3).

[0070] When at t=T₃, switch S₁ is turned off, the circuit enters the topological stage shown in FIG. 5(d), which is identical to the topological stage in FIG. 5(b). During this stage, both switches are off and both inductor currents i_L1 and i_L2 decrease at the same rates given by Eqs. (2) and (3). The circuit enters a new switching cycle at t=T₄ when switch S₂ is turned on again.

[0071] The voltage conversion ratio of the circuit can be calculated from the volt-second balance of the output inductors. From Eqs. (1) and (3), the volt-second balance equation for L₂ is

[0072] $\begin{array}{cc}\left(\frac{V_{\mathrm{IN}}}{2}-V_0\right){\mathrm{DT}}_S=V_0\left(T_S-{\mathrm{DT}}_S\right)\mathrm{so}\mathrm{that}& \left(5\right)\\ \frac{V_0}{V_{\mathrm{IN}}}=\frac{D}{2}\mathrm{for}D\le 0.5.& \left(6\right)\end{array}$

[0073] As can be seen from Eq. (6), the output voltage of the power conversion system in FIG. 3 is one-half of the output voltage of a conventional buck converter when they operate at the same duty cycle. This high conversion ratio makes the converter suitable for applications with a high difference between the input and output voltages, since the power conversion systems do not have to work with narrow switch activation control signals.

[0074] It should be noted that the switch voltages shown in FIGS. 6(b) and (c) are charging only to one-half of input voltage V_IN during the instances when switches S₁ and S₂ are turned on and turned off. Because a switching loss is approximately proportional to the square of the voltage charge across a switch during the switch turn on and turn off time, the switching loss of the power conversion system is approximately one-quarter of that of the conventional buck converter. In addition, because a low-voltage stress on the switches, the converter can employ more efficient low-voltage-rated semiconductor switches.

[0075] For the operation with a duty cycle greater than 0.5, i.e., D>0.5, the voltage conversion ratio of the circuit in FIG. 3 becomes

[0076] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=D^2.& \left(7\right)\end{array}$
FIG. 7 shows the timing diagrams for the current and voltage waveforms of the switching power converter stage when it operates with duty cycle D greater than 0.5 as switches S₁ and S₂ operate according to the switch activation control signals. FIG. 7(a) depicts the activation waveform of switch S₂, FIG. 7(b) depicts the activation waveform of switch S₁, FIG. 7(c) depicts the voltage waveform V_S2 across switch S₂, FIG. 7(d) depicts the voltage waveform V_S1 across switch S₁, FIG. 7(e) depicts the current waveform i_S2 across switch S₂, FIG. 7(f) depicts the current waveform i_S1 across switch S₁, FIG. 7(g) depicts the current waveforms i_L1 and i_L2 across inductors L₁ and L₂, respectively, FIG. 7(h) depicts the voltage waveform V_A measured at interconnection A, FIG. 7(i) depicts the voltage waveform V_B measured at interconnection B, FIG. 7(j) depicts the current waveform i_D1 across diode rectifier D₁, and FIG. 7(k) depicts the current waveform i_D2 across diode rectifier D₂.

[0077] In another embodiment, the non-isolated power conversion system of the invention can be implemented with synchronized rectifier switches instead of diode rectifiers. As shown in FIG. 8, in this implementation, diode rectifiers D₁ and D₂ of the converter in FIG. 3 are replaced by synchronized rectifier switches S_R1 and S_R2. Since the maximum voltages of switches S_R1 and S_R2 are one half of input voltage V_IN, synchronized rectifier switches S_R1 and S_R2 can also be implemented with more efficient low-voltage rated semiconductor devices.

[0078] FIG. 9 shows the timing diagrams of the switches S₁ and S₂ and synchronized rectifier switches S_R1 and S_R2 of the switching power converter stage in FIG. 8 according to switch activation control signals. Since the activation of switches S₁ and S₂ is phase-shifted on the adjacent switching power converter stages, the output current ripple of the power conversion system is reduced, and the size of the output filter capacitor is minimized. FIG. 9(a) depicts the activation waveform of switch S₂, FIG. 9(b) depicts the activation waveform of switch S₁, FIG. 9(c) depicts the activation waveform of synchronized rectifier switch S_R2, and FIG. 9(d) depicts the activation waveform of synchronized rectifier switch S_R1. It would be appreciated that switch S₂ and synchronized rectifier switch S_R2 are conducting at complementary intervals. The switch S₂ conducts during the duty cycle D of its switching power converter stage, whereas the synchronized rectifier switch S_R2 is conducting during the remainder to the switching cycle T_S. The same relationship applies to switch S₁ and synchronized rectifier switch S_R1.

[0079] FIG. 10 shows a 3-stage non-isolated cascaded switching power converter according to the present invention. Switches S₁, S₂, and S₃ are activated at phase-shifted, i.e., delayed, intervals of 120 degrees, as will be described below in FIG. 11. As shown in FIG. 10, the power conversion system employs three switches, S₁, S₂, and S₃, connected in series between an input stage and inductors L₁, L₂, L₃, respectively, to an output stage. The output stage is further coupled to a load R. In this embodiment, the input stage comprises a voltage source V_IN, and the output stage comprises an output capacitor C_F. Between switches S₁ and S₂ is a first junction point connected to inductor L₂ through an energy storage device (blocking capacitor) C_B2. The voltage across capacitor C_B2 is the output voltage V_o divided by the duty cycle D. Between switches S₂ and S₃ is a second junction point connected to inductor L₃ through an energy storage device (blocking capacitor) C_B3. The voltage across capacitor C_B3 is two times the output voltage V_o divided by the duty cycle D. As a general rule for multiple step-down switching power converter stages, the voltage across the blocking capacitor coupled to a junction point is equal to the stage number of the lower switching power converter stage coupled to the junction point, multiplied by the output voltage, and divided by the duty cycle D. Therefore, the junction point between the second and third switching power converter stages is two, multiplied by output voltage V_o, and divided by duty cycle D.

[0080] In this embodiment, the first switching power converter stage comprises switch S₁, inductor L₁, and synchronized rectifier switch S_R1, the second switching power converter stage comprises switch S₂, inductor L₂, and synchronized rectifier switch S_R2, and the third switching power converter stage comprises switch S₃, inductor L₃, and synchronized rectifier switch S_R3. Switching power converter stages that share a junction point are adjacent switching power converter stages. In the present embodiment, the first and second switching power converter stages are adjacent, and the second and third switching power converter stages are adjacent. The first, second, and third switching power converter stages are coupled together and are a cascaded switching power converter stage.

[0081] During the steady-state operation, blocking capacitors C_B2 and C_B3 adjust their DC voltages so that they provide the same volt-second condition across inductors L₁, L₂, and L₃. The voltage conversion ratio of the circuit calculated from the volt-second balance of the inductors is

[0082] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=\frac{D}{3}& \left(8\right)\end{array}$
when it operates with duty cycle D less than one-third. As can be seen from Eq. (8), for the same duty cycle, the output voltage of the power conversion system in FIG. 10 is one-third the output voltage of the conventional buck converter. When the converter operates with a duty cycle greater than one-third, the voltage conversion ratio monotonically increases from D/3 to 1.

[0083] FIG. 11 shows the timing diagrams of switches S₁, S₂ and S₃ and synchronized rectifier switches S_R1, S_R2, and S_R3 of the respective switching power converter stages depicted in FIG. 10 according to the switch activation control signals. Switches S₁, S₂ and S₃ are activated at phase-shifted intervals of 120 degrees, wherein each switch conducts during a duty cycle that is out of phase with the other switches. Since switches S₁, S₂ and S₃ are activated in phase-shifted intervals on adjacent switching power converter stages, the output current ripple of the power conversion system is reduced, and the size of the output filter capacitor is minimized. FIG. 11(a) depicts the activation waveform of switch S₃, FIG. 11(b) depicts the activation waveform of switch S₂, FIG. 11(c) depicts the activation waveform of switch S₁, FIG. 11(d) depicts the activation waveform of synchronized rectifier switch S_R3, FIG. 11(e) depicts the activation waveform of synchronized rectifier switch S_R2, and FIG. 11(f) depicts the activation waveform of synchronized rectifier switch S_R1. It would be appreciated that switch S₃ and synchronized rectifier switch S_R3 are conducting at complementary intervals. The switch S₃ conducts during the duty cycle D of its switching power converter stage, whereas the synchronized rectifier switch S_R3 is conducting during the remainder to the switching cycle T_S. The same relationship applies to switches S₂, S₁ and synchronized rectifier switch S_R2, S_R1, respectively.

[0084] FIG. 12 shows another arrangement of a 4-stage cascaded switching power converter used in accordance with the present invention. This figure depicts an arrangement with four switching power converter stages, where each of the switching power converter stages is identified similar to the description given for FIG. 10. In the present embodiment, the circuit can be operated according to switch activation control signals depicted in FIG. 13 or those in FIG. 14.

[0085] FIG. 13 depicts the timing diagrams of switches S₁, S₂, S₃, and S₄ and synchronized rectifier switches S_R1, S_R2, S_R3 and S_R4 of respective switching power converter stages depicted in FIG. 12 according to the switching activation controls signals. Switches S₁ and S₃ are the odd switches, and switches S₂ and S₄ are the even switches because of which switching power converter stage they are located on. The duty cycles for switches on the even switching power converter stages are phase-shifted by 180 degrees relative to the odd switching power converter stages, therefore the odd and even switches are out of phase with respect to one another. Since the activation of switches S₁, S₂, S₃, and S₄ are on phase-shifted duty cycle intervals of adjacent switching power converter stages, the output current ripple of the power conversion system is reduced, and the size of the output capacitor is thereby minimized. FIG. 13(a) depicts the concurrent activation waveforms of switches S₂ and S₄, FIG. 13(b) depicts the concurrent activation waveforms of switches S₁ and S₃, FIG. 13(c) depicts the concurrent activation waveforms of synchronized rectifier switches S_R2 and S_R4, and FIG. 13(d) depicts the concurrent activation waveforms of synchronized rectifier switches S_R1 and S_R3. It would be appreciated that even switches S₂, S₄ and even synchronized rectifier switches S_R2, S_R4 are conducting at complementary intervals. The even switches S₂, S₄ conduct during the duty cycle D of their respective even switching power converter stages, whereas the even synchronized rectifier switches S_R2, S_R4 are conducting during the remainder to the switching cycle T_S. The same relationship applies to switches S₁, S₃ and synchronized rectifier switches S_R1, S_R3 of the odd switching power converter stages.

[0086] FIG. 14 depicts an alternate scheme for the switching activation controls signals. This figure depicts the timing diagrams of switches S₁, S₂, S₃, and S₄ and synchronized rectifier switches S_R1, S_R2, S_R3 and S_R4 of the respective switching power converter stages depicted in FIG. 12 according to the switch activation control signals. In this embodiment, respective duty cycles of the switches on adjacent switching power converter stages are phase-shifted by T_S/4, wherein each switch conducts during a phase-shifted duty cycle that is out of phase with all of the other switches. The phase-shifted intervals are separated by 90 degrees. Since the activation of switches S₁, S₂, S₃, and S₄ occur on phase-shifted intervals for the adjacent switching power converter stages, the output current ripple of the power conversion system is reduced, and the size of the output capacitor is minimized. FIG. 14(a) depicts the activation waveform of switch S₄, FIG. 14(b) depicts the activation waveform of switch S₃, FIG. 14(c) depicts the activation waveform of switch S₂, FIG. 14(d) depicts the activation waveform of switch S₁, FIG. 14(e) depicts the activation waveform of synchronized rectifier switch S_R4, FIG. 14(f) depicts the activation waveform of synchronized rectifier switch S_R3, FIG. 14(g) depicts the activation waveform of synchronized rectifier switch S_R2, and FIG. 14(h) depicts the activation waveform of synchronized rectifier switch S_R1. It would be appreciated that switch S₄ and synchronized rectifier switch S_R4 are conducting at complementary intervals. The switch S₄ conducts during the duty cycle of its switching power converter stage, whereas the synchronized rectifier switch S_R4 conducts during the remainder to the switching cycle T_S. The same relationship applies to the other switches and synchronized rectifier switches on the remaining switching power converter stages, respectively.

[0087] The voltage-conversion ratio of the circuit in FIG. 12 is given by

[0088] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=\frac{D}{4}& \left(9\right)\end{array}$
when the converter operates with duty cycle D less than one-half for the driving scheme in FIG. 13, and with duty cycle D less than one-fourth for the driving scheme in FIG. 14. When the converter operates with overlapped switch activation control signals, i.e., with a duty cycle greater than one-half for the driving scheme in FIG. 13, or with a duty cycle greater than one-fourth for the driving scheme in FIG. 14, the voltage conversion ratio monotonically changes from D/4 to 1.

[0089] FIG. 15 shows an arrangement where n-switching power converter stages are used in accordance with the present invention, where n is any integer. The voltage conversion ratio of the circuit is

[0090] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=\frac{D}{n}& \left(10\right)\end{array}$
for the operation where the switch activation control signals of the switches on adjacent switching power converters are not overlapped, i.e., for D<1/n. If the switch activation control signals of the switches on adjacent switching power converter stages are overlapped (D>1/n), the voltage conversion ratio of the converter monotonically increases from D/n to 1 as duty cycle D increases. Similar to FIG. 13, the duty cycles of corresponding switches of the even switching power converter stages of the n-switch power converter may be activated simultaneously out of phase with the odd switching power converter stages, where the even converters are activated at phase-shifted intervals relative to the odd converters. Otherwise, each switching power converter stage may be activated at phase-shifted intervals of T_S/n, where all of the switches are out of phase, similar to the description of FIG. 14.

[0091] According to another embodiment of the invention, a step-up power conversion system provides regulated high-voltage outputs from low-voltage inputs. The step-up power conversion system employs multiple switching power converter stages with diode rectifiers connected in series between an output stage and an input stage. In this embodiment, the input stage comprises a voltage source, and the output stage comprises an output capacitor. A switching power converter stage comprises a switch, an inductor, and a diode rectifier. Between each pair of series-connected diode rectifiers is a junction point that is connected to an inductor through one or more energy storage devices. The switching power converter stages that share a junction point are adjacent, and two or more switching power converter stages coupled together are cascaded. The energy storage devices may be blocking capacitors.

[0092] During the steady state operation, these blocking capacitors are temporarily charged to provide the same volt-second product across each individual inductor of the switching power converter stage. It should be noted that the step-up power conversion system of the invention does not require an extremely large duty cycle to provide very high-voltage output in comparison with its conventional boost converter counterpart. In addition, the voltage stress on most of the semiconductor components in the power conversion system of the present invention is also reduced, since it is equal to the output voltage divided by the number of inductors. As such, the step-up power conversion system of the present invention operates with a reduced switching loss, and can employ more efficient low-voltage rated semiconductor devices.

[0093] FIGS. 16–23 show examples of non-isolated step-up cascaded switching power converter stage topologies of this invention and their corresponding voltage, current, and switch activation control signal waveforms. These step-up power converters are suitable for applications where high-voltage output is generated from low-voltage input.

[0094] FIG. 16 shows a 2-stage non-isolated two-switch, two-inductor, cascaded step-up switching power converter of this invention. The depicted power conversion system employs two diode rectifiers D₁ and D₂, connected in series with inductors L₁ and L₂, respectively, to the output stage. Between diode rectifiers D₁ and D₂ is a junction point connected to input inductor L₁ through blocking capacitor C_B. In steady state, the voltage across the blocking capacitor C_B is proportional to the input voltage V_IN and the duty cycle D. In this embodiment, a first step-up switching power converter stage comprises switch S₁, inductor L₁, and diode rectifier D₁, and a second step-up switching power converter stage comprises switch S₂, inductor L₂, and diode rectifier D₂. Switching power converter stages that share a junction point are adjacent switching power converters, therefore the first and second switching power converter stages are adjacent switching power converters. The first and second switching power converter stages are also cascaded because they are interconnected. The input stage applies voltage V_IN to the cascaded switching power converter stages, which is output at the output stage.

[0095] FIG. 17 depicts the activation waveforms for switches S₁ and S₂ of the corresponding switching power converter stages according to switching control activation signals in FIG. 16. FIG. 17(a) illustrates the activation waveform of switch S₁, and FIG. 17(b) illustrates the activation waveform of switch S₂. Switches S₁ and S₂ operate according to a duty cycle D, which has a fractional relationship with the switching cycle T_S. It is noted that the duty cycle for switch S₁ is phase-shifted, i.e., delayed, by 180 degrees relative to switch S₂. In contrast to the step-down power converter, the duty cycle of the step-up converter is overlapped (D>0.50), wherein both switches are conducting during the same interval during the switch transitions of switching cycle T_S,. This implies both of the switches are conducting during the transitions. As illustrated, switch S₂ conducts during the period when switch S₁ is turned off, and switch S₁ conducts during the period when switch S₂ is turned off. As a result, the energy in inductor L₁ stored during the time when switch S₁ turns on is delivered to blocking capacitor C_B when switch S₁ turns off.

[0096] In the steady state, the voltage at the output stage across blocking capacitor C_B can be found from the volt-second balance equation for L₁, which is
V_INDT_S=(V_CB−V_IN)(T_S−DT_S)  (11)
From Eq. (11), voltage V_CB across blocking capacitor C_B is

[0097] $\begin{array}{cc}{V}_{\mathrm{CB}}=\frac{V_{\mathrm{IN}}}{1-D}.& \left(12\right)\end{array}$
The voltage conversion ratio of the power conversion system can be calculated from the volt-second balance of input inductor L₂ and Eq. (12). Since the volt-second balance equation for L₂ is
V_INDT_S=(V_O−V_CB−V_IN)(T_S−DT_S),  (13)
from Eqs. (12) and (13), the voltage conversion ratio of the converter in FIG. 16 is given by

[0098] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=\frac{2}{1-D}& \left(14\right)\end{array}$
when the converter operates with duty cycle D greater than 0.5. As can be seen from Eq. (14), the output voltage of the power conversion system is twice the output voltage of the conventional boost converter when they operate with the same duty cycle. This high conversion ratio makes the switching power conversion system suitable for applications with a high difference between the input and output voltages, since the power conversion systems do not require extremely large switch activation control signals to deliver a high voltage output. It should be noted that the converter has a non-linear voltage conversion ratio that depends on duty cycle D. When the converter operates with duty cycle D at less than 0.5, the voltage conversion ratio is

[0099] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}={\left(\frac{1}{1-D}\right)}^2.& \left(15\right)\end{array}$
As can be seen from FIG. 16, the blocking voltage of switch S₁ and diode rectifier D₁ is equal to the voltage across blocking capacitor C_B, which is one-half of output voltage V_O as it can be derived from Eqs. (12) and (14). The blocking voltage of switch S₂ and diode rectifier D₂ is equal to the voltage difference between output voltage V_O and blocking capacitor voltage V_CB, which is also one-half of output voltage V_O. As a result, the voltage stress of most semiconductor components is only one-half of the output voltage, which makes the switching power converter stage operate with small switching losses and employ efficient low-voltage rated semiconductor devices. In this way, the blocking capacitor C_B temporarily stores energy during a switching cycle for delivery to the output stage such that the temporarily stored energy is proportional to the duty cycle. Similar to the step-up power converter, the duty cycle and the switching cycle have a fractional relationship with each other. The duration of the duty cycle D will always be within the duration of the switching cycle T_S.

[0100] FIG. 18 shows a 3-stage non-isolated three-switch, three-inductor, cascaded step-up switching power converter stage of this invention. The activation of switches S₁, S₂, and S₃ is phase-shifted between the corresponding switching power converter stages, as illustrated and will be described in FIG. 19. As shown in FIG. 18, the power conversion system employs three diode rectifiers, D₁, D₂, and D₃, connected in series between input inductor L₁ and output capacitor C_F. Between diode rectifiers D₂ and D₃ is a first junction point that connects to input inductor L₃ through blocking capacitor C_B3. Between diode rectifiers D₁ and D₂ is a second junction point that connects to input inductor L₂ through blocking capacitor C_B2.

[0101] In this embodiment, the first switching power converter stage comprises switch S₁, inductor L₁, and diode rectifier D₁, the second switching power converter stage comprises switch S₂, inductor L₂, and diode rectifier D₂, and the third switching power converter stage comprises switch S₃, inductor L₃, and diode rectifier D₃. Switching power converter stages that share a junction point are adjacent switching power converters, and since the three switching power converter stages are coupled together, they are cascaded. In steady state, the voltage across a blocking capacitor is determined by the switching power converter stages where the junction point is connected to, the input voltage, and the duty cycle. As a general rule, the number of the lower numbered stage for the adjacent switching power converter stages is multiplied by the input voltage V_IN, and is divided by one minus the duty cycle. Accordingly, for blocking capacitor C_B3 attached to the junction point between the second and third switching power converter stages, the voltage V_CB3 across capacitor C_B3 is two multiplied by input voltage V_IN, and divided by one minus the duty cycle (1−D).

[0102] The activation of switches is phase-shifted, i.e., delayed, 120 degrees in the corresponding switching power converter stages. During steady state operation, blocking capacitors C_B2 and C_B3 maintain DC voltages to provide the same volt-second condition across inductors L₁, L₂, and L₃. The voltage conversion ratio of the circuit can be calculated from the volt-second balance of each inductor, which is

[0103] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=\frac{3}{1-D}& \left(16\right)\end{array}$
when it operates with duty cycle D greater than one-third. As can be seen from Eq. (16), the output voltage of the power conversion system in FIG. 18 is three times the output voltage of a conventional boost converter when they operate with the same duty cycle. When the converter operates with a duty cycle of less than one-third, the voltage conversion ratio monotonically increases from 1 to 3/(1−D).

[0104] FIG. 19 depicts the timing diagrams of switches S₁, S₂, and S₃ of the corresponding switching power converter stages according to switching control activation signals in the circuit of FIG. 18. FIG. 19(a) illustrates the activation waveform of switch S₃, FIG. 19(b) illustrates the activation waveform of switch S₂, and FIG. 19(c) illustrates the activation waveform of switch S₁. Switches S₁, S₂, and S₃ operate with duty cycle D, which has a fractional relationship with the switching cycle T_S. It is noted that the duty cycles for the respective switches are phase-shifted, i.e., delayed, by 120 degrees relative to one another, which makes all of the switches out of phase with one another. Similar to FIG. 17, the duty cycles overlap indicating that at least one switch is conducting at all times.

[0105] FIG. 20 shows a 4-stage non-isolated four-switch, four-inductor, cascaded step-up switching power converter stage of this invention. In this figure, four step-up switching power converter stages are depicted, and the converters are identified similar to the step-up switching power converter stages of FIG. 18. The voltage conversion ratio of the circuit, which is controlled by the switch activation control signals shown in FIG. 21, is

[0106] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=\frac{4}{1-D}& \left(17\right)\end{array}$
when the converter operates with duty cycle D larger than one-half. When the converter operates with a duty cycle of less than one-half, the voltage conversion ratio monotonically changes from 1 to 4/(1−D). The converter that operates with the interleaved switch activation control signals, shown in FIG. 22, has the same voltage conversion ratio described as in Eq. (17) if it operates with duty cycle D greater than one-fourth.

[0107] When the converter operates with a duty cycle of less than one-fourth, the voltage conversion ratio monotonically increases from 1 to 4/(1−D) as duty cycle D increases from zero to one-fourth. The corresponding switches of the switching power converter stages from FIGS. 21 and 22 may be activated similar to the description from FIGS. 13 and 14, respectively. FIG. 21 depicts the corresponding switches of the even switching power converter stages being activated simultaneously out of phase with the activation of the odd switching power converter stages. In this embodiment, the odd converters are phase-shifted 180 degrees relative to the even converters. FIG. 22 depicts each switch of the corresponding switching power converter stage being activated at intervals phase-shifted by T_S/4, relative to the other switches.

[0108] FIG. 23 shows a n-stage non-isolated n-switch, n-inductor, cascaded step-up power converter of this invention where n is any integer. The voltage conversion ratio of the circuit is

[0109] $\begin{array}{cc}\frac{V_0}{V_{\mathrm{IN}}}=\frac{n}{1-D}& \left(18\right)\end{array}$
if the switch activation control signals for the switches of adjacent switching power converters are overlapped. If the switch activation control signals for the switches of adjacent switching power converters are not overlapped when duty cycle D of the converter is small, the voltage conversion ratio of the converter monotonically increases from 1 to n/(1−D) as duty cycle D increases from zero.

[0110] This embodiment of the present invention can be implemented in a variety of ways. Specifically, multiple pairs of switches, diode rectifiers, and capacitors can be connected in parallel to reduce the current stresses in the power conversion system. Also, the power conversion system could be connected in parallel for high current applications.

[0111] FIGS. 24–29 show non-isolated converters that have the same voltage conversion ratio as that of the converters shown in FIGS. 3 and 8 of this invention. The converters shown in FIGS. 24–29 have additional switches, capacitors, and inductors effectively connected in parallel to share the load current and reduce the peak current stress of the components for each of the switching power converter stages.

[0112] FIG. 24 shows a 2-stage non-isolated four-switch, two-inductor, cascaded step-down switching power converter stage of this invention. In this embodiment, the energy storage device is embodied in two blocking capacitors C₁ and C₂, placed in parallel. The first switching power converter stage comprises switch S₁, switch S₃, inductor L₁, blocking capacitor C₁ and synchronized rectifier switches S_R1, the second switching power converter stage comprises switch S₂, switch S₄, inductor L₂, blocking capacitor C₂, and synchronized rectifier switches S_R2. The first and second switching power converter stages are coupled between the input stage and the output stage. In this embodiment, the input stage comprises a voltage source, and the output stage comprises an output capacitor.

[0113] FIG. 25 depicts the timing diagrams for switches S₁, S₂, S₃, and S₄ and synchronized rectifier switches S_R1 and S_R2 of respective switching power converter stages depicted in FIG. 24 according to the switching activation controls signals. As depicted, switches S₁ and S₃ from the first switching power converter stage conduct during the same duty cycle, as do switches S₂ and S₄ of the second switching power converter stage. The respective sets of switches on the different switching power converter stages are activated at phase-shifted intervals of 180 degrees, wherein switches S₁ and S₃ are out of phase with switches S₂ and S₄. Because the activation of switches S₁, S₂, S₃, and S₄ is phase-shifted between the first and second sets of switching power converter stages, the output current ripple of the power conversion system is reduced, and the size of the output filter capacitor is minimized. FIG. 25(a) depicts the concurrent activation of switches S₂ and S₄, FIG. 25(b) depicts the concurrent activation of switches S₁ and S₃, FIG. 25(c) depicts the activation of synchronized rectifier switch S_R1, and FIG. 25(d) depicts the activation of synchronized rectifier switch S_R2. It would be appreciated that switches S₂, S₄ and synchronized rectifier switch S_R2 are conducting at complementary intervals. The switches S₂, S₄ conduct during the duty cycle D of the second switching power converter stage, whereas synchronized rectifier switch S_R2 are conducting during the remainder to the switching cycle T_S. A similar relationship applies to switches S₁, S₃, and synchronized rectifier switch S_R1 of the first switching power converter stage.

[0114] FIG. 26 depicts the timing diagrams for switches S₁, S₂, S₃, and S₄ and synchronized rectifier switches S_R1 and S_R2 of the respective switching power converter stages depicted in FIG. 12 according to the switch activation control signals. In this embodiment, the activation of respective switches S₁, S₂, S₃, and S₄ are phase-shifted by T_S/4 of the corresponding switching power converter stages, wherein each switch conducts during phase-shifted duty cycles that are out of phase with all of the other switches, and the phase-shifted intervals are separated by 90 degrees. Since the activation of switches S₁, S₂, S₃, and S₄ is phase-shifted on adjacent switching power converter stages, the output current ripple of the power conversion system is reduced, and the size of the output capacitor is minimized. FIG. 26(a) depicts the activation waveform of switch S₂, FIG. 26(b) depicts the activation waveform of switch S₁, FIG. 26(c) depicts the activation waveform of switch S₄, FIG. 26(d) depicts the activation waveform of switch S₃, FIG. 26(e) depicts the activation of waveform synchronized rectifier switch S_R1, and FIG. 26(f) depicts the activation waveform of synchronized rectifier switch S_R2. It would be appreciated that switches S₂, S₄ and synchronized rectifier switch S_R2 are conducting at complementary intervals. Switches S₂, S₄ are conducting at out of phase relative to one another. The switches S₂, S₄ conduct during the duty cycle of their respective switching power converter stages, whereas the synchronized rectifier switch S_R2 conducts during the remainder to the switching cycle T_S. The same relationship applies to the switches S₁, S₃ and synchronized rectifier switch S_R1 on the first switching power converter stage.

[0115] The voltage conversion ratio of the circuit in FIG. 24, which is controlled by either of the switch activation control signals shown in FIGS. 25 and 26, is equal to the voltage conversion ratio described in Eq. (6) as long as the switch activation control signals of the adjacent switches in the converter are not overlapped. If the switch activation control signals of the corresponding switches of adjacent switching power converter stages are overlapped when duty cycle D of the converter increases, the voltage conversion ratio of the converter is equal to that described in Eq. (7). Each average current in the switches and the blocking capacitors is approximately one-half of those in the converter in FIG. 8.

[0116] FIG. 27 and FIG. 28 show a 3-stage non-isolated six-switch, three-inductor, cascaded step-down power converter and a 4-stage non-isolated eight-switch, four-inductor, cascaded step-down power converter, respectively. FIG. 29 shows a n-stage non-isolated 2×n-switch, n-inductor, cascaded step-down power converter of this invention, where n can be any integer. The switching power converter stages are identified similarly to the description for FIG. 24.

[0117] FIG. 30 shows a 3-stage non-isolated nine-switch, three-level, three-inductor cascaded power converter. The voltage conversion ratio of the circuit is equal to that of the converter in FIG. 10. The voltage conversion ratio is D/3, as described in Eq. (8), as long as the switch activation control signals of the adjacent switches in the converter are not overlapped. If the switch activation control signals of the adjacent switches are overlapped when duty cycle D of the converter is large, the voltage conversion ratio monotonically increases from D/3 to 1 as duty cycle D increases.

[0118] FIG. 31 shows a n-stage non-isolated m×n-switch, m-level, n-inductor cascaded power converter of this invention, where m and n can be any integers.

[0119] Finally, the circuits of this invention can be implemented in a variety of ways. Specifically, multiple pairs of switches and energy storage devices can be connected in parallel to reduce the current stresses for each of the switching power converter stages in the power conversion system. Also, more than two power conversion systems can be connected in parallel for high current applications. FIG. 32 shows an example of two parallel-connected non-isolated power converters from circuits depicted in FIG. 8.

[0120] FIG. 32 depicts two parallel 2-stage non-isolated power conversion systems connected between a single input stage and a single output stage. Each of the two power conversion systems comprises a 2-switch, 2-inductor cascaded switching power converter. The input stage comprises a voltage source, and the output stage comprises an output capacitor. In the present embodiment, the first cascaded switching power converter comprises switches S₁, S₂, energy storage device (or blocking capacitor) C₁, inductors L₁, L₂, and synchronized rectifier switches S_R1, S_R2. The second cascaded switching power converter comprises switches S₃, S₄, energy storage device (or blocking capacitor) C₂, inductors L₃, L₄, and synchronized rectifier switches S_R3, S_R4. The first and second cascaded switching power converters are coupled between the input stage and the output stage. In the present embodiment, the input stage comprises a voltage source V_IN, and the output stage comprises an output capacitor C_F. Coupled to the output capacitor is load R. The input stage is coupled to switches S₁ and S₃ of the respective cascaded switching power converters, and the output stage is coupled inductors L₁, L₂, L₃, and L₄ of the respective cascaded switching power converters. Through using multiple cascaded switching power converters in parallel, current stresses are reduced in the circuit components of the power conversion system. While the present embodiment depicts two power conversion systems connected in parallel, the device could be arranged to have any number of n-switch, n-inductor cascaded switching power converters in parallel, where n is an integer. Additionally, step-down and step-up power conversion systems may be coupled together in a buck-boost arrangement, without departing from the spirit or scope of the invention.

[0121] While various embodiments of the present invention have been described above, it should be understood that they have been presented by way of example only, and not limitation. Thus, the breadth and scope of the present invention should not be limited by any of the above described embodiments, but should instead by defined only in accordance with the following claims and their equivalents.