Apparatus and method for constant delta current control in a capacitor charger

[0001] The present invention is related generally to a capacitor charger, and more particularly, to an apparatus and method for constant delta current control in a capacitor charger. As generally used in the art, the term “delta current” is referred hereinafter to the difference between the maximum and minimum in the desired range of a current.

[0002] Portable apparatus is more and more popular, and therefore capacitor charger it uses receives more attentions than ever. FIG. 1 shows a diagram of a conventional capacitor charger 100 implemented with ring choke converter (RCC) circuit, in which a transformer T1 connected with an input voltage V_in transforms a primary current I_P to a secondary current I_S that flows through a diode D1 to charge a capacitor C1 to generate an output voltage V_out at the output 112 of the charger 100 to supply for a load R_L. To control the current I_P or I_S, a power transistor Q1 has its collector 104 and emitter 108 connected to the primary winding N1 of the transformer T1 and ground, respectively, and resistors R1 and R2 are connected between the input 102 of the charger 100 and another primary winding N3 of the transformer T1 for the bias to the base of the transistor Q1. Although the capacitor charger 100 is cheaper, the transformer T1 has a more complicated structure to construct the circuit, and the power transistor Q1 causes higher switching loss. In addition, there are too many parameters effective to the current I_P, for example the input voltage V_in, transformer T1 and power transistor Q1, for the current I_P to be precisely controlled. Furthermore, the maximum of the current I_P has to be set much smaller than the base current I_b of the power transistor Q1, and much longer charging time is thus needed.

[0003] FIG. 2 shows a diagram of another conventional capacitor charger 200, in which a transformer T1 connected with a battery voltage V_bat transforms a primary current I_P to a secondary current I_S to charge a capacitor C4 for generating an output voltage V_out. To control the current I_P or I_S, a current sense resistor R_S1 is connected between a transistor Q1 and ground, another current sense resistor R_S2 is connected between the secondary winding N2 of the transformer T1 and ground, and operational amplifiers A1 and A2 are used to amplify the voltage drops across the resistors R1 and R2 to generate detector signals S1 and S2 for the inputs R and S of a flip-flop 206, respectively, so as to generate a switch signal S3 for switching the transistor Q1 through a driver 208. In the capacitor charger 200, the primary current I_P and secondary current I_S of the transformer T1 are directly detected by the current sense resistors R_S1 and R_S2 to determine the maximum and minimum of the primary current I_P. However, for not affecting the operations of the charger 200, only much small resistors can be used for the current sense resistors R_S1 and R_S2, and therefore the voltage drops across the resistors R_S1 and R_S2 are very small. As a result, the currents flowing through the resistors R_S1 and R_S2 cannot be precisely detected, and it is also easily to introduce spikes on the primary current I_P out of the maximum current that the battery could supply for the charger 200. Furthermore, the charging current I_P of the charger 200 is not easy to set.

[0004] FIG. 3 shows an application of yet another conventional capacitor charger 300 for a flash lamp module 306, in which a transformer 304 has a primary winding L1 connected between an input voltage V_bat and a transistor 308 to transform the primary current I_P to a secondary current I_S by switching the transistor 308, so as to charge a capacitor C1 for supplying to the flash lamp module 306. To control the charging current I_S, resistors R1, R2 and R3 are connected in series between the output 316 of the charger 300 and ground to detect the voltage on the capacitor C1 by voltage dividing to provide a feedback signal V_FB to a controller integrated circuit chip 302, so as to stop charging the capacitor C1 when the voltage on the capacitor C1 reaches its threshold. In the capacitor charger 300, it is the maximum duty of the transistor 308 to be used to determine the charging current I_P, and unfortunately, which is easy to have the transformer T1 to be magnetically saturated and inefficiently operated. Moreover, the input voltage V_bat is supplied by battery, and it will decade as time goes by, resulting in longer and longer charging time for the capacitor C1 to reach the same voltage.

[0005] One object of the present invention is to provide a control apparatus and method for a capacitor charger.

[0006] Another object of the present invention is to provide an apparatus and method for a capacitor charger without introducing power loss resulted from current detection.

[0007] Yet another object of the present invention is to provide an apparatus and method for a capacitor charger without having the transformer to be magnetically saturated.

[0008] In a capacitor charger using a transformer to transform a primary current to a secondary current to charge a capacitor and a switch connected in series with the primary winding of the transformer to switch the primary current, according to the present invention, a control apparatus and method are used to adjust the on-time period of the switch depending on the input voltage connected to the capacitor charger. Preferably, the control apparatus includes a voltage generator to provide a voltage varying with the input voltage for the control apparatus to determine the on-time period for the switch. Preferably, the voltage provided by the voltage generator is supplied to a current source to generate a reference current that is mirrored by a current mirror to a charge/discharge circuit to generate a charged voltage for a comparator to compare with a reference voltage to generate a comparison signal, and on-time control logics is preferably included in the control apparatus to determine the on-time period of the switch in response to the comparison signal.

[0009] From one aspect of the present invention, it is easy to determine and set the delta current flowing through the primary or secondary winding of the transformer, thereby achieving simple and precise control.

[0010] From another aspect of the present invention, the delta current of the primary winding of the transformer is independent of the input voltage connected to the primary winding, thereby preventing the transformer from magnetic saturation.

[0011] From still another aspect of the present invention, there are no current sense resistors connected in series to the current path of the primary or secondary winding of the transformer, thereby introducing no power loss resulted therefrom.

[0012] According to the present invention, the on-time period of the switch varies with the input voltage, and therefore the influence to the delta current of the primary current resulted from the variation of the input voltage is eliminated, resulting in the delta current of the charging current could be a constant value and easily to set. In addition, the secondary current could be precisely controlled due to the off-time control, and therefore, the maximum of the primary current and the minimum of the secondary current both could be controlled in a desired range, thereby no magnetic saturation occurred to the transformer. Moreover, no current sense to the primary winding of the transformer is used to determine the on-time period of the switch, and thus there will be no power loss resulted from current detection to be introduced.

[0019] FIG. 4 shows an application of one embodiment according to the present invention for a flash lamp module. In a capacitor charger 400, a transformer 416 has a primary winding L1 and a secondary winding L2 with a turns ratio N_P:N_S, a transistor M1 is connected between the primary winding L1 and ground GND, and the primary current I_P flowing through the primary winding L1 is controlled by the switching transistor M1. Briefly, the capacitor charger 400 has system control logics 414 to switch the transistor M1 through a driver 424 in response to an on-time decision circuit and an off-time decision circuit. The on-time decision circuit includes on-time control logics 412 to determine the on-time period T_on for the transistor M1 according to the output S_C generated by an on-time period generator that includes a voltage generator 402, a current source 404, a current mirror 406, a charge/discharge circuit 408, and a comparator 410. On the other hand, the off-time decision circuit includes off-time control logics 420 to determine the off-time period T_off for the transistor M1 according to the output S_S generated by an off-time period generator 418. Systematically, neither the charging time nor the charging current is dependent of the battery voltage V_bat.

[0020] In the on-time period generator, the voltage generator 402 is connected with the input voltage V_bat to generate a voltage V_d varying with the input voltage V_bat. In this embodiment, as shown in FIG. 4, the voltage generator 402 includes a voltage divider composed of resistors R1 and R2 connected in series, and therefore provides the voltage

[0021] $\begin{array}{cc}{V}_d=V_{\mathrm{bat}}\times \frac{R_2}{R_1+R_2}.& \left[\mathrm{EQ}\text{-}1\right]\end{array}$
The current source 404 determines a current I_A according to the voltage V_d. In this embodiment, the current source 404 includes a voltage-to-current converter that has a transconductive amplifier comprised of an operational amplifier 4042 and a transistor 4044 and a resistor R3 connected between the transistor 4044 and ground GND, in which the operational amplifier 4042 has a non-inverting input connected with the voltage V_d, an inverting input connected to the resistor R3, and an output connected to the gate of the transistor 4044. Due to the virtual ground between the inputs of the operational amplifier 4042, the voltage drop across the resistor R3 is also V_d, and thus determines the current

[0022] $\begin{array}{cc}{I}_A=\frac{V_d}{R_3}.& \left[\mathrm{EQ}\text{-}2\right]\end{array}$
From the equations EQ-1 and EQ-2,it is obtained the current

[0023] $\begin{array}{cc}{I}_A=V_{\mathrm{bat}}\times \frac{R_2}{R_1+R_2}\times \frac{1}{R_3}.& \left[\mathrm{EQ}\text{-}3\right]\end{array}$
The current I_A provided by the current source 404 is used as the reference current for the current mirror 406 to generate a mirror current I_B by mirroring the reference current I_A supplied to the charge/discharge circuit 408. Since the reference current I_A and the mirror current I_B have the same magnitude, they are

[0024] $\begin{array}{cc}{I}_B=I_A=V_{\mathrm{bat}}\times \frac{R_2}{R_1+R_2}\times \frac{1}{R_3}.& \left[\mathrm{EQ}\text{-}4\right]\end{array}$
The charge/discharge circuit 408 includes a capacitor C1 and a transistor M2 serving as a switch shunt to the capacitor C1. The current I_B charges the capacitor C1 to generate a charged voltage V_C that is compared with a reference voltage V_tf by the comparator 410 to generate the comparison signal S_C for the on-time control logics 412 to determine the on-time period

[0025] $\begin{array}{cc}{T}_{\mathrm{on}}=\frac{C_1\times V_{\mathrm{tf}}}{I_B}.& \left[\mathrm{EQ}\text{-}5\right]\end{array}$
From the equations EQ-4 and EQ-5, it is obtained the on-time period

[0026] $\begin{array}{cc}{T}_{\mathrm{on}}=\frac{C_1\times V_{\mathrm{tf}}\times R_3\times \left(R_1+R_2\right)}{V_{\mathrm{bat}}\times R_2}.& \left[\mathrm{EQ}\text{-}6\right]\end{array}$

[0027] In the off-time decision circuit, the off-time period generator 418 generates the signal S_S according to the secondary current I_S, and thus the off-time control logics 420 could determine the off-time period T_off supplied to the system control logics 414. Based on the on-time period T_on and the off-time period T_off, the system control logics 414 could switch the transistor M1 by the driver 424 to control the primary current I_P, and moreover, reset the capacitor C1 by turning on the transistor M2 shunt to the capacitor C1 in the charge/discharge circuit 408.

[0028] The transformer 416 has the primary current
I_P=V_bat×T_on=L₁×ΔI_P,  [EQ-7]
where ΔI_P is the delta current, i.e., the difference between the maximum and minimum, of the primary current I_P. It thus has the on-time period

[0029] $\begin{array}{cc}{T}_{\mathrm{on}}=\frac{L_1\times \Delta I_P}{V_{\mathrm{bat}}}.& \left[\mathrm{EQ}\text{-}8\right]\end{array}$
From the equations EQ-6 and EQ-8, it is obtained the on-time period

[0030] $\begin{array}{cc}{T}_{\mathrm{on}}=\frac{L_1\times \Delta I_P}{V_{\mathrm{bat}}}=\frac{C_1\times V_{\mathrm{tf}}\times R_3\times \left(R_1+R_2\right)}{V_{\mathrm{bat}}\times R_2},& \left[\mathrm{EQ}\text{-}9\right]\end{array}$
and thereby the delta current

[0031] $\begin{array}{cc}\Delta I_P=\frac{C_1\times V_{\mathrm{tf}}\times R_3\times \left(R_1+R_2\right)}{L_1\times R_2}.& \left[\mathrm{EQ}\text{-}10\right]\end{array}$
From the equation EQ-10, it is shown that the delta current ΔI_P of the primary current I_P in the transformer 416 is independent of the input voltage V_bat, and is determined by the device parameters R1, R2, R3, C1, V_tf, and L1.

[0032] Since the maximum I_Pmax and minimum I_Pmin of the primary current I_P follow the relation
I_Pmax=ΔI_P+I_Pmin,  [EQ-11]
and the transformer 416 has the transformation

[0033] $\begin{array}{cc}{I}_{P\min }=\frac{N_S}{N_P}\times I_{S\min },& \left[\mathrm{EQ}\text{-}12\right]\end{array}$
where I_Smin is the minimum of the secondary current I_S, it is obtained the maximum of the primary current

[0034] $\begin{array}{cc}{I}_{P\max }=\Delta I_P+\frac{N_S}{N_P}\times I_{S\min }.& \left[\mathrm{EQ}\text{-}13\right]\end{array}$
If the minimum I_Smin of the secondary current I_S is fixed, then it is only needed to set the delta current ΔI_P of the primary current I_P for setting the maximum I_Pmax of the primary current I_P to prevent the transformer 416 from magnetic saturation. It is shown by the equation EQ-10 that the delta current ΔI_P of the primary current I_P could be set by determining the device parameters of the resistors R1, R2 and R3, capacitor C1, inductor L1, and reference voltage V_tf, for example, by selecting the resistance of the resistor R3 to set the delta current ΔI_P, and from the equation EQ-10 the resistor R3 has the resistance

[0035] $\begin{array}{cc}{R}_3=\frac{\Delta I_P\times L_1\times R_2}{C_1\times V_{\mathrm{tf}}\times \left(R_1+R_2\right)},& \left[\mathrm{EQ}\text{-}14\right]\end{array}$
or by selecting the capacitance of the capacitor C1 to set the delta current ΔI_P of the primary current I_P, and from the equation EQ-10 the capacitor C1 has the capacitance

[0036] $\begin{array}{cc}{C}_1=\frac{\Delta I_P\times L_1\times R_2}{V_{\mathrm{tf}}\times R_3\times \left(R_1+R_2\right)}.& \left[\mathrm{EQ}\text{-}15\right]\end{array}$
Likewise, the secondary current I_S of the transformer 416 has the maximum

[0037] $\begin{array}{cc}{I}_{S\max }=\frac{N_P}{N_S}\times I_{P\max },& \left[\mathrm{EQ}\text{-}16\right]\end{array}$
and together with the equation EQ-13, it is obtained the maximum of the secondary current

[0038] $\begin{array}{cc}{I}_{S\max }=\frac{N_P}{N_S}\Delta I_P\times I_{S\min }.& \left[\mathrm{EQ}\text{-}17\right]\end{array}$
Therefore, if the minimum I_Smin of the secondary current I_S is fixed, it could set the maximum I_Smax of the secondary current I_S by setting the delta current ΔI_P of the primary current I_P.

[0039] It is shown by the embodiment 400 that the delta current ΔI_P of the primary current I_P of the transformer 416 is determined by the resistors R1, R2 and R3, capacitor C1, inductor L1 and reference voltage V_tf, and therefore, it is easy to set the delta current ΔI_P and the maximum I_Pmax of the primary current I_P by selecting these parameters for not having the transformer 416 magnetically saturated.

[0040] FIG. 5 shows an application of another embodiment according to the present invention for a flash lamp module. In a capacitor charger 500, the basic architecture and the on-time decision circuit of the capacitor charger 400 shown in FIG. 4 are employed likewise, and the same numerals for those corresponding elements are also designated hereto. While to generate the signal Ss, a voltage generator 502 composed of resistors R4 and R5 is connected between the secondary voltage V_L2 and ground GND to generate a voltage V_d2 varying with the secondary voltage V_L2 as

[0041] $\begin{array}{cc}{V}_{\mathrm{d2}}=V_{\mathrm{L2}}\times \frac{R_5}{R_4+R_5}.& \left[\mathrm{EQ}\text{-}18\right]\end{array}$
A current source 504 determines a current I_A2 according to the voltage V_d2. In the embodiment, the current source 504 includes a voltage-to-current converter that has a transconductive amplifier comprised of an operational amplifier 5042 and a transistor 5044 and a resistor R6 connected between the transistor 5044 and ground GND, in which the operational amplifier 5042 has a non-inverting input connected with the voltage V_d2, an inverting input connected to the resistor R6, and an output connected to the gate of the transistor 5044. Due to the virtual ground between the inputs of the operational amplifier 5042, the voltage drop across the resistor R6 is V_d2, and thus determines the current

[0042] $\begin{array}{cc}{I}_{\mathrm{A2}}=\frac{V_{\mathrm{d2}}}{R_6}.& \left[\mathrm{EQ}\text{-}19\right]\end{array}$
From the equations EQ-18 and EQ-19, it is obtained the current

[0043] $\begin{array}{cc}{I}_{\mathrm{A2}}=V_{\mathrm{L2}}\times \frac{R_5}{R_4+R_5}\times \frac{1}{R_6}.& \left[\mathrm{EQ}\text{-}20\right]\end{array}$
The current I_A2 provided by the current source 504 is used as the reference current for a current mirror circuit 506 to generate a mirror current I_B2 by mirroring the reference current I_A2 supplied to the charge/discharge circuit 508. Since the reference current I_A2 and the mirror current I_B2 have the same magnitude, they are

[0044] $\begin{array}{cc}{I}_{\mathrm{B2}}=I_{\mathrm{A2}}=V_{\mathrm{L2}}\times \frac{R_5}{R_4+R_5}\times \frac{1}{R_6}.& \left[\mathrm{EQ}\text{-}21\right]\end{array}$
The charge/discharge circuit 508 includes a capacitor C3 and a transistor M3 serving as a switch shunt to the capacitor C3, and the transistor M3 is controlled by the off-time control logics 420. The current I_B2 charges the capacitor C3 to generate a charged voltage V_C2 that is compared with a reference voltage V_tf2 by a comparator 510 to generate a comparison signal Ss for the off-time control logics 420 to determine the off-time period

[0045] $\begin{array}{cc}{T}_{\mathrm{off}}=\frac{C_3\times V_{\mathrm{tf2}}}{I_{\mathrm{B2}}}.& \left[\mathrm{EQ}\text{-}22\right]\end{array}$
From the equations EQ-21 and EQ-22, it is obtained the off-time period

[0046] $\begin{array}{cc}{T}_{\mathrm{off}}=\frac{C_3\times V_{\mathrm{tf2}}\times R_6\times \left(R_4+R_5\right)}{V_{\mathrm{L2}}\times R_5},& \left[\mathrm{EQ}\text{-}23\right]\end{array}$
and the off-time period is also

[0047] $\begin{array}{cc}{T}_{\mathrm{off}}=\frac{L_2\times \Delta I_S}{V_{\mathrm{L2}}}.& \left[\mathrm{EQ}\text{-}24\right]\end{array}$
By combining the equations EQ-23 and EQ-24, it is obtained the off-time period

[0048] $\begin{array}{cc}{T}_{\mathrm{off}}=\frac{L_{21}\times \Delta I_S}{V_{\mathrm{L2}}}=\frac{C_3\times V_{\mathrm{tf2}}\times R_6\times \left(R_4+R_5\right)}{V_{\mathrm{L2}}\times R_5},& \left[\mathrm{EQ}\text{-}25\right]\end{array}$
and thereby the resistance

[0049] $\begin{array}{cc}{R}_6=\frac{\Delta I_S\times L_2\times R_5}{C_3\times V_{\mathrm{tf2}}\times \left(R_4+R_5\right)}.& \left[\mathrm{EQ}\text{-}26\right]\end{array}$
From the equation EQ-26, it is shown that the delta current ΔI_S of the secondary current in the transformer 416 is proportional to the resistance of the resistor R6, and by which it could set the output current of the secondary winding L2 in the transformer 416 by setting the resistance of the resistor R6.

[0050] To control the capacitor chargers 400 and 500 or to switch the switch M1, there is no current sense resistor connected in series to the windings L1 or L2 of the transformer 416, and therefore no power loss is introduced by the current sense resistors. The delta current ΔI_P of the primary current I_P is easily to set and well controlled, and thus the transformer 416 could be prevented from magnetic saturation, thereby improving the efficiency thereof. The variation of the battery V_bat is also used to adjust the on-time period of the switch M1, thereby optimizing the operations of the capacitor chargers 400 and 500 adaptive to the varied input voltage V_bat, and the charging time and charging current are independent of the battery voltage V_bat.

[0051] While the present invention has been described in conjunction with preferred embodiments thereof, it is evident that many alternatives, modifications and variations will be apparent to those skilled in the art. Accordingly, it is intended to embrace all such alternatives, modifications and variations that fall within the spirit and scope thereof as set forth in the appended claims.