Control circuit for controlling output current at the primary side of a power converter

[0001] 1. Field of the Invention

[0002] The present invention relates to a power converter, and more particularly, to the control circuit of the power converter.

[0003] 2. Description of Related Art

[0004] Various power converters have been widely used to provide regulated voltage and current. For safety reasons, galvanic isolation is provided between a primary side and a secondary side of an off-line power converter. In case that the control circuit is configured at the primary side of the power converter, it would be difficult to control an output current of the power converter. Thus it is desirable to provide a control circuit for controlling the output current of the power converter at the primary side of the power converter.

[0005] The present invention provides a control circuit for controlling an output current at a primary side of a power converter. The control circuit comprises a switching control circuit for generating a switching signal for switching a transformer and regulating an output of the power converter. The switching control circuit includes an operational amplifier and a reference voltage developing an error amplifier for output current control, a comparator associated with a PWM circuit controlling the pulse width of the switching signal in response to the output of the error amplifier.

[0006] The control circuit further comprises an oscillator generates an oscillation signal for determining the switching frequency of the switching signal. A waveform detector generates a current-waveform signal by sampling a primary side switching current signal. A discharge-time detector coupled to the transformer detects a discharge-time of the secondary side switching current. An integrator produces an integrated signal by integrating an average current signal with the discharge-time. The integrated signal is coupled to the input of the error amplifier, therefore the output current of the power converter is regulated.

[0007] The average current signal is produced in response to the current-waveform signal, and the time constant of the integrator is correlated with a switching period of the switching signal. Therefore the integrated signal is proportional to the output current of the power converter.

[0017] Referring to FIG. 1, a power converter according to one embodiment of the present invention is illustrated. The power converter comprises a transformer 10 having an auxiliary winding N_A, a primary winding N_P, and a secondary winding N_S. In order to regulate the output voltage V_O and/or the output current I_O of the power converter, a control circuit 70 generates a switching signal V_PWM to switch the transformer 10 by switching a transistor 20. Referring to FIG. 2, it illustrates various signal waveforms of the power converter in FIG. 1 operating in discontinuous conduction mode. A primary side switching current I_P is generated as the switching signal V_PWM becomes high. A peak value I_PA of the primary side switching current I_P is given by, $\begin{array}{cc}{I}_{\mathrm{PA}}=\frac{V_{\mathrm{IN}}}{L_P}\times T_{\mathrm{ON}}& \left(1\right)\end{array}$
where V_IN is an input voltage applied to the transformer 10, L_P is the inductance of the primary winding N_P of the transformer 10, T_ON is an on-time of the switching signal V_PWM.

[0018] Once the switching signal V_PWM drops to low, the energy stored in the transformer 10 will be delivered to the secondary side of the transformer 10 and to an output of the power converter via a rectifier 40. A peak value I_SA of a secondary side switching current I_S can be expressed by, $\begin{array}{cc}{I}_{\mathrm{SA}}=\frac{\left(V_O+V_F\right)}{L_S}\times T_{\mathrm{DSD}}& \left(2\right)\end{array}$
where V_O is the output voltage of the power converter, V_F is the forward voltage dropped across the rectifier 40, L_S is the inductance of the secondary winding N_S of the transformer 10, and T_DSD is a discharging time of the secondary side switching current I_S in discontinuous conduction mode.

[0019] Meanwhile, a reflected voltage V_AUX is generated at the auxiliary winding N_A of the transformer 10. The reflected voltage V_AUX can be given by, $\begin{array}{cc}{V}_{\mathrm{AUX}}=\frac{T_{\mathrm{NA}}}{T_{\mathrm{NS}}}\times \left(V_O+V_F\right)& \left(3\right)\end{array}$
where T_NA and T_NS are respectively the winding turns of the auxiliary winding N_A and the secondary winding N_S of the transformer 10.

[0020] As the secondary side switching current I_S falls to zero, the reflected voltage V_AUX will start to decrease. This also indicates that the energy of the transformer 10 is fully released at this moment. Therefore, as shown in FIG. 2, the discharging time T_DSD in equation (2) can be measured from the falling edge of the switching signal V_PWM to the falling point of the reflected voltage V_AUX. The discontinuous conduction mode of the power converter features that the energy of the transformer 10 is fully released before the next switching cycle starts. The continuous conduction mode features that the energy of the transformer 10 is not fully released as the next switching cycle starts. FIG. 3 shows the waveforms of the power converter operating in continuous conduction mode. A peak value I_P(PEAK) of the primary side switching current I_P in continuous conduction mode is given by,

[0021]
I_P(PEAK)=I_PA+I_PB  (4) $\begin{array}{cc}{I}_{\mathrm{PA}}=\frac{V_{\mathrm{IN}}}{L_P}\times T_{\mathrm{ON}}& \left(5\right)\end{array}$

where I_PB represents the energy stored in the transformer 10.

[0022] Once the switching signal V_PWM drops to low, the energy stored in the transformer 10 will be delivered to the secondary side of the transformer 10. The secondary side switching current I_S is determined by the primary side switching current I_P and the winding turns of the transformer 10. The peak value I_S(PEAK) of the secondary side switching current I_S can be expressed as, $\begin{array}{cc}{I}_{S\left(\mathrm{PEAK}\right)}=\frac{T_{\mathrm{NP}}}{T_{\mathrm{NS}}}\times I_{P\left(\mathrm{PEAK}\right)}=\frac{T_{\mathrm{NP}}}{T_{\mathrm{NS}}}\times \left(I_{\mathrm{PA}}+I_{\mathrm{PB}}\right)& \left(6\right)\end{array}$
where T_NP is the winding turns of the primary winding N_P of the transformer 10.

[0023] Referring to FIG. 1, the control circuit 70 comprises a supply terminal VCC, a voltage-detection terminal VDET, a ground terminal GND, a current-sense terminal VS, and an output terminal VPWM. The output terminal VPWM outputs the switching signal V_PWM. The voltage-detection terminal VDET is coupled to the auxiliary winding N_A via a resistor 50 for detecting the reflected voltage V_AUX. The reflected voltage V_AUX further charges a capacitor 65 via a rectifier 60 for powering the control circuit 70. A current-sense resistor 30 is coupled from a source of the transistor 20 to a ground reference level for converting the primary side switching current I_P to a primary side switching current signal V_IP. The current-sense terminal VS is coupled to a current-sense device such as a current-sense resistor 30.

[0024] Referring to FIG. 4, it illustrates a schematic diagram of the control circuit 70 for output current control according to one embodiment of the present invention. A waveform detector 300 generates current-waveform signals V_A and V_B by sampling the primary side switching current signal V_IP. A discharge-time detector 100 detects the discharging time T_DSD/T_DSC of the secondary side switching current I_S via the auxiliary winding N_A of the transformer 10. An oscillator 200 generates an oscillation signal PLS to determine the switching frequency of the switching signal V_PWM.

[0025] An integrator 500 is used to generate an integrated signal V_X by integrating an average current signal I_AVG with the discharging time T_DSD/T_DSC. The average current signal I_AVG is produced in response to current-waveform signals V_A and V_B. Both discontinuous conduction mode and continuous conduction mode are considered. A time constant of the integrator 500 is correlated with a switching period T of the switching signal V_PWM. The integrated signal V_X is therefore proportional to the output current I_O of the power converter.

[0026] A switching control circuit includes an operational amplifier 71 and a reference voltage V_REF1 developing an error amplifier for output current control, a comparator 75 associated with a PWM circuit 400 controlling the pulse width of the switching signal V_PWM in response to an output of the error amplifier. The error amplifier amplifies the integrated signal V_X and provides a loop gain for output current control. A current control loop is formed from detecting the primary side switching current I_P to modulating the pulse width of the switching signal V_PWM. The current control loop controls the magnitude of the primary side switching current I_P in response to the reference voltage V_REF1. The secondary side switching current I_S is a ratio of the primary side switching current I_P as shown in equation (6). Referring to the waveform in FIG. 2 and/or FIG. 3, the output current I_O of the power converter is the average of the secondary side switching current I_S. The output current I_O of the power converter can be expressed by, $\begin{array}{cc}{I}_O=\left(I_{\mathrm{SB}}\times \frac{T_{\mathrm{DS}}}{T}\right)+\left(I_{\mathrm{SA}}\times \frac{T_{\mathrm{DS}}}{2T}\right)& \left(7\right)\end{array}$
where T_DS represents T_DSD in discontinuous conduction mode or T_DSC in continuous conduction mode. The output current I_O of the power converter is therefore regulated.

[0027] The primary side switching current I_P is converted to the primary side switching current signal V_IP by the current-sense resistor 30. The waveform detector 300 detects the primary side switching current signal V_IP and generates current-waveform signals V_A and V_B. The integrated signal V_X is designed by, $\begin{array}{cc}{V}_X=\left(V_B+\frac{V_A-V_B}{2}\right)\times \frac{T_{\mathrm{DS}}}{T_I}& \left(8\right)\end{array}$
where $\begin{array}{cc}{V}_A=\frac{T_{\mathrm{NS}}}{T_{\mathrm{NP}}}\times R_S\times \left(I_{\mathrm{SA}}+I_{\mathrm{SB}}\right)& \left(9\right)\end{array}$ $\begin{array}{cc}{V}_B=\frac{T_{\mathrm{NS}}}{T_{\mathrm{NP}}}\times R_S\times I_{\mathrm{SB}}& \left(10\right)\end{array}$
where the T_I is the time constant of the integrator 500.

[0028] Referring to equations (7)–(10), the integrated signal V_X can be rewritten as, $\begin{array}{cc}{V}_X=\frac{T}{T_I}\times \frac{T_{\mathrm{NS}}}{T_{\mathrm{NP}}}\times R_S\times I_O& \left(11\right)\end{array}$

[0029] It is noted that the integrated signal V_X is proportional to the output current I_O of the power converter. The integrated signal V_X increases whenever the output current I_O increases. However, the maximum value of the integrated signal V_X is limited to the value of the reference voltage V_REF1 through the regulation of the current control loop. Under feedback control of the current control loop, a maximum output current I_O(MAX) is given by, $\begin{array}{cc}{I}_{O\left(\mathrm{MAX}\right)}=\frac{T_{\mathrm{NP}}}{T_{\mathrm{NS}}}\times \frac{G_A\times G_{\mathrm{SW}}\times V_{\mathrm{R1}}}{1+\left(G_A\times G_{\mathrm{SW}}\times \frac{R_S}{K}\right)}& \left(12\right)\end{array}$
where K is a constant equal to T₁/T, V_R1 is the value of the reference voltage V_REF1, G_A is the gain of the error amplifier, and G_SW is the gain of the switching circuit.

[0030] If the loop gain of the current control loop is high (G_A×G_SW>>1), the maximum output current I_O(MAX) could be given by, $\begin{array}{cc}{I}_{O\left(\mathrm{MAX}\right)}=K\times \frac{T_{\mathrm{NP}}}{T_{\mathrm{NS}}}\times \frac{V_{\mathrm{R1}}}{R_S}& \left(13\right)\end{array}$

[0031] The maximum output current I_O(MAX) of the power converter is thus regulated as a constant current in accordance with the reference voltage V_REF1. FIG. 5 shows the output voltage V_O curve versus the variation of the output current I_O.

[0032] The PWM circuit 400 outputs the switching signal V_PWM for switching the power converter. The PWM circuit 400 according to one embodiment of the present invention comprises a D flip-flop 95, an inverter 93, an AND gate 91 and an AND gate 92. A D input of the D flip-flop 95 is supplied with a supply voltage V_CC. The oscillation signal PLS sets the D flip-flop 95 through the inverter 93. An output of the D flip-flop 95 is coupled to a first input of the AND gate 92. A second input of the AND gate 92 is coupled to an output of the inverter 93. An output of the AND gate 92 is also an output of the PWM circuit 400, which generates the switching signal V_PWM. The D flip-flop 95 is reset by an output of the AND gate 91. A first input of the AND gate 91 is supplied with a voltage-loop signal S_V. The voltage-loop signal S_V is generated by a voltage control loop, in which the voltage control loop is utilized to regulate the output voltage V_O of the power converter. A current-loop signal S_I generated from an output of the comparator 75 is supplied to a second input of the AND gate 91 for achieving output current control. A positive input of the comparator 75 is coupled to an output of the operational amplifier 71. A negative input of the comparator 75 is supplied with a ramp signal RMP that is provided by the oscillator 200. The voltage-loop signal S_V and the current-loop signal S_I can reset the D flip-flop 95 for shorten the pulse width of the switching signal V_PWM so as to regulate the output voltage V_O and the output current I_O.

[0033] Referring to FIG. 6, it illustrates a schematic circuit diagram of the detector 300 according to one embodiment of the present invention. A comparator 310 has a positive input coupled to the current-sense terminal VS. The value of the primary side switching current signal V_IP is proportional to the value of the primary side switching current I_P. A capacitor 321 is used to hold the peak value of the primary side switching current signal V_IP. A negative input of the comparator 310 is coupled to the capacitor 321. A constant current source 305 is applied to charge the capacitor 321. A switch 311 is coupled between the constant current source 305 and the capacitor 321. The switch 311 is turned on/off by an output of the comparator 310. A peak voltage signal V_SP is thus generated across the capacitor 321. The peak voltage signal V_SP is proportional to the sum current of I_PA and I_PB shown in FIG. 3. A transistor 308 is coupled in parallel with the capacitor 321 to discharge the capacitor 321. A switch 312 is used for periodically sampling the peak voltage signal V_SP from the capacitor 321 to a capacitor 322. Then the current-waveform signal V_A is obtained across the capacitor 322.

[0034] A switch 314 is coupled between the current-sense terminal VS and a capacitor 324. The capacitor 324 is used to hold the initial value of the primary side switching current signal V_IP. An initial voltage signal V_SI is therefore generated across the capacitor 324. The initial voltage signal V_SI is proportional to the value of the current I_PB shown in FIG. 3. A transistor 309 is coupled in parallel with the capacitor 324 to discharge the capacitor 324. A switch 315 serves to periodically sample the initial voltage signal V_SI from the capacitor 324 to a capacitor 325. Then the current-waveform signal V_B is generated across the capacitor 325.

[0035] An inverter 351, a current source 352, a transistor 353, a capacitor 354, and an AND gate 355 form a first time-delay circuit. An inverter 361, a current source 362, a transistor 363, a capacitor 364, an AND gate 365 and an inverter 366 form a first one-shot signal generator to output a store signal STR. The store signal STR is a one-shot signal. The switching signal V_PWM is supplied to an input of the first time-delay circuit. A current I₃₅₂ of the current source 352 and the capacitance of the capacitor 354 determine a delay time of the first time-delay circuit. An output of the first time-delay circuit is coupled to an input of the first one-shot signal generator. A current I₃₆₂ of the current source 362 and the capacitance of the capacitor 364 determine the pulse width of the store signal STR. The store signal STR controls the switch 314 to sample the initial value of the primary side switching current signal V_IP. Therefore the store signal STR is generated in response to the rising edge of a delayed switching signal. The delayed switching signal is generated in response to a rising edge of the switching signal V_PWM after the delay time. The delay time serves to avoid sampling from the interference of the switching spike.

[0036] Referring to FIG. 7, it illustrates a schematic circuit diagram of an integrator 500 according to one embodiment of the present invention. An operational amplifier 510, a resistor 511 and a transistor 512 construct a first V-to-I converter to generate a first programmable current I₅₁₂ in response to the voltage of the current-waveform signal V_B. Transistors 514, 515 and 519 form a first current mirror for producing a current I₅₁₅ and a current I₅₁₉ by mirroring the first programmable current I₅₁₂. Transistors 516 and 517 form a second current mirror for generating a current I₅₁₇ by mirroring the current I₅₁₅.

[0037] An operational amplifier 530, a resistor 531 and a transistor 532 form a second V-to-I converter for generating a second programmable current I₅₃₂ in response to the voltage of the current-waveform signal V_A. Transistors 534 and 535 form a third current mirror for producing a current I₅₃₅ by mirroring the second programmable current I₅₃₂. Transistors 536 and 537 develop a fourth current mirror for producing a current I₅₃₇ in response to the current I₅₃₅ and the current I₅₁₇. The current I₅₃₇ can be expressed by,
I₅₃₇=I₅₃₅−I₅₁₇

• • [0038] in which I₅₃₇=I₅₃₅−I₅₁₇. The geometric size of the transistor 536 is twice the size of the transistor 537. Therefore the current I₅₃₇ is the current I₅₃₆ divided by 2. Transistors 538 and 539 form a fifth current mirror for generating a current I₅₃₉ by mirroring the current I₅₃₇. The drains of the transistor 519 and the transistor 539 are coupled together for generating the average current signal I_AVG by summing the current I₁₅₉ and the current I₅₃₉. The average current signal I_AVG can be expressed by, $\begin{array}{cc}{I}_{\mathrm{AVG}}=\frac{V_B}{R_{511}}+\frac{\left(\frac{V_A}{R_{531}}-\frac{V_B}{R_{511}}\right)}{2}& \left(14\right)\end{array}$

[0039] The resistor 511, the resistor 531 and the capacitor 570 determine the time constant of the integrator 500, in which the resistor 531 is correlated to the resistor 511. As the resistance of the resistor 531 is set equal to the resistance of the resistor 511, the equation (14) can be thus rewritten as, $\begin{array}{cc}{I}_{\mathrm{AVG}}=\frac{1}{R_{511}}\times \left(V_B+\frac{V_A-V_B}{2}\right)& \left(15\right)\end{array}$

[0040] A switch 550 is coupled between the drain of the transistor 519 and a capacitor 570. The switch 550 is turned on only during the period of the discharge time T_DS of secondary side switching current I_S. A transistor 560 is coupled in parallel with the capacitor 570 to discharge the capacitor 570. A switch 551 serves to periodically sample the voltage across the capacitor 570 to an output capacitor 571. The integrated signal V_X is therefore generated across the output capacitor 571. $\begin{array}{cc}{V}_X=\frac{1}{R_{511}{C}_{570}}\times \left(V_B+\frac{V_A-V_B}{2}\right)\times T_{\mathrm{DS}}& \left(16\right)\end{array}$

[0041] Referring to FIG. 8, it illustrates a schematic diagram of the oscillator 200 according to one embodiment of the present invention. An operational amplifier 201, a resistor 210 and a transistor 250 form a third V-to-I converter. The third V-to-I converter generates a reference current I₂₅₀ in response to a reference voltage V_REF2. A plurality of transistors, such as 251, 252, 253, 254, 255 form current mirrors for generating an oscillator charge current I₂₅₃ and an oscillator discharge current I₂₅₅ in response to the reference current I₂₅₀. A drain of the transistor 253 generates the oscillator charge current I₂₅₃. A drain of the transistor 255 generates the oscillator discharge current I₂₅₅. A switch 230 is coupled between the drain of the transistor 253 and a capacitor 215. A switch 231 is coupled between the drain of the transistor 255 and the capacitor 215. The ramp signal RMP is obtained across the capacitor 215. A comparator 205 has a positive input connected to the capacitor 215. The comparator 205 outputs a oscillation signal PLS. The oscillation signal PLS determines the switching frequency and turns on/off the switches 312, 315 and 551. A first terminal of a switch 232 is supplied with a high-threshold voltage V_H. A first terminal of a switch 233 is supplied with a low-threshold voltage V_L. A second terminal of the switch 232 and a second terminal of the switch 233 are both coupled to a negative input of the comparator 205. An input of an inverter 260 is coupled to an output of the comparator 205 for generating an inverse oscillation signal /PLS. The oscillation signal PLS turns on/off the switch 231 and the switch 233. The inverse oscillation signal /PLS turns on/off the switch 230 and the switch 232. Inverters 261, 262, 263 and 264 are coupled in series. An input of the inverter 261 is supplied with the oscillation signal PLS. An AND gate 270 generates a clear signal CLR. A first input of the AND gate 270 is coupled to an output of the inverter 264. A second input of the AND gate 270 is coupled to an output of the inverter 261. The clear signal CLR turns on/off the transistors 308, 309 and 560. The resistance R₂₁₀ of the resistor 210 and the capacitance of the capacitor 215 determine the switching period T of the switching signal V_PWM. $\begin{array}{cc}T=\frac{C_{215}\times V_{\mathrm{OSC}}}{V_{\mathrm{REF2}}/R_{210}}=R_{210}\times C_{215}\times \frac{V_{\mathrm{OSC}}}{V_{\mathrm{REF2}}}& \left(17\right)\end{array}$
where V_OSC=V_H−V_L and C₂₁₅ is the capacitance of the capacitor 215.

[0042] Referring to FIG. 9, it illustrates a schematic diagram of the discharge-time detector 100 according to one embodiment of the present invention. An inverter 150, a transistor 122, a constant current source 120, a capacitor 121 and an AND gate 155 form a second time-delay circuit. An input of the second time-delay circuit is supplied with the switching signal V_PWM. The second time-delay circuit provides a propagation delay for the falling edge of the switching signal V_PWM. A current I₁₂₀ of the constant current source 120 and the capacitance of the capacitor 121 determine the timing of the propagation delay. An inverter 151, an inverter 152, a transistor 125, a constant current source 123, a capacitor 124 and an AND gate 156 form a second one-shot signal generator for generating a voltage-sample signal SMP. An input of the second one-shot signal generator is coupled to an output of the second time-delay circuit, which is also an output of the AND gate 155. A current I₁₂₃ of the constant current source 123 and the capacitance of the capacitor 124 determine the pulse width of the voltage-sample signal SMP.

[0043] An operational amplifier 101 performs as a buffer amplifier. A negative input and an output of the operational amplifier 101 are coupled together. A positive input of the operational amplifier 101, which is also an input of the buffer amplifier, is coupled to the voltage-detection terminal VDET. The voltage-detection terminal VDET is coupled to the auxiliary winding N_A of the transformer 10 via the resistor 50 for detecting the reflected voltage V_AUX. A switch 109 is coupled between an output of the buffer amplifier and a capacitor 112. The switch 109 is turned on/off by the voltage-sample signal SMP. Therefore, the reflected voltage V_AUX is sampled as a voltage V_DET. The voltage V_DET will be held across the capacitor 112. A comparator 105 is utilized to detect the decrease of the reflected voltage V_AUX. A positive input of the comparator 105 is coupled to the capacitor 112. A reference voltage 106 is coupled between a negative input of the comparator 105 and the output of the buffer amplifier to provide a threshold for detecting the decrease of the reflected voltage V_AUX. Therefore, the comparator 105 outputs a logic-high signal as the decrement of the reflected voltage V_AUX is beyond the voltage of the reference voltage 106. An input of an inverter 115 is supplied with the switching signal V_PWM. An input of an inverter 116 is supplied with the voltage-sample signal SMP. An AND gate 119 has a first input connected to an output of the comparator 105. A SR flip-flop 117 and a SR flip-flop 118 have a rising-edge triggered set-input and a high-level triggered reset-input respectively. The set-input of the SR flip-flop 118 is coupled to an output of the inverter 116. The reset-input of the SR flip-flop 118 is supplied with the switching signal V_PWM. An output of the SR flip-flop 118 is coupled to a second input of the AND gate 119. An output of the SR flip-flop 117 is coupled to a first input of an AND gate 114. A second input of the AND gate 114 is coupled to the output of the inverter 115. An output of the AND gate 114 serves to generate a discharge-time signal S_DS. The set-input of the SR flip-flop 117 is also coupled to the output of the inverter 115. The reset-input of the SR flip-flop 117 is coupled to an output of the AND gate 119. The discharge-time signal S_DS turns on/off the switch 550. The pulse width of the discharge-time signal S_DS is correlated to the discharge-time T_DS of the secondary side switching current I_S.

[0044] According to the preferred embodiment illustrated in FIGS. 4, and 6˜8, the integrated signal V_X is correlated to the secondary side switching current I_S and the output current I_O of the power converter. Thus, the equation (11) can be rewritten as, $\begin{array}{cc}{V}_X=m\times \frac{T_{\mathrm{NS}}}{T_{\mathrm{NP}}}\times R_S\times I_O& \left(18\right)\end{array}$
where m is a constant, which can be determined by, $\begin{array}{cc}m=\frac{R_{210}\times C_{215}}{R_{511}\times C_{570}}\times \frac{V_{\mathrm{OSC}}}{V_{\mathrm{REF2}}}& \left(19\right)\end{array}$

[0045] The resistance R₅₁₁ of the resistor 511 is correlated to the resistance R₂₁₀ of the resistor 210. The capacitance C₅₇₀ of the capacitor 570 is correlated to the capacitance C₂₁₅ of the capacitor 215. Therefore, the integrated signal V_X is proportional to the output current I_O of the power converter.

[0046] It will be apparent to those skilled in the art that various modifications and variations can be made to the structure of the present invention without departing from the scope or spirit of the invention. In view of the foregoing, it is intended that the present invention covers modifications and variations of this invention provided they fall within the scope of the following claims and their equivalents.