Dynamic current limit adjustments

[0001] This patent application is a continuation-in-part of U.S. patent application Ser. No. 10/703,960, which was filed Nov. 7, 2003, and claims the benefit under 35 U.S.C. 120 of the filing date.

[0002] The present invention relates to a system and method for adaptively adjusting the current limit in a switching regulator such as a buck regulator, a boost regulator, or a buck-boost regulator. A measurement system and method is employed to measure saturation points in an inductor of the regulator. The current limit can be adjusted such that saturation of the inductor is prevented.

[0003] Opposite polarity pulses can be applied to a transformer in a pulse-width-modulated (PWM) power converter. The secondary winding of the transformer is connected to additional circuitry such as a passive filter to generate an average output voltage. The average output voltage is related to the pulse-width by the volts-seconds rule and the characteristics of the transformer.

[0004] An example push-pull type PWM converter is illustrated in FIG. 1. As illustrated in FIG. 1, transformer T1 conducts current (IC1, IC2) from the BP power supply terminal through transistors Q1 and Q2. For example, the primary winding of transformer T1 conducts current IC1 when transistor Q1 is activated, while the primary winding of transformer T1 conducts current IC2 when transistor Q2 is activated. Diodes D1 and D2 operate on opposite polarity cycles to provide current (either ID1 or ID2) through inductor L to a load (not shown). Capacitor C0 is filter ripple in the output voltage (VO).

[0005] Ideally, transistors Q1 and Q2 are activated for equal amounts of time via drive signals DRV1 and DRV2 such that the volt-second integral of the pulses applied to the transformer corresponds to zero and the transformer core does not saturate. However, non-ideal switching times for transistors Q1 and Q2 may result in unequal cycle times such that the transformer core is driven into saturation. Core saturation in a PWM converter may result in secondary effects such as breakdown in the switching transistors (Q1, Q2), excessive voltage and current stress on the rectifier diodes (D1, D2), as well as EMI related problems.

[0016] Various embodiments of the present invention will be described in detail with reference to the drawings, where like reference numerals represent like parts and assemblies throughout the several views. Reference to various embodiments does not limit the scope of the invention, which is limited only by the scope of the claims attached hereto. Additionally, any examples set forth in this specification are not intended to be limiting and merely set forth some of the many possible embodiments for the claimed invention.

[0017] Throughout the specification and claims, the following terms take at least the meanings explicitly associated herein, unless the context clearly dictates otherwise. The meanings identified below are not intended to limit the terms, but merely provide illustrative examples for the terms. The meaning of “a,” “an,” and “the” includes plural reference, the meaning of “in” includes “in” and “on.” The term “connected” means a direct electrical connection between the items connected, without any intermediate devices. The term “coupled” means either a direct electrical connection between the items connected or an indirect connection through one or more passive or active intermediary devices. The term “circuit” means either a single component or a multiplicity of components, either active and/or passive, that are coupled together to provide a desired function. The term “signal” means at least one current, voltage, charge, temperature, data, or other signal.

[0018] Briefly stated, the invention is related to a system, method, and apparatus for providing current limit adjustment in a switching regulator that includes an inductor. A switched voltage divider circuit is selectively activated according to the actuation of various switching circuits in the regulator. The output of the switched voltage divider circuit is compared to a reference signal from a reference circuit to determine when a current limit is reached. At least one of the voltage divider ratio and the reference signal is adjustable in response to measurements related to the inductor saturation such that the current limit is changed to match inductor characteristics.

Measured Inductor Characteristics

[0019] Inductors in switching electronic circuits are typically made out of ferrites or some other core material that has a higher permeability than air. Unlike air core inductors, the amount of current flowing in these inductors is constrained by the quality and quantity of the core material. Once the inductor's operating current exceeds a saturation point, the inductor's effectiveness is reduced and the rate of increase in the current relative to the applied voltage rises measurably. The inductor voltage (V_L) and the inductor current (I_L) are related to one another by: ΔI_L/Δt=ΔV_L/L.

[0020] Magnetic load applications such as motor drivers and switched-mode power supplied (SMPS) incorporate current limiting circuits to protect the control circuits (e.g., drivers, inductive elements, etc.) from currents that are sufficient to cause saturation and induce destructive currents. For example, in a SMPS the saturation current should always be greater than the current limit in the controller.

[0021] The various measurements of voltages and/or currents can be used to calculate values of characteristics associated with the inductor. Example characteristics include: inductor value, series resistance, slew rate in non-saturation, slew rate in saturation, as well as others. Inductors with different core materials have different characteristic curves (e.g., the saturation and non-saturation slew rates change based on the type of core material) such that the type of core material and the value of the inductor can be determined (e.g., a lookup table that identifies the core material type based on characteristic measurements). The characteristic measurements can also be used in a control system such as in a switching-type converter (or regulator) to adaptively change control parameters such as: current limit, switching time, small signal compensation, and slope compensation.

[0022] Conventional circuits often include a current limit to prevent undesirable effects such as a short circuit, or protection from some other event that may damage various portions of the circuit. In the present invention, characteristics associated with the inductor are measured and various control parameters associated with current limit of the circuit are dynamically adjusted by the control circuit to accommodate a wide range of inductor values and saturation ratings. In one example, additional current sources are enabled/disabled to change a reference voltage associated with the current limit. In another example, voltage divider ratios are adjusted to change a feedback voltage associated with the current limit. In still another example, both reference voltages and feedback voltages may be adjusted to change the current limit in the circuit.

[0023] FIG. 2 is an illustration of an example embodiment of a switching converter (200) that is arranged according to an aspect of the present invention. Switching converter 200 includes an inductor (L), a switching transistor (T_SW), a diode (D_S), a capacitor (C_O), a load circuit (Z_L), and a measurement and system control (MSC) block.

[0024] Inductor L is charged when switching transistor T_SW is active. Inductor current (I_L) is delivered to capacitor C_O and load circuit Z_L via diode D_S when transistor T_SW is inactive. In a normal operating mode, the MSC block is arranged to actuate switching transistor T_SW such that inductor L is periodically charged. The MSC block is further arranged to monitor the voltages and currents associated with inductor L when the switching converter 200 is operated in a measurement mode, and provides a measurement of the inductance value based on those measurements.

[0025] The change in inductor voltage (ΔV_L) is determined by monitoring the input voltage (V_IN) and the switch voltage (V_SW). In one example embodiment, the change in inductor current (ΔI_L/Δt) is determined by monitoring the change in voltage across the switching transistor (ΔV_DS) and calculating: (ΔI_L/Δt)=(ΔV_DS/Δt)/rdson, where rdson is the on resistance of switching transistor T_SW. An optional resistor (R_S) may be placed in series with switching transistor T_SW to facilitate measurement of the inductor current.

[0026] FIG. 3 is an illustration of example waveforms for a switching converter such as that illustrated in FIG. 2. Ideally, the inductor does not reach a saturation point and the inductor current (I_L) and the voltage across transistor T_SW (V_DS) changes linearly in proportion to the value of the inductor. The non-ideal effects of inductor saturation are also illustrated as the inductor current (IL) and the voltage across transistor TSW (V_DS) experience a measurable change in slope when the saturation points (P₁ and P₂) are reached. The voltage across resistor R_S (when found in circuit 200) has the same shape as V_DS.

[0027] FIG. 4 is an illustration of a procedural flow for an example measurement system that is arranged according to an aspect of the present invention. After the measurement mode is activated by some triggering event, processing continues at block 410 where the inductor is operated for one or more cycles. At block 420, voltages (e.g., V_IN, V_SW, V_DS, V_RS, etc.) associated with the inductor are monitored (420) while the inductor is operated. At block 430, the monitored voltages are recorded and/or used to detect the saturation point (depending on the circuit implementation). From decision block 440, the process returns to block 420 when additional monitoring is necessary. Processing continues to block 450 when no additional voltages need be monitored. At block 450, values associated with the inductor (e.g., inductance value, slew rate, saturation point, etc.) are calculated. The calculated values can be used by other circuits or systems to adjust one or more parameters (e.g., small-signal compensation, current limit, slope compensation, etc).

[0028] In one example, voltages are monitored by an analog-type circuit that detects the slope associated with the current and/or voltage associated with the inductor. In another example, the inductor voltages and/or currents are monitored by an analog-to-digital converter. In this example, either the digital values can be recorded in a memory and later retrieved for calculation purposes, or the digital values can be evaluated in real time. In a further example, the digital values can be processed (e.g., averaged, throwing out extreme values, filtered, etc.) over many operating cycles of the inductor to minimize the effects of noise and switching transients on the measurement values. The processing of the digital values can be provided as a post-processing function or in real-time based on the processing power of the system implementation.

[0029] FIG. 5 illustrates another example embodiment of a switching converter (500) that is arranged in accordance with an aspect of the present invention. Switching converter 500 is buck-type converter that includes a p-type transistor (T_p), an n-type transistor (T_N), drivers (DRV_P, DRV_N), a controller (CNTL), a differentiator (DIFF), a comparator (CP), an SR-type latch (LATCH), a decrement circuit (DECR), a high-side turn-off circuit (TOHS), an inductor (L), a capacitor (C_O), and a load circuit (Z_L).

[0030] The controller (CNTL) is arranged to provide drive signals D_H and D_L to drivers DRV_P and DRV_N, respectively. Driver DRVP provides a gate control signal (G_P) to p-type transistor T_P, while driver DRVN provides another gate control signal (G_N) to n-type transistor T_N. The p-type transistor (T_p) is arranged to operate as a high side switch that couples the inductor (L) to the high power source (V_S). The n-type transistor (T_N) is arranged to operate as a low side switch that couples the inductor (L) to the low power source (GND). Capacitor C_O is coupled in parallel with load circuit Z_L, which provides a feedback signal (FB) from V_O to the controller (CNTL).

[0031] An example differentiator illustrated in FIG. 5 includes a capacitor (C), a resistor (R), and an amplifier (A_V). The differentiator is arranged to monitor the drain-source voltage (V_DS) of p-type transistor T_P and provide a signal (V_DIFF) that is proportional to ΔV_DS/Δt. The signal (V_DIFF) is used to adjust parameters within the circuit such as, for example, small-signal compensation, slope compensation, and current limit.

[0032] FIG. 6 illustrates an example of a digital processing circuit (600) that is arranged in accordance with aspects of the present invention. The digital processing circuit (600) includes an analog-to-digital converter (ADC) circuit, a memory circuit, and a digital processing logic (DPL) circuit. The ADC circuit is activated by a trigger mechanism (TRIGG) and converts a series of measurement signals (ISNS, VSNS, etc) to a data stream (raw data). The data stream is received by the memory circuit for storage. Data in the memory circuit can be retrieved by the DPL circuit for further processing.

[0033] In one example, the memory circuit stores sampled data points that are utilized to find the value of ΔV_DS/Δt, where the DPL circuit is arranged to calculate the value after all of the sampled points are stored. In another example, the memory circuit stores values that correspond to an average value, and the DPL circuit calculates a running average. The DPL circuit can further be arranged to: locate the saturation point of the inductor, calculate the slope of the inductor current, filter noise from the sampled data, discard anomalous data from the sampled data points, determine an appropriate current limit, determine a slope compensation parameter(s), determine a small-signal compensation parameter(s), as well as provide other calculations.

Dynamic Current Limit Adjustment Generally

[0034] FIG. 7 illustrates an example regulator system (700) that includes a dynamic current limit adjustment that is arranged according to at least one aspect of the present invention. System 700 includes an inductor (L), two switching transistors (T_SW, T_SW2), a diode (D_S), a reference circuit (REF), a comparator (COMP), an adjustable resistance circuit (R_LIM), an optional delay circuit (DLY), and a measurement and system control (MSC) block.

[0035] The system (700) illustrated in FIG. 7 is substantially similar in operation to that described with respect to FIG. 2, and like components are labeled identically. Delay circuit DLY is responsive to the control signal that is provided to switching transistor T_SW and provides another control signal to switching transistor T_SW2 after a short time delay. Switching transistor T_SW2 is arranged in cooperation with the adjustable resistance circuit (R_LIM) to operate as a switched voltage divider that provides a feedback voltage (V_FB). Resistance circuit R_LIM is optionally adjusted in response to a resistance control signal (S_LIM). Reference circuit REF is arranged to provide a voltage reference (V_REF), which is optionally adjusted in response to a reference control signal (S_REF). Comparator COMP is arranged to assert a current limit signal (LIMIT) in response to the comparison between voltage reference V_REF and feedback voltage V_FB.

[0036] The measurement system and control (MSC) block is arranged to control the switching of transistor T_SW as previously described with respect to FIG. 2. The MSC block is also arranged to adjust the current limit by adjusting at least one of the reference voltage (V_REF) and the level associated with the feedback voltage (V_FB). The amount of the feedback voltage is related to the voltage at the switch terminal (SW), the drain-to-source voltage of switching transistor T_SW2, the value associated with adjustable resistance circuit R_LIM, and the inductor current (I_L). Control signal SLIM is provided by the MSC block such that the resistance level associated with resistance circuit RLIM can be changed. Control signal SREF is also provided by the MSC block such that the voltage associated with the reference circuit can be changed.

[0037] The current limit may be adjusted to account for certain non-ideal effects in the overall system, including but not limited to, avoiding saturation of the core material in the inductor. For example, the MSC block may be configured to measure the current level associated with the inductor for which the core is detected as saturating (see previous discussions). After the current level is measured, the MSC block may then adjust at least one of the reference voltage (V_REF) and the feedback voltage (V_FB) so that the current limit level is reduced and core saturation is avoided during operation of the system. The process of adjusting the current limit in the system (700) can be performed a single time, periodically, after the occurrence of a detected event, or continually.

Dynamic Current Limit with Adjustable Feedback Voltage Level

[0038] FIG. 8 illustrates an example schematic diagram for a system (800) that employs a first current limit adjustment means that is arranged according to at least one aspect of the present invention. System 800 includes a switching transistors (T_SW2), an adjustable resistance circuit (R_LIM), and a reference circuit (REF) that is illustrated by a two transistors (M_REF1, M_REF2) and a current source (I_S).

[0039] The system (800) illustrated in FIG. 8 is substantially similar in operation to that is described with respect to FIG. 7, and like components are labeled identically. Adjustable resistance circuit R_LIM includes an array of N selectable transistors (M_R1−M_RN), and a transistor M_R0 that is always active. The N selectable transistors are configured to operate as switched resistances that are selectively coupled together in parallel with transistor M_R0. The parallel circuit is coupled to the switching transistor (T_SW) to operate as a switched voltage divider that can be adjusted in response to a control signal S_LIM (e.g., an N-bit signal bus with bits B₁−B_N).

[0040] Each of transistors (M_R0−M_RN) has a corresponding drain-source impedance when activated. The parallel combination of all of the active transistors yields an effective resistance that functions as part of the switched voltage divider that was previously described. The effective drain-source impedance of each transistor may be the same as one another, different from one another, ratioed to one-another according to binary values, or arranged to provide a specific set of effective impedances as may be desired. The number of adjustable resistance elements (the transistors) can be changed from 1 to N based on a desired granularity for adjusting the current limit.

[0041] In one example, the adjustable resistance elements (M_R1−M_RN) are scaled with respect to one another according to binary weighting such that the current limit has an adjustment range of approximately 20%. For example, all of the resistance elements can be initially activated such that the effective impedance is very low and the feedback voltage (V_FB) is reduced. By selectively deactivating various resistive elements from the parallel circuit, the overall impedance increases and the feedback voltage (V_FB) will also increase.

[0042] Transistors M_REF1 and M_REF2 are arranged as stacked transistors that have similar operating characteristics to switching transistor M_TSW (see FIG. 7). Current I_REF is provided to the stacked transistors (M_REF1 and M_REF2) by current source I_S, whereby a reference voltage (V_REF) is generated. The reference circuit is arranged such that changes in processing, operating temperature and other environmental influences are accounted for in the reference voltage.

Dynamic Current Limit with Adjustable Reference Voltage Level

[0043] FIG. 9 illustrates an example schematic diagram (900) for a system that employs a second current limit adjustment means, arranged in accordance with at least one aspect of the present invention. System 900 includes a switching transistors (T_SW2), a resistance circuit (M_NFB), and a reference circuit (REF) that is illustrated by a nine transistors (M_NR0−M_NRM, M_D, P_D, P_S1−P_SM) and a current source (Is).

[0044] The system (900) illustrated in FIG. 9 is substantially similar in operation to that is described with respect to FIG. 7, and like components are labeled identically. However, the adjustable resistance circuit RLIM from FIG. 7 is replaced with a fixed resistance value that is illustrated by transistor M_NFB.

[0045] Current source I_S is arranged to couple current I_REF to transistor MD, which is configured to operate as a diode. Transistors M_NR0−M_NRM are arranged in a current mirror configuration with transistor MD. Transistor M_NR0 is coupled to transistor PD, which is configured to operate as a diode. Transistors M_NR1−M_NRM are selectively coupled to transistor PD such that the total operating current for transistor PD can be adjusted in response to control signal S_REF. For the example illustrated in FIG. 9, control signal S_REF is an M-bit signal with bits B₁−B_M. Since the operating current of transistor P_D is selectable, the overall reference voltage provided as V_REF is adjustable.

[0046] In an alternative embodiment, the current mirror can provide a fixed current and a number of selectable devices can be placed in parallel with transistor PD to adjust the reference voltage. In yet another alternative embodiment, the value associated with current I_REF can be selectable. In still another embodiment, the circuit of FIG. 8 can be combined with the circuit of FIG. 9 to provide adjustments to both the feedback voltage and the reference voltage.

[0047] The above specification, examples and data provide a complete description of the manufacture and use of the composition of the invention. Since many embodiments of the invention can be made without departing from the spirit and scope of the invention, the invention resides in the claims hereinafter appended.