Apparatus and method for controlling voltage regulator and power supply apparatus

[00002] 1. Field of the Invention

[00003] The present invention relates to an apparatus and method for controlling a linear voltage regulator.

[00004] 2. Description of the Related Art

[00005] Linear voltage regulators are well known electronic devices. They are used to produce a steady output voltage at a predetermined level from an input voltage which may vary. The input voltage is higher than the output voltage and heat is dissipated in the regulator. The power dissipated given a constant load current is proportional to the voltage drop between the input and the output. For example, in a battery operated system where the operating voltage of the circuitry is significantly lower than the battery voltage, reducing the power dissipation in the regulator will lead to improved battery life. It will also reduce the thermal efficiency requirements of the regulator thereby allowing a small and cheaper package and pass transistor to be used. Conventionally, power supply efficiency has been improved by replacing the pass transistor with a switched inductor. A buck converter uses such an arrangement. However, the inductor tends to be a large and expensive component and is generally not suitable for miniaturization.

[00006] Therefore, an object of the present invention is to provide a method and circuitry for improving the efficiency of a linear regulator without using ferroelectric components.

[00007] Another object of the present invention is to reduce the input voltage to a linear regulator be switching small amounts of change between capacitors.

[00008] In an aspect of the present invention, a method for controlling a voltage regulator is achieved by a) providing first and second charge storage devices switchably connected between a voltage source and the voltage regulator; by b) switching the first storage device into connection with the voltage source until the voltage on it reaches a predetermined level; by c) disconnecting the first storage device from the voltage source and switching it into connection with the second storage device and the voltage regulator until the voltage input to the voltage regulator falls below a predetermined level; and by d) repeating steps b) and c).

[00009] Here, the storage devices may be capacitors connected in parallel with the voltage regulator, across the voltage source.

[00010] Also, the switching may be performed by two switches connected in series, one between the voltage source and the first storage device and the other between the first and second storage devices.

[00011] Also, the first storage device may be significantly larger than the second storage device.

[00012] In another aspect of the present invention, an apparatus for controlling a voltage regulator includes a voltage source, and a first and second charge storage devices connected between the voltage source and the voltage regulator. A section connects the first storage device to the voltage source and disconnects it from the second storage device and the voltage regulator until the voltage on the first storage device reaches a predetermined level. Another section disconnects the first storage device from the voltage source and connects it to the second storage device and the voltage regulator until the input voltage to the voltage regulator falls below a predetermined level. Still another section switches the storage devices between the 2 modes of operation.

[00013] Here, the storage devices may be capacitors, and the connecting section may include two switches are connected in series between the voltage source and the first storage device, the other between the two storage devices.

[00014] Also, it is preferable that the first storage device is substantially larger than the second storage device.

[00015] In still another aspect of the present invention, a power supply apparatus includes a power supply, a voltage regulator, and first and second capacitors provided between the power supply and the voltage regulator in parallel to the power supply. The apparatus further includes a first switch provided between the power supply and the first capacitor to open or close in response to a first control signal, and a second switch provided between the power supply and the second capacitor to open or close in response to a second control signal. A control circuit generates the first and second control signals to the first and second switches such that the second switch opens and then the first switch closes when a voltage of the second capacitor decreases to a first predetermined level, and such that the first switch opens and the second switch closes after a first predetermined time period from the closing the first switch.

[00016] Here, the first predetermined time may be a time period until a voltage of the first capacitor reaches a second predetermined level after the first switch is closed.

[00017] Also, the control circuit may generate the first and second control signals to repeat a switching operation in which the second switch opens and then the first switch closes when the voltage of the second capacitor decreases to the first predetermined level, and the first switch opens and the second switch closes after the first predetermined time period from the closing the first switch.

[00018] Also, the control circuit may monitor the voltage of the second capacitor and generates the first and second control signals based on the monitoring result.

[00019] Also, it is preferable that the second capacitor is larger in capacitance than the first capacitor.

[00020] In yet still another aspect of the present invention, a power supply apparatus includes a power supply, a voltage regulator, and first and second capacitors provided between the power supply and the voltage regulator in parallel to the power supply. The apparatus further includes a first switch provided between the power supply and the first capacitor to open or close in response to a first control signal, and a second switch provided between the power supply and the second capacitor to open or close in response to a second control signal. A control circuit generates the first and second control signals to the first and second switches such that the second switch opens and then the first switch closes when a voltage of the first capacitor decreases to a first predetermined level, and such that the first switch opens and the second switch closes after a first predetermined time period from the closing the first switch.

[00021] Here, the first predetermined time may be a time period until a voltage of the first capacitor reaches a second predetermined level after the first switch is closed.

[00022] Also, the control circuit may generate the first and second control signals to repeat a switching operation in which the second switch opens and then the first switch closes when the voltage of the first capacitor decreases to the first predetermined level, and the first switch opens and the second switch closes after the first predetermined time period from the closing the first switch.

[00023] Also, the control circuit may monitor the voltage of the first capacitor and generate the first and second control signals when the voltage of the first capacitor decreases to the first predetermined level.

[00024] Also, it is preferable that the second capacitor is larger in capacitance than the first capacitor.

[00025] Also, the control circuit may monitor the voltage of the first capacitor and an output voltage of the voltage regulator and generate the first and second control signals based on the voltage of the first capacitor to the output voltage of the voltage regulator.

[00031] A control apparatus of a linear regulator according to a first embodiment of the present invention is shown in FIG. 1. The control apparatus is comprised of a DC power supply or battery 2 which supplies an input voltage V_in. The power supply is connected in parallel with two capacitors C₁ and C₂. The capacitor C₁ is separated from the voltage V_in by a switch S₁. A further switch S₂ separates the capacitors C₁ and C₂. A linear regulator 4 is connected across the circuit downstream of the capacitor C₂ and has an output which produces a voltage V_out and a current I_load.

[00032] A control circuit 6 monitors the input voltage V_in to the linear regulator 4 and in response to the monitoring result, supplies control signals to the switches S₁ and S₂ which can be closed. The switch S₁ when closed will enable the capacitor C₁ to charge. The switch S₂ when closed will allow the capacitor C₂ to charge from the capacitor C₁ at the same time as providing input charge to the linear regulator 4.

[00033] The control circuit 6 is responsive to the voltage of the input to the linear regulator 4. When this voltage falls below a predetermined level, corresponding to the minimum required to maintain the output voltage V_out, the control circuit 6 opens the switch S₂ and closes the switch S₁, in that order. This causes the capacitor C₁ to be charged up to the battery voltage, whereupon the switch S₁ is reopened and the switch S₂ is closed (again in that order). A charge is transferred from the battery to the capacitor C₁ in the first stage where the switch S₁ is closed and in the second stage when the switch S₁ is opened and the switch S₂ is closed, the charge is transferred from the capacitor C₁ to the capacitor C₂ until the voltages across the capacitors are equalized. Subsequently, if a constant load current I^load is drawn from the regulator 4, the voltage on the capacitors C₁ and C₂ will decrease linearly until the switching threshold is reached again. Typically, the switching threshold will be set to such a level that recharging the capacitor C₁ by closing the switch S₁ and opening the switch S₂, and switching back to discharge of the capacitor C₁ by opening the switch S₁ and closing the switch S₂ can happen before the input voltage to the linear regulator 4 falls beneath the minimum required to maintain voltage V_out.

[00034] The traces in FIG. 2 show the voltage across the capacitors linked to the switching cycle, assuming that there are no resistive losses in the circuit. In practice, there will of course be resistive losses and the traces will be modified accordingly.

[00035] The average voltage at the input to the regulator 4 is the average of V_C2, and is given by: $\begin{array}{cc}{V}_{\mathrm{ave}}=\frac{V_{\mathrm{set}}+V_{\mathrm{out}}+V_{\mathrm{do}}}{2}& \left(1\right)\end{array}$

[00036] A voltage V_set is determined by considering the energy transferred between the capacitors. The energy stored in the capacitor C₁ while the switch S₁ is closed is given by: $\begin{array}{cc}{E}_1=\frac{C_1·V_{\mathrm{in}}^2}{2}& \left(2\right)\end{array}$

[00037] The energy remaining in the capacitor C₂ at the moment the switch S₂ closes is given by: $\begin{array}{cc}{E}_2=\frac{C_1·{\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)}^2}{2}& \left(3\right)\end{array}$

[00038] By conservation of energy, the combined energy of the capacitors C₁ and C₂ in parallel is given by: $\begin{array}{cc}{E}_c=E_1+E_2=\frac{C_1·V_{\mathrm{in}}^2}{2}+\frac{C_2·{\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)}^2}{2}& \left(4\right)\end{array}$
Also: $\begin{array}{cc}{E}_c=\frac{\left(C_1+C_2\right)·V_{\mathrm{set}}^2}{2}& \left(5\right)\end{array}$
Therefore: $\begin{array}{cc}{V}_{\mathrm{set}}=\sqrt{\frac{2·E_c}{C_1+C_2}}=\sqrt{\frac{C_1·V_{\mathrm{in}}^2+{C_2\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)}^2}{C_1+C_2}}& \left(6\right)\end{array}$

[00041] From the above equations, it can be deducted that the power drawn from the battery 2 is given by: $\begin{array}{cc}P=I_{\mathrm{load}}·V_{\mathrm{set}}=\frac{I_{\mathrm{load}}}{2}.\left[\sqrt{\frac{C_1·V_{\mathrm{in}}^2+{C_2\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)}^2}{C_1+C_2}}+V_{\mathrm{out}}+V_{\mathrm{do}}\right]& \left(7\right)\end{array}$

[00042] The power drawn from the battery 2 without the switch/capacitor circuit is given by:
P_old=I_load·V_in  (8)

[00044] Therefore, the improvement in power efficiency given by the circuit (ignoring power lost during the switching due to gate capacitance and switch/capacitor series resistance) is: $\begin{array}{cc}\frac{P}{P_{\mathrm{old}}}=\frac{1}{2·V_{\mathrm{in}}}·\left[\sqrt{\frac{C_1·V_{\mathrm{in}}^2+{C_2\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)}^2}{C_1+C_2}}+V_{\mathrm{out}}+V_{\mathrm{do}}\right]& \left(9\right)\end{array}$

[00045] It can be seen by examination of equation (1) that the best efficiency is obtained when C₂>>C₁ such that V_set=>(Vout+V_do). Then the improvement in efficiency approaches the ratio: $\begin{array}{cc}\frac{P}{P_{\mathrm{old}}}=\frac{V_{\mathrm{out}}+V_{\mathrm{do}}}{V_{\mathrm{in}}}& \left(10\right)\end{array}$

[00046] The period T between successive activations of the switches is dependent on the load current: $\begin{array}{cc}T=\frac{\left[V_{\mathrm{set}}-\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)\right]·\left(C_1+C_2\right)}{I_{\mathrm{load}}}& \left(11\right)\end{array}$
or substituting for V_set: $\begin{array}{cc}T=\frac{\left(C_1+C_2\right)}{I_{\mathrm{load}}}\left[\sqrt{\frac{C_1·V_{\mathrm{in}}^2+{C_2\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)}^2}{C_1+C_2}}-\left(V_{\mathrm{out}}+V_{\mathrm{do}}\right)\right]& \left(12\right)\end{array}$

[00048] So the switching period is inversely proportional to the load current, as would be expected. The period can be increased (to save power lost in switching) by making the capacitor C₁ as large as possible.

[00049] FIG. 3 shows the control apparatus of the regulator according to the second embodiment of the present invention. The portion of the circuit corresponding to the control circuit 6 of FIG. 1 is shown in dotted outline. The switches S₁ and S₂ are P-channel FET devices. The feedback arrangement of the voltage which in FIG. 1 is from the input voltage to the linear regulator (given by the voltage on the capacitor C₂) is in the embodiment replaced by feedback from the voltage on the capacitor C₁ and from the voltage output of the regulator 4 for V_out. The output voltage V_out is the voltage input to a voltage divider R₁ and R₂ and the output of this divider is provided to one input of a comparator 8. The voltage on the capacitor C₁ is fed by a further voltage divider R₃ and R₄ to the other inverted input of a comparator 8. The comparator has a hysteresis characteristic and outputs a pulse with a predetermined duration time when the output of the voltage divider R₃ and R₄ is lower than that of the voltage divider R₁ and R₂. The pulse duration time is sufficient to charge the capacitor C₁ to the power supply voltage.

[00050] The output of the comparator 8 is provided to the two inputs of a flip-flop comprising a pair of NAND gates and a pair of AND gates. The output of the comparator 8 goes directly to the input of a first NAND gate 10 and by the inverter 12 to the second NAND gate 14. The output of the second NAND gate 14 is connected to the other input of the first NAND gate 10, and the output of the first NAND gate 10 is connected to the second input of the second NAND gate 14. The output of the first NAND gate 10 is also connected to an input of an AND gate 16 whilst the output of the second NAND gate 14 is connected to an input of an AND gate 18. The other input of each of these AND gates 16 and 18 is connected to a start/enable line. The purpose of the AND gates is to enable the regulator 4 to start up.

[00051] FIG. 4 shows the voltage at various points in the circuit during start up and subsequent operation following the application of a signal to the start/enable line.

[00052] When the start/enable signal is low (probably by default when the battery 2 power V_bat is applied) as shown in FIG. 4A, both FET switches S₁ and S₂ are forced on, and the battery 2 voltage is applied directly to the regulator 4 input. This enables the regulator 4 to start up as normal in a low efficiency mode. This mode may also be used if the battery voltage falls to the point where the control apparatus ceases to provide any efficiency improvement or, alternatively, may be used in situations where harmonic interference caused by switching is undesirable (e.g., in a radio subsystem).

[00053] When the start/enable signal is set high as shown in FIG. 4A, e.g., by a micro-controller I/O port, the voltage on the capacitor C₁ will be higher than output of the regulator 4 and the output of the comparator 8 will be low as shown in FIG. 4B. Therefore, the voltage V_gs1 (the S₁ enabling voltage) will be high as shown in FIG. 4D, and so the switch S₁ will be open, and the voltage V_gs2 (S₂ enabling voltage) will be low as shown in FIG. 4E, which means the switch S₂ will be closed. If a load current is drawn from the regulator 4, the voltage on the capacitors C₁ and C₂ will fall as shown in FIGS. 4F and 4G. When the voltage V_c1 reaches a predetermined switching point, V_threshold (V_thr in FIG. 4), the comparator 8 output will go high as shown in FIG. 4B. The switching point is set relative to the output voltage V_out of the regulator 4, and can be adjusted by varying the ratio of R₃ to R₄. It should be chosen such that the voltage across the regulator 4 remains larger than the maximum dropout voltage V_do (the voltage drop across the regulator 4) at all times and under all load current conditions. Clearly, allowance needs to be made for the time taken to recharge the capacitor C₁, considering that in a practical implementation there will be a finite switching time for the FET's, and series resistance means that the capacitors do not charge instantaneously.

[00054] When the comparator 8 output switches to high in response to a reduction in the capacitor voltage V_c1 or the output voltage V_out, the flip-flop will cause the voltage V_gs2 to go high as shown in FIG. 4E. Thereby, the switch S₂ is opened. Also, the flip-flop will cause the voltage V_gs1 to go low as shown in FIG. 4D, thereby closing the switch S₁. The capacitor C₁ will then be charged and the voltage V_c1 will increase as shown in FIG. 4F. This will cause the comparator 8 output to go low as shown in FIG. 4B when the voltage V_c1 reaches a predetermined level. At this time, the switch S₁ is opened again as shown in FIG. 4D and the switch S₂ is closed as shown in FIG. 4E. Then, the cycle begins again.

[00055] The small amount of resistance in the switching circuit which was mentioned above consists of the series resistance of the battery, or voltage source, the series resistance of the switches and interconnections, and the series resistance of the capacitors. The effect of this is twofold. Firstly, it will reduce the efficiency of the circuit due to energy dissipation. Secondly, it will introduce a delay of the transfer of charge between the battery 2 and the capacitors C₁ and C₂. Any inductance in the circuit will also add to this delay. This means that the “on” time of the switch S₁ must be increased to allow the capacitor C₁ to be fully charged.

[00056] The circuit shown in FIG. 3 relies on propagation delays through the feedback circuitry to provide this delay. A more deterministic method might include a hysteresis component in the comparator, such that the voltage on C₁ has to approach the battery voltage before the comparator switches back.

[00057] A second consideration is the time required to turn the transistor on and off. This is determined by the size of the transistor, the gate capacitance, and the drive capability of the AND gates. Clearly, if there is a period during which both transistors are switched on, the capacitor C₂ will be directly charged from the battery 2, and the voltage of the regulator 4 input will be consequently higher. This leads to a reduction in efficiency of the circuit which can be dramatic. The NAND/inverter circuit is used to prevent any overlap between switching off one transistor and switching on the other. Nevertheless, this relies on the propagation delay through the NAND gates being greater than the switching time of the transistors FET1 and FET2. The delay can be increased by inserting extra delay buffers in the feedback path between the output of one NAND gate and the input of the other.

[00058] Another effect of the transistor switching time increases loss whilst the transistors are partially on and, hence resistive. Ideally, vary fast transistors should be used. However, this can usually only be achieved at the expense of series resistance and/or maximum current capability. Therefore, a compromise must be made based on the load requirements. Furthermore, it should be noted that the peak current flow from the battery I_peak can be high if the switch S₁ switches very quickly. A slower turn on time may be desirable to limit the transient current and possible associated noise problems.

[00059] The most obvious practical consideration is in the selection of the capacitors. Clearly, low ESR dielectrics such as ceramics will contribute less to the overall loss in the switching system. However, the larger the value of the capacitor C₁ becomes, the lower the switching frequency becomes, (furthermore, the capacitor C₂ should be significantly larger than the capacitor C₁) and this will increase efficiency. This is because a significant amount of power is lost in the switching of the gates and the transistors and therefore a low switching frequency is desirable. Preferably, therefore, the capacitor C₂ should be chosen to be as large as possible within the space and cost limitations of the system.

[00060] The switching transistors and feedback circuit could be integrated into a BiCMOS process with the linear regulator Bipolar/CMOS (CMOS: Complementary Metal Oxide Semiconductor). This would mean that the only external components required would be the capacitors. All linear regulators do in fact require an input and an output capacitor for stability and smoothing and, therefore, in fact only one extra capacitor C₁ would be required. The system therefore offers considerable efficiency gains over a standard linear regulator through the addition of one extra capacitor. The system also offers advantages over ferro-electric switch mode converters such as buck regulators. Capacitors are generally cheaper, smaller, have lower series resistance and radiate less than inductors and transformers. All of these are qualities which are of particular importance for portable telecommunications systems.

[00061] Where higher current applications are required, it is necessary to use tantalum or electrolytic capacitors. Large transistors are also required for low resistance in high current applications.

[00062] Interference due to high peak charge currents may occur, and the switching frequency is not predictable, this being dependent on the load current. This can easily be overcome by using a fixed frequency clock, rather than a comparator to drive the switches. This arrangement, however, would be less efficient at low loads.