Analog-to-digital converter bank based ultra wideband communications

[0002] This invention relates generally to ultra wideband communications.

[0003] Ultra wideband communications (UWB) is true digital radio communication; completely unlike the radios we listen to and communicate every day. UWB is a wireless broadband communications technology fundamentally different from all other radio frequency (RF) communications. UWB achieves wireless broadband communication without using a RF carrier. Instead, UWB is a sequence of very short electrical pulses, billionths of a second long, which exist not on any particular frequency but on all frequencies simultaneously. UWB uses modulated pulses with less one nanosecond in duration. The modulated pulse is usually assigned a digital representation of 0 or 1 to the transmitted and received pulse based on where the pulse is place in time. The key to turning the digital pulses into wireless broadband communication lies in the timing of the pulses. In order to hear the information in that code, a UWB receiver has to know the exact pulse sequence used by the transmitter.

[0004] Each pulse can exist simultaneously across an extensive band of frequencies if the distributed energy of the pulse at any given frequency exists in the noise floor. Therefore, UWB can co-exist with RF carrier-based communications with no discernable interference. This opens vast new communications with providing tremendous wireless bandwidth to ease the growing bandwidth crunch.

[0005] The U.S. Federal Communications Commission (FCC) on Feb. 14, 2002 authorized limited commercial use of wireless devices based on a communication technology called ultra wideband. The FCC's restrictions require that commercial ultra wideband devices must operate in radio spectrum in the frequency ranges from 3.1 GHz to 10.6 GHz. UWB communication devices should also satisfy by Part 15.209 rule, which set emission limits for operation.

[0006] UWB communication transceivers can transfer information data at rates of 100 mega-bit per second (Mbps) to 1 giga-bit per second (Gbps), with sending repeated ultra-short pulse signals across distances as great as 500 feet, even up to 2 kilometers.

[0007] With transmitting repeated ultra-short pulse signals for the high data rate up to 1 Gbps in the frequency ranges from 3.1 to 10.6 GHz, an analog-to-digital (A/D) converter should operate at very high sampling rate F_sso that UWB communication receiver can implement in a digital domain. Usually, the sampling rate F_smust be greater than two-time the highest frequency F_maxin the ultra-short pulse signals. This may lead to have a difficult problem to design an A/D converter with such high-speed operation in an UWB communication transceiver.

[0008] In addition, digital down conversion (DDC) should shift the bandpass ultra-shout pulse signals of the output of the A/D converter into the baseband signals and perform the decimation of the baseband signals with high sampling rate into low sampling rate due to the repeated pulse signals during transmitting.

[0009] Thus, there is a continuing need for an A/D converter bank with operating at a lower-speed to substitute a very-high-speed A/D converter along with a digital down conversion for a digital UWB communication transceiver.

[0010] In accordance with one aspect, an A/D converter bank based UWB receiver comprises a low noise amplifier (LNA) coupled to an anti-aliasing analog filter, an A/D converter bank with a sampling frequency rate of F_s, a digital down conversion coupled to a rake receiver and a channel estimate, a template pulse generator coupled to the rake receiver to calculate a correlation between a received monocycle pulse and an ideal monocycle pulse, a sequence generator coupled to the template pulse generator, a synchronization coupled to the template pulse generator, and a clock control coupled to the A/D converter bank, the digital down conversion and the template pulse generator.

[0011] Other aspects are set forth in the accompanying detailed description and claims.

[0012] FIG. 1 is a block diagram of showing one embodiment of an UWB communication transceiver in accordance with the present invention;

[0013] FIG. 2 is a block diagram of showing a transmitter section of the UWB communication transceiver as shown in FIG. 1;

[0014] FIG. 3 is a block diagram of the receiver section of the UWB communication transceiver as shown in FIG. 1;

[0015] FIG. 4 is a block diagram of showing one embodiment of the present invention of the A/D converter bank, including an analyzed sequence switch, a synthesized sequence switch, a set of A/D converters, a set of adaptive amplifiers, and a set of digital FIR filters, with operating in parallel;

[0016] FIG. 5 is a block diagram of showing one embodiment of the present invention of another A/D converter bank, including an analyzed sequence switch, a synthesized sequence switch, a set of A/D converters, a set of adaptive amplifiers, with operating in parallel, and one Mth-band digital FIR filter;

[0017] FIG. 6 is a block diagram of showing one embodiment of the present invention, including a complex multiplier, a complex oscillator, a decimation lowpass FIR filter H_N(z), a down sampling N selector, a clock control, and a set of down sampling with a selectable MUX function;

[0018] FIG. 7 is a block diagram of showing one embodiment of the present invention, including a complex multiplier, a complex oscillator, a decimation lowpass FIR filter H_D(z), a down sampling N selector, a clock control, and a set of down sampling with a selectable MUX function; FIG. 7 has the exact output comparing with the output as shown in FIG. 6, but the present invention as shown in FIG. 7 is different from the present invention in FIG. 6.

[0019] Referring to FIG. 1, an UWB communication transceiver 8 in accordance with one embodiment of the present invention includes a low-noise amplifier (LNA) and power amplifier (PA) section 10, which is coupled to transmitting and receiving antennas. The low-noise amplifier/power amplifier section 10 is coupled to an A/D and D/A converter section 12. The A/D and D/A converter section 12 is coupled to the digital signal processing section 14. The digital signal processing section 14 is coupled to a network interface section 16. The network interface 16 interfaces with Ethernet network 18. In accordance with one embodiment of the present invention, the system 8 is a so-called ultra wideband communication transceiver that both transmits and receives speech, audio, image, and video and data information by using a sequence of the ultra-short pulses.

[0020] A dedicated physical channel of showing transmitter in the UWB communication transceiver, as shown in FIG. 2, receives dedicated physical data channel 20 user data bits, such as information data at 1 Gbps. The information data 20 is passed through a one-second-rate convolution encoder 22 that may produce the double data rate of 2 giga-symbols per second (Gsps) by adding redundancy. The data is then interleaved and produced 2 Gsps by using a block interleaver 24. Thus, the output data symbols from the block interleaver 24 are modulated by using a pulse position modulation (PPM) 26, which is able to produce eight digital Gaussian-monocycle pulses based on one symbol data. The PPM technique 26 is to assign a time-window, and shift the position of the Gaussian-monocycle pulses within the window in time. The sequence generator 28 is a time-hopping encoding sequence generator. Using the sequence generator 28 and the clock control 32 controls the pulse generator 30 to produce the ultra-short Gaussian monocycle pulses and the pulse position in time for the PPM technique 26. The output ultra-short Gaussian-monocycle pulses from the PPM 26 are then passed through the pulse-shaping digital FIR filter 34 to create the frequency spectrum of the pulses in which can meet the FCC's requirements. Then, the output pulses from the pulse-shaping digital FIR filter 34 are used for the digital-to-analog (D/A) converter 36, which is operated at the sampling frequency of 22 gigahertz (GHz). The analog reconstruct filter 38, which is a bandpass filter, is used for reconstructing the analog ultra-short pulse signals. Thus, the analog pulse signals from the output of the analog reconstruct filter 38 is passed the power amplifier 40 through an antenna into air.

[0021] The transmitter in an UWB communication transceiver, as shown in FIG. 2, can also transmit the dedicated physical data channel 20 user data bits with scalability, such as information data of 50 Mbps, 100 Mbps, 200 Mbps, 250 Mbps, and 500 Mbps. In these cases, the PPM 26 produces 160, 80, 40, 32, 16 ultra-short Gaussian-monocycle pulses based on one symbol data for the information data of 50 Mbps, 100 Mbps, 200 Mbps, 250 Mbps, and 500 Mbps, respectively.

[0022] Referring to FIG. 3, which is the dedicated physical channel of showing the receiver in an UWB communication transceiver, the LNA 50 receives the ultra-short Gaussian-monocycle pulses from an antenna. The analog signals are passed through the analog anti-aliasing filter 52, which is a bandpass filter. The bandlimited analog signals are then sampled and quantized by using an A/D converter 54, with the sampling rate at 22 GHz. The digital bandpass signals of the output of the A/D converter 54 are then shifted into the baseband signals with down sampling by eight by using the digital down conversion (DDC) 56. Thus, the DDC 56 produces the digital data of 2 Gsps. Both the A/D converter 54 and the DDC 56 are controlled by the clock control 64. The output data from the DDC 56 is used for the channel estimate 62, and the rake receiver 58. The channel estimate 62 is used to estimate the channel phase and frequency that are passed into the rake receiver 58. The rake receiver 58 calculates the correlation between the received ultra-short pulses and the template pulses, which are generated by using the template pulse generator 66, and performs coherent combination. The template pulse generator 66 is controlled by three functions: clock control 64, sequence generator 68, and synchronization 70. The output of the rake receiver 58 is passed through the block de-interleaver 60. Thus, the output data of the block de-interleaver is used for the Viterbi decoder 72 to decode the encoded data and produce the information data of 1 Gbps.

[0023] The receiver in an UWB communication transceiver, as shown in FIG. 3, can also receive the symbol data with scalability to produce the information data of 50 Mbps, 100 Mbps, 200 Mbps, 250 Mbps, and 500 Mbps. In these cases, the DDC 56 has to shift the bandpass ultra-short pulse signals into baseband signals, and performs the down sampling by a factor of 160, 80, 40, 32, and 16, respectively.

[0024] An effective flexibility and scalability analog-to-digital (A/D) converter bank 54, as shown in FIG. 4, is used to substitute a very-high-speed A/D converter 54 in FIG. 3. The A/D converter bank 54, shown in FIG. 4, contains forty low-speed A/D converters 86a-86M, forty adaptive amplifiers 88a-88M, forty digital FIR filters 90a-90M, with operating in parallel, as well as one analyzed sequence switch 80, 82, 84, and one synthesized sequence switch 92, 94, 96.

[0025] The A/D converter bank 54, shown in FIG. 4, is to first decompose the analog bandpass signals into subband signals by using the analyzed sequence switch 80, 82, 84. The analyzed sequence switch 80, 82, 84 is a counterclockwise commutator-model circuit that is equivalent to polyphase implementation for downsampling. The analyzed sequence switch 80, 82, 84 rotates at uniform speed and takes on the forty positions in the way as shown in FIG. 4. The analog bandpass signals are achieved the downsampling of forty for each branch by using the analyzed sequence switch 80, 82, 84, and sampled by forty low-speed A/D converters 86a-86M in parallel. Each one of low-speed A/D converters 86a-86M is sampled at a sampling rate of 550 MHz with 8-bit resolution. Thus, these analog bandpass signals are converted into the digital subband signals, with non-overlapping frequency bands of bandwidth F_s/40. The digital subband signals are parallel passed through the forty different adaptive amplifiers 88a-88M, and forty different digital FIR filters 90a-90M. Then these digital subband signals are sequentially rotated for polyphase implementation of upsampling by using the synthesized sequence switch 92, 94, 96 to recover the desired sampling rate F_sand to obtain the digitally reconstructed signals.

[0026] The forty adaptive amplifiers 88a-88M are used for eliminating the gain error within the A/D converter bank due to the mismatch problem among the forty A/D converters 86a-86M, and for reducing the narrow interference as well due to other radio operation within the frequency band from 3.1 GH to 10.6 GHz.

[0027] As shown in FIG. 4, the adaptive amplifiers A_k(k=0, 1, 2, . . . 40) 88a-88M are mainly used to compensate and reduce the gain error among all of the A/D converters 86a-86M since all of the A/D converters 86a-86M are not exactly equal in practical implementation. Furthermore, K out of 40 adaptive amplifiers, where K is an integer and is less than 40, are setting to −50 dBm to reduce the narrow interference within the A/D converter bank, such as the interference of IEEE WLAN 802.1 lain the frequency ranges from 5.15 GHz to 5.35 GHz and from 5.725 GHz to 5.825 GHz.

[0028] For the digital filter bank R_k(z), (where k=0, 1, 2, . . . 40), 90a-90M, as shown in FIG. 4, the z-transform function X(z) is expressed in terms of {circumflex over (X)}(z) as follows: $\begin{array}{cc}\hat{X}\left(z\right)=\left(\frac{z^{-\left(M-1\right)}}{M}\right)\left(\sum _{i=0}^{M-1}X\left(z{W}_M^i\right)\right)\left(\sum _{k=0}^{M-1}{R}_k\left(z^M\right)W_M^{-ki}\right),& \left(1\right)\end{array}$

[0029] where W_M=e^−j2π/M. So, the A/D converter bank system 54 in FIG. 4 is alias-free if and only if, $\begin{array}{cc}\left(\sum _{k=0}^{M-1}{R}_k\left(z^M\right)W_M^{-ki}\right)=0,1\le i\le M-1.& \left(2\right)\end{array}$

[0030] Because of W_MW^*_M=MI, equation (2) can be rewritten as $\begin{array}{cc}\left[\begin{array}{c}{R}_0\left(z\right)\\ R_1\left(z\right)\\ ⋮\\ R_{M-1}\left(z\right)\end{array}\right]=W_M\left[\begin{array}{c}R\left(z\right)\\ 0\\ ⋮\\ 0\end{array}\right].& \left(3\right)\end{array}$

[0031] This implies that the A/D converter bank system 54 is aliasing free if

[0032] R_k(z)=R(z), for all of k.

[0033] Thus, under the condition of equation (4), the z-transform function {circumflex over (X)}(z) in FIG. 4 is given by

[0034] {circumflex over (X)}(z)=z^−(M−1)R(z^M)X(z),  (5)

[0035] where R(z^M) is a Mth frequency band digital FIR filter of R(z). Thus, for the A/D converter bank 54 as shown in FIG. 4, the aliasing is completely canceled. However, this A/D converter bank system 54 has amplitude distortion

[0036] T(z)=z^−(M−1)R(z^M).  (6)

[0037] The amplitude distortion can also be canceled if the filter R(z^M) is one allpass filter. The A/D converter bank system 54 does not have phase distortion since the digital filter R(z) is a FIR filter with a linear phase. In addition, the FIR filter R(z) is able to eliminate the narrow interference of other radio operations within each branch of the A/D converter bank 54 in FIG. 4.

[0038] In accordance with another embodiment of the present invention, instead of using the forty different digital FIR filters 90a-96M with operating in parallel for the A/D converter bank in FIG. 4, another present invention of this A/D converter bank system 54, as shown in FIG. 5, uses only one Mth frequency band digital FIR filter H(z^M) 116 after the synthesized sequence switch 110, 112, 114. Thus, memory of the filter coefficients 116 can be used in a minimum size. The architecture of the A/D converter bank 54 can be simplified. The power consumption of the A/D converter bank 54 can be reduced. This A/D converter bank system 54 is also aliasing free, no phase distortion, but has amplitude distortion as shown in equation (6). This A/D converter bank system 54 is also able to eliminate the narrow interference within each branch by using the Mth frequency band digital FIR filter H(z^M) 116.

[0039] Referring to FIG. 6, one embodiment of the present invention is called the digital down conversion (DDC) 56. The DDC 56 works by first shifting the ultra wideband signals with a frequency range from 3.1 GHz to 10.6 GHz of interest to baseband signals by using the complex multiplying 120 the received signals of the scalable A/D converter 54 by a complex oscillator 122. The baseband signals of output of the complex multiplier 120 are passed through the decimation lowpass FIR filter H_N(z) 124, which is controlled by the clock control 128 and is selected one of the down sampling N (N=8, 16, 32, 40, 80, 160) for the filter cutoff frequency by using the down sampling N selector 126. The output signals from the decimation lowpass FIR filter H_N(z) 124 are then passed one of the down sampling blocks 130a-130gthrough the selectable MUX 132, which is controlled by the clock control 128 and the down sampling N selector 126. Thus, the selectable MUX 132 produces one of the data rates of 1 Gbps, 500 Mbps, 250 Mbps, 200 Mbps, 100 Mbps, 50 Mbps based on the down sampling blocks 130b, 130c, 130d, 130e, 130f, 130g, respectively.

[0040] In accordance with another embodiment of the present invention 56, shown in FIG. 7, the down sampling blocks 144a-144gin the DDC 56 are implemented before the decimation lowpass filter FIR filter H_D(z) 152, which is designed to be a Nth-band decimation lowpass FIR filter H_D(Z^N). Thus, the operation of the decimation lowpass filter FIR filter H_D(z) 152 is based on the low sampling rate to achieve the efficient implementation.

[0041] Referring to FIG. 7, one embodiment of the present invention of the DDC 56 operates by first shifting the interested ultra wideband (3.1 GHz to 10.6 GHz) to the baseband signals by using the complex multiplying 140 the received signals of the scalable A/D converter 54 by a complex oscillator 142. The output baseband signals of the complex multiplier 140 are passed one of the down sampling blocks 144a-144gthrough the selectable MUX 146. The selectable MUX 146 is controlled by the clock control 150 and the down sampling N selector 148 for selecting one down sampling rate N, (including N=8, 16, 32, 40, 80, 160). Then, the output signals of the selectable MUX 146 are passed through the decimation lowpass FIR filter H_D(z) 152 to produce one of the band-limited signal with the data rates of 1 Gbps, 500 Mbps, 250 Mbps, 200 Mbps, 100 Mbps, 50 Mbps based on the down sampling blocks 144b, 144c, 144d, 144e, 144f, 144g, respectively.

[0042] While the present inventions have been described with respect to a limited number of embodiments, those skilled in the art will appreciate numerous modifications and variations therefrom. It is intended that the appended claims cover all such modifications and variations as fall within the true spirit and scope of these present inventions.