CMOS relaxation oscillator circuit with improved speed and reduced process/temperature variations

[0002] This application claims priority from provisional application entitled “A CMOS Relaxation Oscillator Circuit with Improved Speed and Reduced Process/Temperature Variations”, Application No. 60/341,481, filed Dec. 14, 2001.

[0003] 1. Technical Field

[0004] This invention relates to CMOS relaxation oscillator circuits. More particularly, this invention relates to high speed CMOS relaxation oscillator circuits with reduced dependence on process and temperature variations.

[0005] 2. Background

[0006] Oscillators are widely used throughout the microelectronics industry to provide a steady and stable periodic output waveform. For example, a typical oscillator could be used to generate the clock signals for digital applications. Another application where oscillator circuits are used is in the generation of an amplitude modulated carrier signal or a frequency modulated carrier signal for the transmission of data.

[0007] Oscillator circuits are generally broken down into two subclasses: tuned oscillators and relaxation oscillators.

[0008] Tuned oscillators have the advantage of high frequency stability, high frequency capability and higher spectral purity than relaxation oscillators. Tuned oscillators, however, require expensive and bulky components, such as inductors and crystals. Furthermore, these components are difficult to integrate into monolithic circuits. Additionally, tuned oscillators suffer from a narrow frequency band of operation.

[0009] Relaxation oscillators, however, have some unique advantages when used in integrated circuit design. For instance, relaxation oscillators do not require inductors or crystals. Thus, the relaxation oscillator can be easily incorporated into monolithic circuits. Additionally, the operative frequency band of a relaxation oscillator circuit is proportional to a charging current and inversely proportional to a voltage level across an energy storage device (e.g. capacitor) and the energy storage capacity of the energy storage device. The frequency of the relaxation oscillator circuit can easily be set using a single external component (e.g. a resistor) and can be varied linearly over a wide frequency band. A relaxation oscillator circuit, however, suffers from relatively poor frequency stability, particularly for high-speed operation. Furthermore, the relaxation oscillator circuit has a poor spectral purity characteristic.

[0010] Further details of the present invention are explained with the help of the attached drawings.

[0011] FIG. 1 depicts a relaxation oscillator circuit found in the prior art.

[0012] FIG. 2 depicts the voltage waveforms within the prior art relaxation oscillator circuit during the transitions between the two stable states.

[0013] FIG. 3 depicts an improved relaxation oscillator circuit with improved speed and reduced dependency on process and temperature variations.

[0014] FIG. 4 depicts the voltage waveforms within the improved relaxation oscillator circuit depicted in FIG. 3.

[0015] FIG. 1 depicts a relaxation oscillator circuit found in the prior art. The circuit is biased with a first voltage source 199 at a voltage potential of VSUP, and a second voltage source 198 at a voltage potential of VSUP2. The first voltage source 199 has a higher voltage potential than the voltage potential of the second voltage source 198 (VSUP>VSUP2). The value of VSUP2 can be zero volts. Connected between the first voltage source 199 and the second voltage source 198 is a right side of the relaxation oscillator circuit and a left side of the relaxation oscillator circuit.

[0016] The right side of the relaxation oscillator circuit includes a current source 101 connected in parallel with a diode connected transistor 110. Thus, a drain and a gate of the transistor 110 are connected together as well as being connected with the first voltage source 199. A first terminal of the current source 101 is connected to the first voltage source 199. The current source 101 is connected in such a manner as to source current in a direction that is generally from the first voltage source 199 and towards the second voltage source 198. A source of the transistor 110, a second terminal of the current source 101, a drain of a transistor 112, and the gate of a transistor 116 are connected together to form a first node, VOR. A source of the transistor 112 is connected with a first terminal of a capacitor 118 and a first terminal of a current sink 105 to form a second node, VSR. The second terminal of the current sink 105 is connected with the second voltage source 198, such that current flows into the second voltage source, 198.

[0017] The left side of the relaxation oscillator circuit essentially mirrors the right side of the relaxation oscillator circuit. Thus, the left side of the relaxation oscillator circuit includes a current source 103 connected in parallel with a diode connected transistor 114. Thus, a drain and a gate of the transistor 114 are connected together as well as being connected with the first voltage source 199. Additionally, the gate of the transistor 114 is connected with the gate of the transistor 110. A first terminal of the current source 103 is connected with the first voltage source 199. The current source 103 is connected in such a manner as to supply current in a direction that is generally from the first voltage source 199 towards the second voltage source 198. A source of the transistor 114, a second terminal of the current source 103, a drain of a transistor 116, and a gate of the transistor 112 are connected together to form a third node, VOL. A source of the transistor 116 is connected with a second terminal of the capacitor 118 and a first terminal of a current sink 107 to form a fourth node, VSL. A second terminal of the current sink 107 is connected with the second voltage source 198, such that current flows into the second voltage source 198.

[0018] The current sink 105 and the current sink 107 sinks current from a first gain stage that includes the transistor 112 and a second gain stage that includes the transistor 116. Additionally, the current sink 105 is typically set to sink the same amount of current as the current sink 107 and has a current value of ISINK.

[0019] The current source 101 and the current source 103 are non-ideal. Thus, they both have a zero amp output current when the voltage across their terminals is zero. If the voltage across their terminals is not zero, then the value of the current can be any arbitrary number from zero amps to two times ISINK. Typically, the current provided by the current source 101 is equal to the current provided by the current source 103 and has a current value of ISOURCE. Furthermore, ISINK is typically equal to ISOURCE.

[0020] For simplicity, the ratio of the transistor width to the transistor length is balanced along the different stages of the relaxation oscillator circuit. Thus, the width-to-length (W/L) of the transistor 110 is approximately equal to the W/L of the transistor 114. Likewise, the W/L of the transistor 112 is approximately equal to the W/L of the transistor 116. Thus, W/L(112)=W/L(116) and W/L(110)=W/L(114). Consequentially, the gate to source voltage, VGS, drop across the transistor 110 when it is on is equal to the VGS drop of the transistor 114 when it is on. The VGS associated with the transistor 110 or the transistor 114 is referred to as VGS1. Furthermore, the VGS drop across the transistor 112 when it is on is equal to the VGS drop of the transistor 116 when it is on. The VGS associated with the transistor 112 or the transistor 116 is referred to as VGS2.

[0021] In operation, the cross-coupling between the transistor 112 and the transistor 116 by the capacitor 118 creates a positive feedback around the loop of the two gain stages. Thus, the relaxation oscillator circuit will oscillate and the voltage amplitude of an output signal on the output nodes, VOL and VOR, will grow until the output signal is clipped. At any one time, it can be assumed that either the transistor 112 or the transistor 116 will be active, but not both. Thus, the capacitor 118 is alternatively charged with equal but opposite currents. One current that charges the capacitor 118 originates from the transistor 112 when it is active and another current originates from the transistor 116 when it is active. The source of the charging current for the capacitor 118 alternates in half cycles of the relaxation oscillator circuit operation.

[0022] During one half cycle of the relaxation oscillator circuit operation, the transistor 116 is supposed to be off and the transistor 112 is on. Meanwhile, the voltage at the third node, VOL, is equal to the voltage potential of the first voltage source 199, or [VOL=VSUP]. The first node, VOR, however, is clamped to a voltage potential equal to the first voltage source199, less VGS1, the gate to source voltage drop across the transistor 110, or [VOR=VSUP−VGS1]. The source current of the transistor 110 is equal to the sum of the current sink 105 and the current sink 107 less the current source 101, or [2*ISINK−ISOURCE=ISINK]. The transistor 112 is conducting a current equal to the sum of the current sink 105 and the current sink 107 or [2*ISINK]. The voltage at the second node, VSR, is clamped to the voltage at the third node, VOL, less VGS2, the gate to source voltage of the transistor 112 or [VSR=VOL−VGS2=VSUP−VGS2]. It can be assumed that at the beginning of the half cycle, the voltage at the fourth node, VSL, is equal to the voltage at the node of VSR. Since the transistor 116 is off and the transistor 112 is on with a current of 2*ISINK, there is a current of ISINK flowing from the node VSR to the node VSL. The charging current through the capacitor 118 will cause the voltage level at the node, VSL, to ramp down with a constant slew rate of ISINK/C1, where C1 is the capacitance value of the capacitor 118. When this voltage ramps to [VSUP−VGS1−VGS2], the transistor 116 is turned on like a switch. At this point, the positive feedback around the loop of the two stages will force the transistor 112 to be off like a switch. The output voltage at the third node, VOL, will be pulled down to VSUP−VGS1 and the output voltage of the first node, VOR, is regenerattively pulled up to VSUP. The voltage at the node VSL is restored to be [VSUP−VGS2]. As a result, the circuit has now changed its state and the capacitor 118 discharges in the opposite direction with a constant current equal to ISINK. The circuit maintains this state until the voltage level at the source of the transistor 112 ramps downward to [VSUP−VGS1−VGS2], causing the transistor 112 to turn on and switch the circuit to its prior state, whereby the cycle repeats itself.

[0023] FIG. 2 depicts the voltage waveforms within the prior art relaxation oscillator circuit during transitions between two stable states. The voltage level at the source of the off-transistor (transistor 116 or 112, which is off) is linearly ramping in a negative direction with a slew rate of ISINK/C1, for a total voltage swing of (VSUP−VGS2)−(VSUP−VGS1−VGS2)=VGS1 during each half-cycle. Recall that VGS1 is the gate to source voltage across either transistor 110 or transistor 114, whichever is on. Each half-cycle of oscillation is equal to the time T/2 for the voltage across capacitor 118 to ramp down by an amount equal to VGS1. Thus, T/2=C1*VGS1/ISINK. The total period of oscillation is T and the frequency of oscillation can be expressed as [F=1/T=ISINK/(2*C1*VGS1)]. Thus, the frequency is proportional to the charging current of ISINK, and is inversely proportional to the value of the timing of capacitor 118 and VGS1, which is the voltage across the diode connected transistors, transistor 110 and transistor 114 when they are on with a source current of ISOURCE=ISINK, and the voltage swing across the capacitor 118.

[0024] By neglecting the body effect of the transistors for simplicity, VGS1 can be expressed as VGS1=Vt0+((2* ISINK/k){circumflex over ( )}½). In this equation, Vt0 is the threshold voltage and k is the device transconductance of either transistor 110 or transistor 114, depending on which transistor is conducting. Both Vt0 and k are determined by the process used for the design of the relaxation oscillator circuit. The frequency dependence on VGS1 prevents the use of ISINK to compensate for process and temperature variations. Thus, the frequency will be unstable over process and temperature. Additionally, VGS1 has a minimum value of Vt0 when ISINK is zero. Consequentially, a certain amount of current, ISINK, is required for a minimum operating frequency. Moreover, VGS1 increases linearly in proportion to the square root of ISINK. This relationship between VGS1 and ISINK forces the oscillator to need more charging current for higher speed operation due to the increased signal swing.

[0025] FIG. 3 depicts modifications to the circuit of FIG. 1 to create an improved relaxation oscillator circuit with improved speed and reduced dependency on process variations and temperature. Components carried over from FIG. 1 to FIG. 3 are similarly labeled for convenience. In FIG. 3, the transistor 110 and the transistor 114 are no longer diode connected. Thus, the gate of these two transistors are not connected with the voltage source 199. Rather, the gate of transistor 110 is connected with the gate of transistor 114, the gate of an additional transistor 130 and the output of an amplifier 132. In addition to the previously identified connections to the voltage source 199, the drain of the transistor 130, a first terminal of a current source 138, and a first terminal of voltage source 134 are connected to the voltage source 199. A second terminal of current source 138 is connected with the source of the transistor 130, an inverting input to the amplifier 132, and a first terminal of a current sink 136. A second terminal of voltage source 134 is connected to a non-inverting input to amplifier 132. Lastly, a second terminal of current sink 136 is connected to the voltage source 198.

[0026] The reference voltage source 134, the amplifier 132 together with the transistor 130, the current source 138 and the current sink 136 provide proper biasing for the gates of the transistor 114 and the transistor 110. The current source 138 has the same properties as the current source 103 and the current source 101. Thus, the current source 138 is non-ideal and requires a voltage across its terminals for it to generate a current. Also, current source 138 provides the same current as the current source 103 and the current source 101, that being ISOURCE. The current sink 136 provides a current equal to two times ISINK or [2*ISINK]. The voltage source 134 has a voltage value of VCLMP. Lastly, the W/L ratio of the transistor 130 is equal to W/L ratio of the transistor 114 and the transistor 110, [W/L(130)=W/L(114)=W/L(110)].

[0027] The relaxation oscillator circuit of FIG. 3 performs similarly to the relaxation oscillator circuit of FIG. 1, with a few exceptions. For instance, the value of voltage source 134, VCLMP, now controls the voltage at the node VOR to be VSUP−VCLMP when transistor 112 is on (same for the voltage at the node VOL when transistor 116 is on), rather than the gate-to-source voltage drop of the transistor 110 setting the voltage at the node VOR to be VSUP−VGS1 (same for the node VOL). As the circuit settles, the voltage difference between the non-inverting and the inverting terminals of the amplifier 132 seek equilibrium. Thus, the voltage at the source of the transistor 130 settles to a value equal to VSUP−VCLMP. The voltage on the gate and source of transistor 114 and transistor 110 (when they are on) mirror the voltage values on the gate and source of the transistor 130, to make VOR=VSUP−VCLMP and VOL=VSUP−VCLMP.

[0028] While the transistors shown in FIG. 3 are NMOS transistors, these transistors can be replaced with PMOS transistors. Thus, a drain-source current path is formed in the transistors. Current in the drain-source path is controlled at a transistor control input (e.g. transistor gate). Furthermore, the transistors shown in FIG. 3 can be BJT transistors, either npn BJTs or pnp BJTs. If BJTs are used, then a collector-emitter path is formed. Current in the collector-emitter path is set using a transistor control input (e.g. transistor base). Collectively, a transistor can be described as comprising a first terminal coupled with a second terminal and a control terminal coupled between the first terminal and the second terminal.

[0029] FIG. 4 depicts the voltage waveforms within the improved relaxation oscillator circuit depicted in FIG. 3. In operation, the voltage at the first node, VOR, varies between VSUP and (VSUP−VCLMP); the voltage at the second node, VSR, varies between (VSUP−VGS2) and (VSUP−VCLMP−VGS2); the voltage at the third node, VOL, varies between VSUP and (VSUP−VCLMP); and, the voltage at the fourth node, VSL, varies between (VSUP−VGS2) and (VSUP−VCLMP−VGS2). Furthermore, the voltage at node VSR and the node VSL continue to ramp down with a slew rate of ISINK/C1, where C1 is the value of capacitor 118. From these voltage characteristics of the relaxation oscillator circuit in FIG. 3, it can be seen that the voltage source 134 with a value of VCLMP can be used to control the voltage variation on node VOR and node VOL. Also, each half-cycle of oscillation is equal to the time T/2 for the voltage across capacitor 118 to ramp down by a voltage equal to VCLMP, or T/2=C1*VCLMP/ISINK. Thus, the total period of oscillation is T and the frequency of oscillation can be expressed as F=1/f=ISINK/(2*C1*VCLMP). Hence, the frequency is proportional to the charging current of ISINK, and inversely proportional to the value of the timing capacitor 118 and the voltage swing across the capacitor, VCLMP.

[0030] The value of VCLMP can be designed to be less than the threshold voltage, Vt0, therefore less than the VGS1 of the relaxation oscillator circuit depicted in FIG. 1 for any charging current ISINK. Thus, the improved relaxation oscillator circuit of FIG. 3 will have a higher speed than the relaxation oscillator circuit of FIG. 1 for the same charging current ISINK and the same timing capacitor 118. Another way to describe the improvement is to say that for the same speed and capacitor, the improved relaxation oscillator circuit of FIG. 3 consumes less current than the relaxation oscillator circuit of FIG. 1. The amplifier 132 can also be designed to have a higher supply voltage than the voltage source 199, or VSUP. Hence, the amplifier output can exceed the voltage source 199; and, the gate voltage of the transistor 110, the transistor 114 and the transistor 130 can also be higher than the value of the voltage source 199, or VSUP.

[0031] In addition to the above-described improvements, the value of VCLMP is independent of the value of ISINK. Hence, the frequency increases linearly with ISINK in the relaxation oscillator circuit of FIG. 3 rather than increasing with the square root of ISINK as the relaxation oscillator circuit of FIG. 1 behaves. Consequentially, the frequency tuning range of the relaxation oscillator circuit of FIG. 3 is improved over the prior art.

[0032] The value of VCLMP allows a frequency to be stable irrespective of process and temperature variations. Thus, voltage source 134 can be a neighboring circuit on an integrated circuit that includes the relaxation oscillator circuit of FIG. 3. Additionally, the value of VCLMP can be set such that it is proportional to a reference voltage (Vref) on an integrated circuit that includes the relaxation oscillator circuit of FIG. 3. For instance, VCLMP can be proportional to a bandgap voltage. Furthermore, VCLMP can equal k times Vref, where k is constant. Moreover, ISINK can be obtained from the reference voltage, Vref, and a precision resistor such as an off-chip resistor, Rext. Thus, ISINK=Vref/Rext. If such a configuration is used, then the frequency would be equal to, F=1/(2*k*C1*Rext). Thus, the frequency can be inversely proportional to the value of an on-chip capacitor 118 and an off-chip resistor, Rext. The reference voltage, Vref, is no longer part of the frequency expression. Since an off-chip resistor can be more resistant to process and temperature variations, the improved relaxation oscillator circuit of FIG. 3 will have better process and temperature independence over the relaxation oscillator circuit of FIG. 1.