Receiving circuit

[0002] This application is a divisional of application Ser. No. 08/682,717 filed Jul. 29, 1996 which is now U.S. Pat. No. 5,914,980 which is a 371 application of PCT/JP95/02448 filed Nov. 30, 1995.

[0003] 1. Field of the Invention

[0004] The present invention relates to a receiving circuit, and more particularly to a receiving circuit which permits simplification of its circuit arrangement and which further allows less power for a receiving system concurrent with reduction of power consumption.

[0005] 2. Description of the Prior Art

[0006] One of important points needed for a receiving circuit in communications is to possibly reduce high-frequency circuit portions to thereby lessen a high power consumption factor, unstable operation factor and high manufacturing cost inherent in the high-frequency circuit, coupled with a space occupied by that circuit. These points are particularly important for mobile communication systems and others. For the reduction of the high-frequency circuit portions, a direct demodulation method has been proposed heretofore in terms of multiple frequency conversion and carrier frequency, thus accomplishing the direct conversion into a low frequency and the direct demodulation into a baseband. The aforesaid high-frequency circuit portion mainly constitutes a space diversity reception function necessitating two antenna systems.

[0007] As the direct demodulation method, there have been developed a number of methods in which a local oscillator generates a signal with a frequency equal to the carrier frequency which in turn, is mixed with a received input wave to derive a baseband signal therefrom. This direct demodulation method is made to produce a high-frequency signal with a frequency equal to the received signal frequency, and the high-frequency signal can be easy to release or emit in the air through an antenna of the receiver. Accordingly, another receiver adjacent thereto undergoes interferences to be inhibited from establishing communications. For this reason, this method has chiefly been adopted for communications based on frequency modulation methods which are relatively strong to single frequency interferences.

[0008] On the other hand, radio portable telephones, being recently rapidly put in widespread use, rely on a so-called PSK which is one amplitude transport modulation method, and the single frequency interference produces an offset in a demodulated output to deteriorate the error rate of a received signal. That is, since the local oscillation frequency can not take the carrier frequency, in this kind of communication method difficulty is encountered in direct frequency conversion and direct demodulation. One approach to resolve such a technical problem involves a method which, if a carrier frequency for a radio portable telephone is taken as fc and an offset frequency is taken to be fo, obtains fc+fo and fc−fo in order to provide a frequency-offset complementary local oscillation frequency for a frequency conversion. For carrying out this method, fc+fo and fc−fo are obtainable by multiplication process of fc and fo through a mixer (frequency mixer), while both signals fc+fo and fc−fo coexist in an output. More specifically, although the aforesaid process independently requires signals with the respective frequencies, the prior system can not practically satisfy this requirement. The prior system essentially employs filters for the respective frequencies, while suffering a disadvantage that a carrier frequency for a desired signal is variable and hence the filters need to designed to cope with the frequency variations.

[0009] The present invention has been developed with a view to resolving the prior problems, and it is therefore a general object of the present invention to provide a receiving circuit in a communication system with a plurality of channels, which is capable of lessening the power necessary in a receiving system and of reducing the power consumption with a simplified circuit.

[0010] Another general object of this invention is to provide a receiving circuit which can provide the frequencies fc+fo and fc−fo for elimination of the problems related to the aforementioned ordinary methods.

[0011] More specifically, an object of this invention is to provide a receiving circuit which performs a direction frequency conversion of a frequency in between the channels of a receiving system as a local frequency of a receiver and prevents the occurrence of the frequency offset in its output signal and the mixing of a signal of the adjacent channels thereinto.

[0012] In addition, a further object of this invention is to structurally review functional parts making up a receiving circuit to reduce the functional parts suffering from a large power consumption or replace them with different ones.

[0013] For these purposes, as one example, a receiving circuit according to this invention is composed of first and second frequency converting circuits for accepting a received signal obtained through an antenna, a local frequency signal generating circuit for generating middle (intermediate) frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to said first frequently converting circuit and for outputting the downside frequency thereof as a conversion frequency input to said second frequency converting circuit, a common wave extracting circuit for extracting a component present in common in both outputs of the first and second frequency converting circuits, a frequency offset circuit for removing a frequency offset remaining in an output of the common wave extracting circuit, and a filter for removing an unnecessary frequency component remaining in an output of the frequency offset circuit.

[0014] In addition, in this invention, a means to resolve the prior problems is added to a local frequency complementary offset type direct frequency converting system, which bases this invention. A space diversity function can be realized with a receiving circuit based on a single direct quadrature detector.

[0015] With this arrangement, a received signal obtained from an antenna is fed to the first and second frequency converting circuits, while the local frequency signal generating circuit supplies, to the first and second frequency converting circuits, two different frequencies, i.e., the upside and downside frequencies corresponding to central values between channels, thus producing two output signals in terms of each of a desired wave, an upside channel and a downside channel. Further, a desired wave channel signal being a signal component present in common in both the first and second frequency converting circuits is extracted in the common wave extracting circuit. Since a frequency offset of ωo remains in the output of the common wave extracting circuit, a minute frequency conversion is carried out in the offset frequency circuit to remove the offset amount. Moreover, an unnecessary frequency component generated in this process is removed through a filter, before the resultant signal is supplied as a baseband signal to a baseband signal processing section.

[0016] In accordance with a preferred form of this invention, a receiving circuit comprises first and second frequency converting circuits for accepting a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second frequency converting circuits for generating middle frequencies between a ratio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit and for outputting the downside frequency thereof as a conversion frequency input to the second frequency converting circuit, a common wave extracting circuit for extracting a component present in common in both outputs of the first and second frequency converting circuits, a frequency offset circuit for removing a frequency offset remaining in an output o the common wave extracting circuit, and a filter for removing an unnecessary frequency component remaining in an output of the frequency offset circuit.

[0017] In another preferred form of this invention, a receiving circuit comprises first and second frequency converting circuits for accepting a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second frequency converting circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit and for outputting the downside frequency thereof as a conversion frequency input to the second frequency converting circuit, a first frequency offset circuit for removing a frequency offset contained in an output of the first frequency converting circuit, a second frequency offset circuit for removing a frequency offset contained in an output of the second frequency converting circuit, a common wave extracting circuit for extracting a component present in common in both outputs of the first and second frequency offset circuits, and a filter for removing an unnecessary frequency component left in an output of the common wave extracting circuit.

[0018] In a further preferred form of this invention, a receiving circuit comprises first and second frequency converting circuits for accepting a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second frequency converting circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit and for outputting the downside frequency thereof as a conversion frequency input to the second frequency converting circuit, first quantizing means for quantizing an output of the first frequency converting circuit, second quantizing means for quantizing an output of the second frequency converting circuit, a common wave extracting circuit for extracting a component present in common in both outputs of the first and second quantizing means, a frequency offset circuit for removing a frequency offset remaining in an output of the common wave extracting circuit; and a filter for removing an unnecessary frequency component left in an output of the frequency offset circuit.

[0019] In a further preferred form of this invention, a receiving circuit comprises first and second frequency converting circuits for accepting a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second frequency converting circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit and for outputting the downside frequency thereof as a conversion frequency input to the second frequency converting circuit, first quantizing means for quantizing an output of the first frequency converting circuit, second quantizing means for quantizing an output of the second frequency converting circuit, a first frequency offset circuit for removing a frequency offset contained in an output of the first quantizing means, a second frequency offset circuit for removing a frequency offset contained in an output of the second quantizing means, a common wave extracting circuit for extracting a component present in common in both outputs of the first and second frequency offset circuits, and a filter for removing an unnecessary frequency components left in an output of the common wave extracting circuit.

[0020] In a further preferred form of this invention, a receiving circuit comprises first and second quadrature (orthogonal) demodulating circuits for receiving a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second quadrature demodulating circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating circuit and for outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit, a first common wave extracting circuit coupled to the first and second quadrature demodulating circuits for extracting a component present in common in both I outputs of the fist and second quadrature demodulating circuits, a second common wave extracting circuit coupled to the first and second quadrature demodulating circuits for extracting a component present in common in both a Q output of the first quadrature demodulating circuit and a Q output of the second quadrature demodulating circuit reversed in polarity, a first frequency offset circuit for removing a frequency offset remaining in the I output extracted through the first common wave extracting circuit, a second frequency offset circuit for removing a frequency offset remaining in the Q output extracted through the second common wave extracting circuit, a first filter for removing an unnecessary frequency component remaining in an output of the first frequency offset circuit, and a second filter for removing an unnecessary frequency component remaining in an output of the second frequency offset circuit.

[0021] In a further preferred form of this invention, a receiving circuit comprises first and second quadrature demodulating circuits for receiving a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second quadrature demodulating circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating circuit and for outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit, first and second frequency offset circuits for removing a frequency offset contained in common in I outputs and Q outputs of the fist and second quadrature demodulating circuits, a first common wave extracting circuit for extracting a component present in common in both the I outputs of the first and second frequency offset circuits, a second common wave extracting circuit for extracting a component present in common in both the Q output o the first frequency offset circuit and the Q output of the second frequency offset circuit 6I reversed in polarity, and first and second filters for removing unnecessary frequency components remaining in outputs of the first and second common wave extracting circuits.

[0022] In a further preferred form of this invention, a receiving circuit comprises first and second quadrature demodulating circuits for receiving a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second quadrature demodulating circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating circuit and for outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit, first and second quantizing means for quantizing an I output and a Q output of the first quadrature demodulating circuit, respectively, third and fourth quantizing means for quantizing an I output and a Q output of the second quadrature demodulating circuit, respectively, a first common wave extracting circuit for extracting a component present in common in the I outputs of the first and third quantizing means, a second common wave extracting circuit for extracting a component present in common in the Q output of the second quantizing means and the Q output of the fourth quantizing means reversed in polarity, a first frequency offset circuit for removing a frequency offset remaining in the I output extracted through the first common wave extracting circuit, a second frequency offset circuit for removing a frequency offset remaining in the Q output extracted through the second common wave extracting circuit, a first filter for removing an unnecessary frequency component left in an output of the first frequency offset circuit, and a second filter for removing an unnecessary frequency component left in an output of the second frequency offset circuit.

[0023] In a further preferred form of this invention, a receiving circuit comprises first and second quadrature demodulating circuits for receiving a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second quadrature demodulating circuits for generating middle frequency between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating a circuit and for outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit, first and second quantizing means for quantizing an I output and a Q output of the first quadrature demodulating circuit, respectively, third and fourth quantizing means for quantizing an I output and a Q output of the second quadrature demodulating circuit, respectively, a frequency offset circuit for removing frequency offsets remaining in the I outputs and the Q outputs of the first to fourth quantizing means, a first common wave extracting circuit for extracting a component present in common in the I outputs of the frequency offset circuit, a second common wave extracting circuit for extracting a component present in common in the Q output and a polarity-reversed Q output of the frequency offset circuit, a first filter for removing an unnecessary frequency component left in the I output extracted through the first common wave extracting circuit, and a second filter for removing an unnecessary frequency component left in the Q output of the second common wave extracting circuit.

[0024] Furthermore, according to this invention, in a receiving circuit, a correlator for calculating a mutual correction is used in placed of the common wave extracting circuit.

[0025] In a receiving circuit according to this invention, only the first frequency converting circuit is used for accepting the received signal, and after a frequency conversion by the first frequency converting circuit, a frequency-converted output of a side where no frequency conversion is effected by the first frequency converting circuit is obtained by another frequency converting circuit to attain two frequency-converted outputs necessary for the common wave extraction.

[0026] According to this invention, in a receiving circuit, only the first frequency converting circuit is used for accepting the received signal and only one of the first and second quantizing means is used for quantization, and after the quantization by the quantizing means, a frequency-converted output of a side where no frequency conversion is performed by the first frequency converting circuit is obtained by the digital frequency converting circuit to obtain two frequency-converted digital outputs necessary for the common wave extraction.

[0027] According to this invention, in a receiving circuit, only one of the first and second quadrature demodulating circuits is used for accepting the received signal, and two outputs of the quadrature demodulating circuit are frequency-converted by the frequency converted circuit to obtain a frequency-converted output of a side where no quadrature demodulation is made, to obtain two quadrature-demodulating outputs necessary for the common wave extraction.

[0028] According to this invention, in a receiving circuit, only one of the first and second quadrature demodulating circuits is used for accepting the received signal and only the second and third quantizing means are used, and after the quantization by the quantizing means, two outputs of the quantizing means are frequency-converted by the digital frequency converting circuit to obtain a frequency-converted output of a side no quadrature demodulation is effected, to obtain two quadrature-modulated outputs necessary for the common wave extraction.

[0029] Furthermore, in a preferred form of this invention, a receiving circuit comprises first and second frequency converting circuits for accepting a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the first and second frequency converting circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit 2 and for outputting the downside frequency thereof as a conversion frequency input to the second frequency converting circuit, first and second integrating circuits, serving additionally as low-pass filters, for receiving outputs of the first and second frequency converting circuits through input lines, first and second buffer amplifiers for receiving outputs of the first and second integrating circuits, first and second transformers having the same construction and having primary coils whose one ends accepts outputs of the first and second buffer amplifiers, the other ends of the primary coils of the first and second transformers being grounded in the form of alternating currents, and secondary coils of the first and second transformers being coupled in parallel to each other in accordance with polarities of the primary coils and one ends of the secondary coils equal in polarity to the primary coils is used as an output terminal while the other ends of the secondary coils are grounded, a third buffer amplifier for accepting an output from the first and second transformers, a frequency offset circuit for removing a frequency offset remaining in an output of the third buffer amplifier, and a filter for removing an unnecessary frequency component remaining in an output of the frequency offset.

[0030] In a further preferred form of this invention, a receiving circuit comprises first and second frequency converting circuits for accepting a received signal obtained through an antenna, a local frequency signal generating circuit coupled to the fist and second frequency converting circuits for generating middle frequencies between a radio carrier frequency of the received signal and radio carrier frequencies of adjacent upside and downside channels and further for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit and for outputting the downside frequency thereof as a conversion frequency input to the second frequency converting circuit, first and second differential amplifiers for accepting outputs of the first and second frequency converting circuits through input lines, first and second integrating circuits, serving additionally as low-pass filters, for correspondingly accepting outputs of the first and second differential amplifiers, first and second buffer amplifiers for supplying outputs of the first and second integrating circuits, means for making a feedback from an output side of each of the first and second buffer amplifiers to a negative input side thereof, first and second transformers having the same construction and having primary coils whose one ends accepts outputs of the first and second buffer amplifiers, the other ends of the primary coils of the first and second transformers being at least grounded in the form of alternating currents, and secondary coils of the first and second transformers being coupled in parallel to each other in accordance with polarities of the primary coils and one ends of the secondary coils equal in polarity to the primary coils is used as an output terminal while the other ends of the secondary coils are at least grounded in the form of alternating currents, a third buffer amplifier for accepting outputs of the first and second transformers, means for comparing an output of the third buffer amplifier with an average of the output of the first frequency converting circuit and the output of the second frequency converting circuit to correct the outputs of the first and second frequency converting circuits, means for causing an output of the third buffer amplifier to be feedbacked to one of the first and second frequency converting circuits to correct a difference between the outputs of the first and second frequency converting circuits, a frequency offset circuit for removing a frequency offset remaining in the output of the third buffer amplifier, and a filter for removing an unnecessary frequency component remaining in an output of the frequency offset circuit.

[0031] In accordance with this invention, in receiving circuit, the one end of the secondary coil different in polarity from the primary coil is connected with the third buffer amplifier in place of the end end of the secondary coil equal in polarity to the primary coil being connected therewith.

[0032] In accordance with this invention, in a receiving circuit, the one end of the secondary coil different in polarity from the primary coil is connected with the third buffer amplifier in place of the end end of the secondary coil equal in polarity to the primary coil being connected therewith.

[0033] A receiving circuit according to this invention further comprises local frequency generating means including a first frequency signal source for generating a signal with a frequency equal to a carrier frequency of a desired receiving signal, first means for accepting a signal from the first frequency signal source to phase-shift the accepted signal by a phase amount of π/2 with respect to a frequency of the accepted signal, a second frequency signal source for generating a frequency signal equal to ½ of a frequency channel spacing, second means for accepting a signal from the second frequency signal source to phase-shift a frequency of the accepted signal by a phase amount of π/2, a first quadrature modulator comprising a multiplier for receiving the two signals from the first and second frequency signal sources and a multiplier for receiving two signals from the first and second phase-shifting means, and a second quadrature modulator comprising two multipliers for receiving one of the two signals from the first and second frequency signal sources through the phase-shifting means.

[0034] A receiving circuit according to this invention further comprises local frequency generating means including a first frequency signal source for generating a frequency signal equal to a carrier frequency of a desired receiving signal, first means for accepting a signal from the first frequency signal source to phase-shift the accepted signal by a phase amount of π/2 with respect to a frequency of the accepted signal, a second frequency signal source for generating a frequency signal equal to ½ of a frequency channel spacing, second means for accepting a signal from the second frequency signal source to phase-shift a frequency of the accepted signal by a phase amount of π/2 multipliers for receiving the two signals from the first and second frequency signal sources, and means for inverting the polarity of an output of one of the multipliers to add the polarity-inverted output to an output of the other multiplier.

[0035] A receiving circuit according to this invention further comprises local frequency generating means including a frequency signal source for generating a frequency signal equal to a carrier frequency of a desired receiving signal, means for accepting a signal from the frequency signal source to phase-shift the accepted signal by a phase amount of π/2 in relation to a frequency of the accepted signal, a quadrature modulator comprising two multipliers for accepting a frequency signal from the frequency signal source through the phase shifting means, and means for inverting the polarity of an output of one of the two multipliers and for adding the polarity-inverted output to an output of the other multiplier.

[0036] In accordance with this invention, a receiving circuit includes means for analog-to-digital-converting first and second received signals obtained through one of the frequency converting circuits and the quadrature demodulating circuits, first and second Fourier transformers for accepting digital outputs of the analog-to-digital converting means, respectively, a correlator for accepting outputs of the first and second Fourier transformers at every frequency component, a weighting function device for accepting an output of the correlator, a weighting value multiplier for accepting an output of the weighting function device, an adder for accepting the outputs of the first and second Fourier transformers, and means for inputting an addition result of the adder to the multiplier, and an inverse Fourier transformer for accepting an output of the weighting value multiplier, wherein an inverse Fourier transformed output of the inverse Fourier transformer is used as a desired wave extraction result.

[0037] Moreover, in a preferred form of this invention, a receiving circuit comprises received signal inputting means for receiving a received signal from an antenna, a quadrature demodulator for carrying out a frequency converting process of the received signal from the received signal inputting means, first and second analog-to-digital converters for accepting an output signal of the quadrature demodulator to convert its analog signal into a digital signal, a sampling clock generator for generating, to the first and second analog-to-digital converters, a clock with a frequency which is more than two times a frequency corresponding to a bandwidth of the received signal, an adding circuit for adding a delayed pulse train (sequence) to a pulse train from the sampling clock generator, means for providing the pulse train from the sampling clock generator and the delayed pulse train as an sampling pulse for the analog-to-digital converters, and means for extracting a quadrature component of a desired receiving channel signal from digital output data of the analog-to-digital converters.

[0038] In a further preferred form of this invention, a receiving circuit comprises received signal inputting means for receiving a received signal from an antenna, a quadrature demodulator for carrying out a frequency converting process of the received signal from the received signal inputting means, first and second analog-to-digital converters for accepting an output signal of the quadrature demodulator to convert its analog signal into a digital signal, a sampling clock generator for generating, to the first and second analog-to-digital converters, a clock with a frequency higher than a frequency corresponding to a bandwidth of the received signal, an adding circuit for adding a delayed pulse train to a pulse train from the sampling clock generator, means for providing the pulse train from the sampling clock generator and the delayed pulse train as an sampling pulse for the analog-to-digital converters, and means for extracting a quadrature component of a desired receiving channel signal from digital output data of the analog-to-digital converters, wherein one or more delayed pulse trains is generated so that the pulse train from the sampling clock generator and the delayed pulse train from the delayed pulse train adding circuit take a delayed time other than a phase difference corresponding to π of a frequency of the desired channel signal.

[0039] In accordance with this invention, in a receiving circuit, the pulse train is generated from the sampling clock generator and the delayed pulse train taking a phase difference time corresponding to π/2 relative to a frequency of the desired channel signal is generated from the delayed pulse train adding circuit.

[0040] In a further preferred form of this invention, a receiving circuit comprises received signal inputting means for receiving a received signal from an antenna, first and second analog-to-digital converters for accepting the received signal to convert the received signal into a digital signal, a sampling clock generator for generating, to the first and second analog-to-digital converters, a clock with a frequency higher than a frequency corresponding to a bandwidth of the received signal, an adding circuit for adding a delayed pulse train to a pulse train from the sampling clock generator, means for providing the pulse train from the sampling clock generator and the delayed pulse train as an sampling pulse for the analog-to-digital converters, and means for extracting a desired receiving channel signal from digital output data of the analog-to-digital converters.

[0041] Furthermore, in accordance with this invention, in a receiving circuit, in addition to the pulse train from the sampling clock generator, the delayed pulse train taking a phase difference time corresponding to π/2 with respect to a frequency of the desired channel signal is generated from the delayed pulse train adding circuit.

[0042] Moreover, in accordance with this invention, in a receiving circuit, in addition to the pulse train from the sampling clock generator, a polarity of delayed pulse trains taking a phase difference time corresponding to π/2 in relation to a frequency of the desired channel signal are generated from the delayed pulse train adding circuit.

[0043] In a further preferred form of this invention, a receiving circuit comprises received signal inputting means for receiving a received signal from an antenna, an analog-to-digital converter for accepting the received signal to convert the received signal into a digital signal, a sampling clock generator for generating, to the analog-to-digital converter, a clock with a frequency higher than a frequency corresponding to a bandwidth of the received signal, an adding circuit for adding a delayed pulse train to a pulse train from the sampling clock generator, means for providing the pulse train from the sampling clock generator and the delayed pulse train as an sampling pulse for the analog-to-digital converter, and means for extracting a desired receiving channel signal from digital output data of the analog-to-digital converter.

[0044] In a still preferred form of this invention, a receiving circuit comprises received signal inputting means for accepting received signals from a plurality of antennas, first and second frequency converting means for accepting the received signals, a local oscillator for supplying, to the first and second frequency converting means, an output at a frequency produced by providing a frequency offset of ½ of a channel spacing frequency to a desired carrier frequency, first and second analog-to-digital converters for accepting signals from the first and second frequency converting means, a sampling clock generator for generating, to the analog-to-digital converters, a clock with a frequency higher than a frequency corresponding to a bandwidth of the received signal, an adding circuit for adding a delayed pulse train to a pulse train from the sampling clock generator, means for providing the pulse train from the sampling clock generator and the delayed pulse train as an sampling pulse for the analog-to-digital converters, and means for extracting a desired receiving channel signal from digital output data of the analog-to-digital converters.

[0045] In accordance with this invention, in a receiving circuit, the local oscillator for providing the output to the first and second frequency converting means is provided independently so that local oscillation frequencies are frequencies produced by providing positive and negative frequency offsets of ½ of a channel spacing frequency with respect to the desired carrier frequency.

[0046] Furthermore, in accordance with this invention, in a receiving circuit, the two received signals are supplied to the first and second analog-to-digital converters without being frequency-converted.

[0047] The object and features of the present invention will become more readily apparent from the following detailed description of the preferred embodiments taken in conjunction with the accompanying drawings in which:

[0048] FIG. 1 is a block digram showing an arrangement of a receiving circuit according to a first embodiment of this invention;

[0049] FIG. 2 is a block digram showing an arrangement of a receiving circuit according to a second embodiment of this invention;

[0050] FIG. 3 is a block digram showing an arrangement of a receiving circuit according to a third embodiment of this invention;

[0051] FIG. 4 is a block digram showing an arrangement of a receiving circuit according to a fourth embodiment of this invention;

[0052] FIG. 5 is a block digram showing an arrangement of a receiving circuit according to a fifth embodiment of this invention;

[0053] FIG. 6 is a block digram showing an arrangement of a receiving circuit according to a sixth embodiment of this invention;

[0054] FIG. 7 is a block digram showing an arrangement of a receiving circuit according to a seventh embodiment of this invention;

[0055] FIG. 8 is a block digram showing an arrangement of a receiving circuit according to a eighth embodiment of this invention;

[0056] FIG. 9 is a block digram showing an arrangement of a receiving circuit according to a ninth embodiment of this invention;

[0057] FIG. 10 is a block diagram showing an arrangement of a receiving circuit according to a tenth embodiment of this invention;

[0058] FIG. 11 is a block diagram showing an arrangement of a receiving circuit according to a eleventh embodiment of this invention;

[0059] FIG. 12 is a block diagram showing an arrangement of a receiving circuit according to a twelfth embodiment of this invention;

[0060] FIG. 13 is a block diagram showing an arrangement of a receiving circuit according to a thirteenth embodiment of this invention;

[0061] FIG. 14 is a block diagram showing an arrangement of a receiving circuit according to a fourteenth embodiment of this invention;

[0062] FIG. 15 is a illustration of a concept of a transformer in the fourteenth embodiment;

[0063] FIG. 16 is a block diagram showing an arrangement of a receiving circuit according to a fifteenth embodiment of this invention;

[0064] FIG. 17 is a block diagram showing a concrete arrangement of a receiving circuit according to a fifteenth embodiment of this invention;

[0065] FIG. 18 is a block diagram showing an arrangement of a receiving circuit according to a sixteenth embodiment of this invention;

[0066] FIG. 19 is a block diagram showing an arrangement of a receiving circuit according to a seventeenth embodiment of this invention;

[0067] FIG. 20 is a diagrammatic view useful for describing a local frequency setting method in embodiments of this invention;

[0068] FIG. 21 is a block diagram showing an arrangement of a receiving circuit according to a eighteenth embodiment of this invention;

[0069] FIG. 22 is a block diagram showing an arrangement of a receiving circuit according to a nineteenth embodiment of this invention;

[0070] FIG. 23 is a block diagram showing an arrangement of a receiving circuit according to a twentieth embodiment of this invention;

[0071] FIG. 24 is a block diagram showing an arrangement of a receiving circuit according to a twenty first embodiment of this invention;

[0072] FIG. 25 is a concept illustration of a transformer in a twenty second embodiment of this invention;

[0073] FIG. 26 is an illustration for explaining a state of an alias generated as a result of sampling by an A/D converter in the twenty second embodiment of this invention;

[0074] FIG. 27 is an illustration of a state of a receiving channel in a multi-channel communication system in this invention;

[0075] FIG. 28 is an illustration of an A/D converted output with a negative frequency region appearing in the twenty second embodiment of this invention;

[0076] FIG. 29 is an illustration of a method of dividing a signal component into quadrature components on the basis of a cosine function and a sin function by utilizing a π/2 phase difference in the twenty second embodiment of this invention;

[0077] FIG. 30 is an illustration fo one example of a quadrature sampling operation in the case of A/D conversion of two orthogonal signals in the twenty second embodiment of this invention;

[0078] FIG. 31 is an illustration available for describing quadrature sampling taking offset in consideration;

[0079] FIG. 32 is an illustration of an example of a quadrature sampling operation different from those of FIGS. 30 and 31 in the case of the A/D conversion of two orthogonal signals in the twenty second embodiment of this invention;

[0080] FIG. 33 is an illustration of sampling pulses obtainable in performing the quadrature sampling operation shown in FIG. 32 in the twenty second embodiment of this invention;

[0081] FIG. 34 is a block diagram showing an arrangement of a receiving circuit according to a twenty third embodiment of this invention;

[0082] FIG. 35 is an illustration of a quadrature sampling operation in the case of A/D conversion of two orthogonal signals in the twenty third embodiment of this invention;

[0083] FIG. 36 is a block diagram showing an arrangement of a receiving circuit according to a twenty fourth embodiment of this invention;

[0084] FIG. 37 is an illustration of a quadrature sampling operation in the case of A/D conversion of two orthogonal signals in the twenty fourth embodiment of this invention;

[0085] FIG. 38 is a block diagram showing an arrangement of a receiving circuit according to a twenty fifth embodiment of this invention;

[0086] FIG. 39 is a simplified block diagram for describing one operation of a sampling signal generating source portion in the twenty fifth embodiment of this invention;

[0087] FIG. 40 is a simplified block diagram for describing one operation of a sampling signal generating source portion in the twenty fifth embodiment of this invention;

[0088] FIG. 41 is a simplified block diagram for describing a further operation of a sampling signal generating source portion in the twenty fifth embodiment of this invention;

[0089] FIG. 42 is a schematic illustration of the frequency disposition for the Japanese standard digital type automobile telephone system used for description of a twenty sixth embodiment of this invention;

[0090] FIG. 42 is a schematic illustration of the channel disposition for the Japanese standard digital type automobile telephone system in FIG. 42;

[0091] FIG. 44 is a block diagram showing an arrangement of a receiving circuit according to a twenty sixth embodiment of this invention;

[0092] FIG. 45 is a block diagram showing an arrangement of a receiving circuit according to a twenty seventh embodiment of this invention;

[0093] FIG. 46 is a block diagram showing an arrangement of a receiving circuit according to a twenty eighth embodiment of this invention;

[0094] FIG. 47 is a block diagram showing an arrangement of a receiving circuit according to a twenty ninth embodiment of this invention;

[0095] FIG. 48 is a block diagram showing an arrangement of a receiving circuit according to a thirtieth embodiment of this invention.

[0096] Prior to explanation of preferred embodiments of the present invention, a description will be made hereinbelow of a logical ground supporting the present invention. First of all, the description starts with a binary PSK, i.e., BPSK, which has currently mostly been employed in a digital modulation ways. A BPSK signal S_Bon a base frequency, i.e., baseband is expressible as follows.

[0097] S_B=Acos(θk)

[0098] where A represents an amplitude, and θk designates a phase indicative of BPSK information, i.e., θk=0, π.

[0099] When this baseband signal is modulated with a carrier angular frequency, ωc, the resulting modulated signal S_Cis expressible as follows. $\begin{array}{cc}\begin{array}{c}{S}_C=S_B\left(\theta k\right)\times a\cos \omega \mathrm{ct}+S_B\left(\theta k-\frac{1}{2}\pi \right)\times a\cos \left(\omega \mathrm{ct}-\frac{1}{2}\pi \right)\\ A_C=a\times A,\theta k=0,\pi \\ =A_C\cos \left(\omega \mathrm{ct}+\theta k\right)\end{array}& \left(1\right)\end{array}$

[0100] When this modulated signal is received and frequency-converted with a local frequency ωc for a frequency conversion, the frequency-converted output S_Rcan be given as follows. $\begin{array}{cc}\begin{array}{c}{S}_R=S_C\times b\cos \omega \mathrm{ct}\\ =\frac{A_C·b}{2}[\cos \left\{2\omega \mathrm{ct}+\theta k\right\}+\cos (\theta k\}]\mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ =A_R\left[\cos \left\{2\omega \mathrm{ct}+\theta k\right\}+\cos \left\{\theta k\right\}\right]\end{array}& \left(2\right)\end{array}$

[0101] Further, this frequency-converted output S_Ris passed through a low-pass filter in order to remove (reject) a high-frequency component 2ωc, the resulting output S_RFbecomes as follows, thus allowing the demodulation of the binary PSK, i.e., BPSK.

[0102] S_RF=A_Rcos(θk) θk=0, π  (3)

[0103] However, since in the receiving frequency conversion the local oscillation frequency signal is set to ωc equal to the carrier frequency, this local oscillation frequency signal radiates from the receiver into the air and can interfere with other nearby receivers.

[0104] In this invention, the local oscillation frequency is set as follows to eliminate such a problem. FIG. 20 illustrates a method of setting the local oscillation frequency according to this invention, where character A indicates a band of a desired channel with its carrier frequency being ωc, character B denotes a band of an upside adjacent (upper neighboring) channel, whose carrier frequency is ωcu, and character C depicts a band of a downside adjacent (lower neighboring) channel which has a carrier frequency of ωCL. The spacing between the carrier frequencies of the respective channels is approximately four times that of the base frequency ωb in the BPSK.

[0105] The bands of the respective channels assume the carrier frequency ±2 ωb. Accordingly, the locations remote by an amount corresponding to the base frequency 2ωb from the respective carrier frequencies come to gorge positions between the channels. Even if interference waves with a line spectrum exist at these positions, all the channels are hardly subjected to interference. Taking notice of this point, as a main means to resolve the aforesaid problem, according to this invention the local oscillation frequency of the receiver is set to middle values with respect to the frequencies of the adjacent channels.

[0106] Secondly, a description will be taken hereinbelow in conjunction with equations in terms of another point of this invention which shows how to construct circuits (which will be mentioned later) so that the demodulation is possible as well as the prior art in the case of setting the local oscillation frequency of the receiver as described above. When a modulated signal is received and the local oscillation frequency for the frequency conversion in the receiver is set to ωc+ωo as mentioned before, the frequency-converted output S_Rbecomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_R=S_C\times b\cos \left(\omega c+\omega o\right)t\\ =\frac{A_C·b}{2}\cos \left\{\omega \mathrm{ct}+\theta k\right\}\cos \left(\omega c+\omega o\right)t\mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ =A_R\left[\cos \left\{\left(2\omega c+\omega o\right)t+\theta k\right\}+\cos \left(-\omega \mathrm{ot}+\theta k\right)\right]\end{array}& \left(4\right)\end{array}$

[0107] If this frequency-converted output S_Ris passed through a low-pass filter to eliminate the high-frequency component 2ωc, the resultant output S_RFis as follows, thus resulting in producing a binary PSK, i.e., BPSK signal, which is offset by only the frequency ωo.

[0108] S_RF=A_R{cos(−ωot+θk)} θk=0, π  (50)

[0109] Subsequently, the local oscillation frequency is set to ωc−ωo lower by ωo than the carrier frequency ωc of a desired channel. In this instance, the frequency-converted output S_Lbecomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_L=S_C\times b\cos \left(\omega c-\omega o\right)t\\ =\frac{A_C·b}{2}\cos \left\{\omega \mathrm{ct}+\theta k\right\}\cos \left(\omega c-\omega o\right)t\mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ =A_R\left[\cos \left\{2\omega c-\omega o+\theta k\right\}+\cos \left\{\omega \mathrm{ot}+\theta k\right\}\right]\end{array}& \left(6\right)\end{array}$

[0110] If this frequency-converted output S_Lis passed through a low-pass filer to remove the high-frequency component 2ωc, the resultant output S_LFbecomes as follow, thus producing a BPSK signal which is also equal in phase to the output S_LF.

[0111] S_LF=A_Rcos(ωot+θk) θk=0, π  (7)

[0112] Since the reception local oscillation frequency is also an equal distance from the adjacent channels, the components of the adjacent channels are also introduced into a signal being demodulated. When the carrier frequency is taken as ωch, the signal of the upside adjacent channel assumes ωch=ωc+ωo, and hence the aforesaid reception side frequency conversion is performed as follows. If the modulated signal is received and the local oscillation frequency for the frequency conversion in the receiver is set to ωc+ωo as mentioned before, the frequency-converted output S_Rhis given as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{Rh}}=S_{\mathrm{Ch}}\times b\cos \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Ch}}·b}{2}\left[\cos \left\{\omega \mathrm{ct}+2\omega \mathrm{ot}+\theta h\right\}\cos \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Ch}}·b}{2}=A_{\mathrm{Rh}}\\ =A_{\mathrm{Rh}}\left[\cos \left\{\left(2\omega c+3\omega o\right)t+\theta h\right\}+\cos \left\{\omega \mathrm{ot}+\theta h\right\}\right]\end{array}& \left(8\right)\end{array}$

[0113] If this frequency-converted output S_Rhis passed through a low-pass filter to eliminate the high-frequency component 2ωc, the resultant output S_RFhis given as follows, thus producing a BPSK signal present in the same band as the desired channel.

[0114] S_RFh=A_Rhcos(ωot+θh) θh=0, π  (9)

[0115] On the other hand, when the carrier frequency is expressed by ωc₁, the signal of the downside adjacent channel represents ωc₁=ωc−2ωo, and the aforesaid reception side frequency conversion is carried out as follows. If the local oscillation frequency of the receiver is set to ωc+ωo as noted before, the frequency-converted output S_R1becomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{R1}}=S_{\mathrm{Cl}}\times b\cos \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Cl}}·b}{2}\left[\cos \left\{\left(\omega c-2\omega o\right)t+\theta l\right\}\cos \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Cl}}·b}{2}=A_{\mathrm{R1}}\\ =A_{\mathrm{R1}}\left[\cos \left(2\omega \mathrm{ct}-\omega \mathrm{ot}+\theta l\right)+\cos \left(\text{-}3\omega \mathrm{ot}+\theta l\right)\right]\\ \theta h=0,\pi \end{array}& \left(10\right)\end{array}$

[0116] If this frequency-converted output S_R1is passed through a low-pass filer to eliminate the high-frequency component 2ωc, the resultant output S_RF1is given as follows, which generates a BPSK signal at a frequency separated by 3ωo from the desired channel.

[0117] S_RF1=A_R1[cos(3ωot−θ1)]  (11)

[0118] Furthermore, a verification will be made hereinbelow of a frequency conversion condition of an adjacent channel in the case that the local frequency is ωc−ωo. When the carrier frequency is expressed by ωch, the signal of the upside adjacent channel takes ωch=ωc+2ωo, and therefore, when the local oscillation frequency is ωc−ωo, the frequency conversion is made as follows. First, if the modulated signal is received and the local oscillation frequency for the frequency conversion in the receiver is set to ωc−ωo as noted before, the frequency-converted output S_Lhbecomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{Lh}}=S_{\mathrm{Ch}}\times b\cos \left(\omega c-\omega o\right)t\\ =\frac{A_{\mathrm{Ch}}·b}{2}\left[\cos \left\{\omega \mathrm{ct}+2\omega \mathrm{ot}+\theta h\right\}\cos \left(\omega c-\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Ch}}·b}{2}=A_{\mathrm{Rh}}\\ =A_{\mathrm{Rh}}\left[\cos \left\{\left(2\omega c+\omega o\right)t+\theta h\right\}+\cos \left\{3\omega \mathrm{ot}+\theta h\right\}\right]\end{array}& \left(12\right)\end{array}$

[0119] If the frequency-converted output S_Lhis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_LFhbecomes as follows.

[0120] S_LFh=A_Rh[cos(3ωot+θh)] θh=0, π  (13)

[0121] On the other hand, when the carrier frequency is expressed by ω_C1, the signal of the downside adjacent channel takes ω_C1=ωc−2ωo, and therefore, the aforesaid reception side frequency conversion is carried out as follows. First, if the local oscillation frequency of the receiver is set to ωc−ωo as noted before, the frequency-converted output S_L1is given as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{L1}}=S_{\mathrm{Cl}}\times b\cos \left(\omega c-\omega o\right)t\\ =\frac{A_{\mathrm{Cl}}·b}{2}\left[\cos \left\{\left(\omega c-2\omega o\right)t+\theta l\right\}\cos \left(\omega c-\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Cl}}·b}{2}=A_{\mathrm{R1}}\\ =A_{\mathrm{L1}}\left[\cos \left\{\left(2\omega c-3\omega o\right)t+\theta l\right)\right\}+\cos \left(-\omega \mathrm{ot}+\theta l\right)]\\ \theta h=0,\pi \end{array}& \left(14\right)\end{array}$

[0122] If this frequency-converted output S_L1is passed through a low-pass filter to eliminate the high-frequency component 2ωc, the resultant output SLF1 becomes as follows, which produces a BPSK signal at a frequency equal to that of the desired channel.

[0123] S_LF1=A_R1cos(ωot−θ1)   (15)

[0124] From the above description, in the case that the local frequency is shifted upward by ωo, the output takes the following three kinds. $\begin{array}{cc}\begin{array}{ccc}{S}_{\mathrm{RF}}=A_R\cos \left(-\omega \mathrm{ot}+\theta k\right)& \dots & \mathrm{desired}\mathrm{channel}\\ S_{\mathrm{RFh}}=A_{\mathrm{Rh}}\cos \left(\omega \mathrm{ot}+\theta h\right)& \dots & \mathrm{upside}\mathrm{channel}\\ S_{\mathrm{RF1}}=A_{\mathrm{R1}}\cos (3\omega \mathrm{ot}-\theta l)& \dots & \mathrm{downside}\mathrm{channel}\end{array}& \left(16\right)\end{array}$

[0125] Going the the other way, when the local frequency is shifted downward by ωo, the output takes the following three kinds. $\begin{array}{cc}\begin{array}{ccc}{S}_{\mathrm{LF}}=A_R\cos \left(\omega \mathrm{ot}+\theta k\right)& \dots & \mathrm{desired}\mathrm{channel}\\ S_{\mathrm{LFh}}=A_{\mathrm{Rh}}\cos \left(3\omega \mathrm{ot}+\theta h\right)& \dots & \mathrm{upside}\mathrm{channel}\\ S_{\mathrm{LF1}}=A_{\mathrm{R1}}\cos \left(\omega \mathrm{ot}-\theta l\right)& \dots & \mathrm{downside}\mathrm{channel}\end{array}& \left(17\right)\end{array}$

[0126] The common component of these two groups is present in only the desired channel. Accordingly, if both are supplied as two inputs to an adder, only the desired channel can be derived from its output. Although the output is frequency-offset by ωo, this frequency offset is removable with a simple frequency offset circuit.

[0127] This invention is on the basis of the above-described principle an can be embodied as being described in the following embodiments.

[0128] First Embodiment

[0129] FIG. 1 is an illustration of an arrangement of a receiving circuit according to a first embodiment of this invention. In FIG. 1, the receiving circuit is shown, as comprising an antenna 1 for accepting a signal being received, first and second frequency converting circuits 2, 3 for accepting the received signal, a local frequency signal generating circuit 4 for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channels and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit 2 while outputting the downside frequency as a conversion frequency input to the second frequency converting circuit 3, a common wave extracting circuit 5 for extracting a component present in common in (common to) the output of the first frequency converting circuit 2 and the output of the second frequency converting circuit 3, a frequency offset circuit 6 for removing a frequency offset remaining in the output of the common wave extracting circuit 5, an offset frequency generating circuit 7 for performing a minute frequency conversion to supply an offset amount to the frequency offset circuit 6, and a filter 8 for eliminating an unnecessary frequency component remaining in the output of the frequency offset circuit 6.

[0130] Secondly, a description will be made hereinbelow of an operation of the receiving circuit according to this first embodiment. According to the above-mentioned equations, the received signal obtained through the antenna 1 is supplied to both the first and second frequency converting circuits 2, 3, while the local frequency signal generating circuit 4 separately supplies two different frequencies, i.e., the upside and downside frequencies equivalent to the central value between the channels, to the first and second frequency converting circuits 2, 3, which produces two output signals in terms of each of the three signals: the desired channel signal, the upside channel signal and the downside channel signal. According to the equation developments, the signal component present in common in the first and second frequency converting circuits 2, 3 is only the desired channel signal. When it is fed to the common wave extracting circuit 5 for extracting a balanced component, the balanced component mainly comprising the desired wave is obtainable. Since in the output of the common wave extracting circuit 5 a frequency offset of ωc remains, a minute frequency conversion is conducted in the offset frequency generating circuit 7 and the offset amount is removed in the frequency offset circuit 6. In addition, the unnecessary frequency component generated in this process is removed in the filter 8, before the resultant signal is fed as a baseband signal to a baseband signal processing circuit (not shown).

[0131] Second Embodiment

[0132] FIG. 2 shows an arrangement of a receiving circuit according to a second embodiment of this invention. In FIG. 2, the receiving circuit is shown, as comprising an antenna 1 for accepting a signal being received, first and second frequency converting circuits 2, 3 for accepting the received signal, a local frequency signal generating circuit 4 for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channel and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit 2 while outputting the downside frequency as a conversion frequency input to the second frequency converting circuit 3, a first frequency offset circuit 6A for removing a frequency offset contained in the output of the first frequency converting circuit 2, a second frequency offset circuit 6B for removing a frequency offset contained in the output of the second frequency converting circuit 3, an offset frequency generating circuit 7A for performing a minute frequency conversion to supply an offset amount to the respective frequency offset circuits 6A and 6B, a common wave extracting circuit 5A for extracting a component present in common in both the outputs of the first and second frequency offset circuits 6A, 6B, and a filter 8A for removing an unnecessary frequency component left in the output of the common wave extracting circuit 5A.

[0133] Secondly, a description will be taken hereinbelow of an operation of the receiving circuit according to the second embodiment. In this embodiment the process for performing a frequency offset is interchanged with the common wave extracting process as compared with the aforementioned first embodiment. That is, the frequency offset process is carried out ahead, so that the desired channel signal directly becomes a baseband signal whereby more stable extraction is expectable.

[0134] A description will be made hereinbelow of the appropriateness of the preference of the frequency offset process. In the case of the frequency offset for the signal group in which the local frequency is shifted upward by ωo, the shifting is made for eliminating only ωo, thus resulting in the following three kinds of outputs. $\begin{array}{cc}{S}_{\mathrm{RF}}=A_R\cos \left(2\omega \mathrm{ot}-\theta k\right)\dots \mathrm{desired}\mathrm{channel}{S}_{\mathrm{RFh}}=A_{\mathrm{Rh}}\cos \left(\theta h\right)\dots \mathrm{upside}\mathrm{channel}{S}_{\mathrm{RFl}}=A_{\mathrm{R1}}\cos \left(4\omega \mathrm{ot}-\mathrm{\theta 1}\right)\dots \mathrm{downside}\mathrm{channel}& \left(18\right)\end{array}$

[0135] On the other hand, in the case of the frequency offset for the signal group in which the local frequency is shifted downward by ωo, the shifting is made for eliminating only ωo, thus resulting in the following three kinds of outputs. $\begin{array}{cc}{S}_{\mathrm{LF}}=A_R\cos \left(2\omega \mathrm{ot}-\theta k\right)\dots \mathrm{desired}\mathrm{channel}{S}_{\mathrm{LFh}}=A_{\mathrm{Rh}}\cos \left(4\omega \mathrm{ot}+\theta h\right)\dots \mathrm{upside}\mathrm{channel}{S}_{\mathrm{LFl}}=A_{\mathrm{R1}}\cos \left(\mathrm{\theta 1}\right)\dots \mathrm{downside}\mathrm{channel}& \left(19\right)\end{array}$

[0136] The common component of both the groups is also present in only the desired channel. Accordingly, if both are fed as two inputs to an adder, a BPSK signal of only the desired channel can be derived from its output signal.

[0137] Third Embodiment

[0138] FIG. 3 illustrates an arrangement of a receiving circuit according to a third embodiment of this invention. In FIG. 3, included therein are an antenna 1 for accepting a signal being received, first and second frequency converting circuits 2, 3 for accepting the received signal, a local frequency signal generating circuit 4 for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channels and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit 2 while outputting the downside frequency as a conversion frequency input to the second frequency converting circuit 3, a first bandpass filter 9A for waveform-shaping the output of the first frequency converting circuit 2, a first A/D converter 10A for converting the output of the first bandpass filter 9A into a digital signal, a second bandpass filter 9B for waveform-shaping the output of the second frequency converting circuit 3, a second A/D converter 10B for converting the output of the second bandpass filter 9B into a digital signal, a common wave extracting circuit 5B for extracting a component present in common in both the outputs of the first and second A/D converters 10A and 10B, a frequency offset circuit 6C for removing a frequency offset remaining in the output of the common wave extracting circuit 5B, an offset frequency generating circuit 7B for performing a minute frequency conversion to supply the offset amount to the frequency offset circuit 6C, and a filter 8B for removing an unnecessary frequency component left in the output of the frequency offset circuit 6C.

[0139] Secondly, a description will be made hereinbelow of an operation of the receiving circuit according to the third embodiment of this invention. According to this embodiment, the outputs of the same two frequency converting circuits 2, 3 as in the aforementioned first embodiment are quantized with the A/D converters 10A, 10B, respectively, thereby providing processes equivalent to that of the first embodiment according to a digital calculation, that is, carrying out the common wave extraction, frequency offset, and filtering. The common wave extraction and filtering are possible with a digital filter technique, whereas the frequency output is possible with a digital quadrature modulation.

[0140] Furthermore, the principle of this embodiment will be described hereinbelow in terms of a quadrature PSK, i.e., QPSK or quadruplet QAM. The base frequency, i.e., a QPSK signal S_Bin the baseband, is expressible as follows.

[0141] S_B=Acos(θk)+jAsin(θk)   (20)

[0142] where j denotes an imaginary unit indicative of an imaginary axis perpendicular to a real axis, A depicts an amplitude, and θk designates a phase representative of QPSK information:

[0143] θk=(±¼, ¾)π

[0144] When this baseband signal is modulated with the carrier angular frequency ωc, the resultant modulated output Sc can be expressed as follows. $\begin{array}{cc}\begin{array}{c}{S}_C=S_B\times a\left(\cos \omega \mathrm{ct}+j\sin \omega \mathrm{ct}\right)\\ =A_C\left\{\cos \left(\omega \mathrm{ct}+\theta k\right)+j\sin \left(\omega \mathrm{ct}+\theta k\right)\right\}\\ A_C=a\times \sqrt{2}A,\theta k=\left(\pm \frac{1}{4},\pm \frac{3}{4}\right)\pi \\ =\frac{A_C}{2}\left[\cos \left\{\omega \mathrm{ct}+\theta k\right\}\right]+\frac{A_C}{2}\left[j\sin \left(\omega \mathrm{ct}+\theta k\right)\right]\end{array}& \left(21\right)\end{array}$

[0145] Generally, a real axis component is called an I-axis signal and an imaginary axis component is called a Q-axis signal. If this modulated signal is received and quadrature-demodulated with the frequency conversion local frequency ωc, the quadrature-demodulated I-axis output S_IRis given as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{IR}}=S_C\times b\cos \omega \mathrm{ct}\\ =\frac{A_C·b}{2\times 2}\left[\cos \left\{\omega \mathrm{ct}+\theta k\right\}\cos \omega \mathrm{ct}\right]\\ \mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ =A_R\left[\cos \left\{2\omega \mathrm{ct}+\theta k\right\}+\cos \left\{\theta k\right\}\right]\end{array}& \left(22\right)\end{array}$

[0146] If this quadrature-demodulated I-axis output S_IRis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_IRFbecomes as follows, thus allowing the demodulation of the I-axis signal of the quadrature PSK, i.e., QPSK.

[0147] S_IRF=A_Rcos(θk) θk=(±¼, ±¾)π  (23)

[0148] However, as well as the above description, in this instance, since the local oscillation frequency on the quadrature demodulation is equal to the carrier frequency ωc, this local oscillation frequency signal radiates from the receiver into the air, which can interfere with other nearby receivers. For this reason, when the local oscillation frequency of the receiver is set to ωc+ωo as noted before, the I-axis output S_IRon the quadrature demodulation becomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{IR}}=S_C\times b\cos \left(\omega c+\omega o\right)t\\ =\frac{A_C·b}{2}\left[\cos \left\{\omega \mathrm{ct}+\theta k\right\}\cos \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ =A_R\left[\cos \left\{\left(2\omega c+\omega o\right)t+\theta k\right\}+\cos \left(\omega \mathrm{ot}+\theta k\right)\right]\end{array}& \left(24\right)\end{array}$

[0149] If this quadrature-demodulated I-axis output S_IRis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_IRFbecomes as follows, thus providing an I-axis output of the quadrature PSK, i.e., QPSK.

[0150] S_IRF=A_R[cos(ωot+θk)]  (25)

[0151] Subsequently, the local oscillation frequency is set to ωc−ωo lower by ωo than the carrier frequency of the desired channel. In this case, the frequency-converted output S_ILbecomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{IL}}=S_C\times b\cos \left(\omega c-\omega o\right)t\\ =\frac{A_C·b}{2}\left[\cos \left\{\omega \mathrm{ct}+\theta k\right\}\cos \left(\omega c-\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ =A_R[\cos \left\{2\omega c+\omega \mathrm{ot}+\theta k\right\}+\cos \{-\omega \mathrm{ot}+\theta k)]\end{array}& \left(26\right)\end{array}$

[0152] If this quadrature-demodulated I-axis output S_ILis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_ILFis given as follows.

[0153] S_ILF=A_Rcos{−ωot+θk} θk=(±¼, ±¾)π  (27)

[0154] If the quadrature-demodulated I-axis output S_IRhis passed through a low-pass filter to remove the high-frequency component 2ωc, the output S_IRFhis obtainable.

[0155] The signal, demodulated, of adjacent channels being an equal distance from the reception local oscillation frequency is as follows. That is, when the carrier frequency is taken as ωch, the upside adjacent channel signal takes ωch=ωc+2ωo, and hence, if the local oscillation frequency is ωc+ωo, the quadrature-demodulated I-axis output S_IRhbecomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{IRh}}=S_{\mathrm{Ch}}\times b\cos \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Ch}}·b}{2}\left[\cos \left\{\omega \mathrm{ct}+2\omega \mathrm{ot}+\theta h\right\}\cos \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Ch}}·b}{2}=A_{\mathrm{Rh}}\\ S_{\mathrm{IRh}}=A_{\mathrm{Rh}}\left[\cos \left\{\left(2\omega c+3\omega o\right)t+\theta h\right\}+\cos \left\{\omega \mathrm{ot}+\theta h\right\}\right]\end{array}& \left(28\right)\end{array}$

[0156] If this quadrature-demodulated I-axis output S_IRhis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_IRFhbecomes as follows, thus producing a quadrature PSK, i.e., QPSK signal, present in the same band as that of the desired channel.

[0157] S_IRFh=A_Rh[cos(ωot+θh)] θh=(±¼, ±¾)π  (29)

[0158] Going the other way, when the carrier frequency is given by ω_c1, the downside adjacent channel signal represents ω_c1=ωc−2ωo. Accordingly, when the local oscillation frequency is set to ωc+ωo as noted before, the reception side quadrature-demodulated I-axis output S_IR1becomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{IR1}}=S_{\mathrm{Cl}}\times b\cos \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Cl}}·b}{2}\left[\cos \left\{\left(\omega c-2\omega o\right)t+\theta l\right\}\cos \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Cl}}·b}{2}=A_{\mathrm{R1}}\\ =A_{\mathrm{R1}}\left[\cos \left(2\omega \mathrm{ct}-\omega \mathrm{ot}+\theta l\right)+\cos \left(-\omega \mathrm{ot}+\theta l\right)\right]\end{array}& \left(30\right)\end{array}$

[0159] If this quadrature-demodulated I-axis output S_IR1is passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_IRF1is given as follows, this producing a quadrature PSK, i.e., QPSK signal, at the same frequency as that of the desired channel.

[0160] S_IRF1=A_R1[cos(−ωot+θ1)] θ1=(±{fraction (1/4)}, ±{fraction (3/4)})π  (31)

[0161] Subsequently, when the local oscillation frequency of the receiver is set to ωc+ωo as noted before and its phase is delayed by π/2, the quadrature-demodulated Q-axis output S_QRis obtainable as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{QR}}=S_C\times b\sin \left(\omega c+\omega o\right)t\\ =\frac{A_C·b}{2}\left[\cos \left(\omega \mathrm{ct}+\theta k\right)\sin \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ =A_R\left[\sin \left\{\left(2\omega c+\omega o\right)t+\theta k\right\}+\sin \left(-\omega \mathrm{ot}+\theta k\right)\right]\end{array}& \left(32\right)\end{array}$

[0162] If this quadrature-demodulated Q-axis output S_QRis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_QRFis given as follows, whereby the Q-axis output of the quadrature PSK, i.e., QPSK, is obtainable.

[0163] S_QRF=A_Rsin(−ωot−θk)   (33)

[0164] Let it be assumed that, in the case that the local oscillation frequency is set to ωc−ωo lower by ωo than the carrier frequency of the desired channel, the phase is delayed by π/2. In this instance, the quadrature-demodulated Q-axis output S_QLbecomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{QL}}=S_C\times b\sin \left(\omega c-\omega o\right)t\\ =\frac{A_C·b}{2}\left[\cos \left\{\omega \mathrm{ct}+\theta k\right\}\sin \left(\omega c-\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_C·b}{2}=A_R\\ S_{\mathrm{QL}}=A_{\mathrm{RL}}\left[\sin \left\{\left(2\omega c+\omega o\right)t+\theta k\right\}+\sin \left(\omega \mathrm{ot}+\theta k\right)\right]\end{array}& \left(34\right)\end{array}$

[0165] If this quadrature-demodulated Q-axis output S_QLis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_QLFis given as follows, whereby the quadrature PSK signal Q-axis output different in polarity from S_QLFis obtainable.

[0166] S_QLF=A_Rsin(ωot+θk) θk=(±¼, ±¾)π  (35)

[0167] Furthermore, the analysis of the adjacent channels will be made hereinbelow in terms of the quadrature-demodulated A-axis output. When the carrier frequency is ω_ch, the upside adjacent channel signal represents ω_ch=ωc+2ωo, and therefore, in the case that the local oscillation frequency is set to ωc+ωo, the quadrature-demodulated Q-axis output S_QRhof the upside channel becomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{QRh}}=S_{\mathrm{Ch}}\times b\sin \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Ch}}·b}{2}\left[\cos \left\{\omega \mathrm{ct}+2\omega \mathrm{ot}+\theta h\right\}\sin \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Ch}}·b}{2}=A_{\mathrm{Rh}}\\ =A_{\mathrm{Rh}}\left[\sin \left\{\left(2\omega c+3\omega o\right)t+\theta h\right\}+\sin \left\{\omega \mathrm{ot}+\theta h\right\}\right]\end{array}& \left(36\right)\end{array}$

[0168] If this quadrature-demodulated Q-axis output S_QRhis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_QLFhbecomes as follows, which produces a quadrature PSK, i.e., QPSK signal Q-axis output present in the same band as that of the desired channel.

[0169] S_QRFh=A_Rh[sin(ωot+θh)] θh=(±¼, ±¾)π  (37)

[0170] A similar process is effected for the downside adjacent channel. Since the carrier frequency ω_c1is ω_c1=ωc−2ωo, the quadrature-demodulated Q-axis output S_QR1at the local oscillation frequency ωc+ωo becomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{QRl}}=S_{\mathrm{Cl}}\times b\sin \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Cl}}·b}{2}\left[\cos \left\{\left(\omega c-2\omega o\right)t+\theta l\right\}\sin \left(\omega c+\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Cl}}·b}{2}=A_{\mathrm{Rl}}\\ =A_{\mathrm{Rl}}\left[\sin \left(2\omega \mathrm{ct}-\omega \mathrm{ot}+\theta l\right)+\sin \left(-3\omega \mathrm{ot}+\theta l\right)\right]\end{array}& \left(38\right)\end{array}$

[0171] If this quadrature-demodulated Q-axis output S_QR1is passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_QRF1is given as follows, thus producing a quadrature PSK, i.e., QPSK, signal Q-axis output at a frequency separated by 3ωo from the desired channel.

[0172] S_QRF1=A_R1sin(−3ωot+θ1) θ1=(±¼, ±¾)π  (39)

[0173] On the other hand, if the local oscillation frequency is set to ωc−ωo, the quadrature-demodulated Q-axis output S_QRof the adjacent channel becomes as follows. That is, since the carrier frequency ω_chfor the upside adjacent channel signal is ω_ch=ωc+2ωo, the quadrature-demodulated Q-axis output S_QRhat the local oscillation frequency ωc−ωo becomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{QRh}}=S_{\mathrm{Ch}}\times b\sin \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Ch}}·b}{2}\left[\cos \left\{\omega \mathrm{ct}+2\omega \mathrm{ot}+\theta h\right\}\sin \left(\omega c-\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Ch}}·b}{2}=A_{\mathrm{Rh}}\\ S_{\mathrm{QRh}}=A_{\mathrm{Rh}}\left[\sin \left\{\left(2\omega c+\omega o\right)t+\theta h\right\}+\sin \left(3\omega \mathrm{ot}+\theta h\right)\right]\end{array}& \left(40\right)\end{array}$

[0174] If the quadrature-demodulated Q-axis output S_QRhis passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_QRFhis given as follows, thus producing a quadrature PSK, i.e., QPSK, signal Q-axis output present in a band separated by 3ωo from the desired channel.

[0175] S_QRFh=A_QRh[sin(3ωot+θh)] θh=(±¼, ±¾)π  (41)

[0176] A similar process is implemented in terms of the downside adjacent channel signal. The carrier frequency ω_c1is ω_c1=ωc−2ωo, and the local oscillation frequency of the receiver is ωc−ωo. Thus, the quadrature-demodulated Q-axis output S_QR1becomes as follows. $\begin{array}{cc}\begin{array}{c}{S}_{\mathrm{QRl}}=S_{\mathrm{Cl}}\times b\sin \left(\omega c+\omega o\right)t\\ =\frac{A_{\mathrm{Cl}}·b}{2}\left[\cos \left\{\left(\omega c-2\omega o\right)t+\theta l\right\}\sin \left(\omega c-\omega o\right)t\right]\\ \mathrm{if}\mathrm{using}\frac{A_{\mathrm{Cl}}·b}{2}=A_{\mathrm{Rl}}\\ =A_{\mathrm{Rl}}\left[\sin \left\{\left(2\omega c-3\omega o\right)t+\theta l\right\}+\sin \left(-\omega \mathrm{ot}+\theta l\right)\right]\end{array}& \left(42\right)\end{array}$

[0177] If this quadrature-demodulated Q-axis output S_QR1is passed through a low-pass filter to remove the high-frequency component 2ωc, the resultant output S_QRF1is given as follows, thus producing a quadrature PSK, i.e., QPSK, signal Q-axis output at the same frequency as that of the desired channel.

[0178] S_QRF1=A_R1[sin(−ωot+θ1)] θ1=(±¼, ±¾)π  (43)

[0179] In summary, the above description becomes as follows.

[0180] I-Axis Quadrature-Demodulated Output Output at Upside Local Frequency $\begin{array}{cc}\mathrm{Output}\mathrm{at}\mathrm{Upside}\mathrm{Local}\mathrm{Frequency}{S}_{\mathrm{IRF}}=A_R\left[\cos \left(-\omega \mathrm{ot}+\theta k\right)\right]\dots \mathrm{desired}\mathrm{channel}{S}_{\mathrm{IRFh}}=A_{\mathrm{Rh}}\left[\cos \left(\omega \mathrm{ot}+\theta h\right)\right]\dots \mathrm{upside}\mathrm{channel}{S}_{\mathrm{IRFl}}=A_{\mathrm{Rl}}\left[\cos \left(-3\omega \mathrm{ot}+\theta l\right)\right]\dots \mathrm{downside}\mathrm{channel}\mathrm{Output}\mathrm{at}\mathrm{Downside}\mathrm{Local}\mathrm{Frequency}{S}_{\mathrm{ILF}}=A_R\left[\cos \left(\omega \mathrm{ot}+\theta k\right)\right]\dots \mathrm{desired}\mathrm{channel}{S}_{\mathrm{ILFh}}=A_{\mathrm{Rh}}\left[\cos \left(3\omega \mathrm{ot}+\theta h\right)\right]\dots \mathrm{upside}\mathrm{channel}{S}_{\mathrm{ILFl}}=A_{\mathrm{R1}}\left[\cos \left(-\omega \mathrm{ot}+\theta l\right)\right]\dots \mathrm{downside}\mathrm{channel}& \left(44\right)\\ Q\text{-}\mathrm{Axis}\mathrm{Quadrature}\text{-}\mathrm{Demodulated}\mathrm{Output}\mathrm{Output}\mathrm{at}\mathrm{Upside}\mathrm{Local}\mathrm{Frequency}{S}_{\mathrm{QRF}}=A_R\left[\sin \left(-\omega \mathrm{ot}+\theta k\right)\right]\dots \mathrm{desired}\mathrm{channel}{S}_{\mathrm{QRFh}}=A_{\mathrm{Rh}}\left[\sin \left(\omega \mathrm{ot}+\theta h\right)\right]\dots \mathrm{upside}\mathrm{channel}{S}_{\mathrm{QLFh}}=A_{\mathrm{Rh}}\left[\sin \left(-3\omega \mathrm{ot}+\theta h\right)\right]\dots \mathrm{downside}\mathrm{channel}\mathrm{Output}\mathrm{at}\mathrm{Downside}\mathrm{Local}\mathrm{Frequency}{S}_{\mathrm{QLF}}=A_R\left[\sin \left(\omega \mathrm{ot}+\theta k\right)\right]\dots \mathrm{desired}\mathrm{channel}{S}_{\mathrm{QRFl}}=A_{\mathrm{Rl}}\left[\sin \left(3\omega \mathrm{ot}+\theta l\right)\right]\dots \mathrm{upside}\mathrm{channel}{S}_{\mathrm{QLFl}}=A_{\mathrm{Rl}}\left[\sin \left(-\omega \mathrm{ot}+\theta l\right)\right]\dots \mathrm{downside}\mathrm{channel}& \left(45\right)\end{array}$

[0181] It is found from the above equations that, as described before, in the I-axis side the desired channel is contained in common in the outputs of the two quadrature-demodulating circuits, while in the Q-axis side the desired channel is contained in common and in antiphase in the outputs of the two quadrature-demodulating circuits. The third embodiment of this invention is realized on the basis of this principle.

[0182] Fourth Embodiment

[0183] FIG. 4 shows an arrangement of a receiving circuit according to a fourth embodiment of this invention. In FIG. 4, included therein are an antenna 1 for accepting a signal being received, first and second frequency converting circuit 2, 3 for accepting the received signal, a local frequency signal generating circuit 4 for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channels and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first frequency converting circuit 2 while outputting the downside frequency as a conversion frequency input to the second frequency converting circuit 3, a first bandpass filter 9A for waveform-shaping the output of the first frequency converting circuit 2, a first A/D converter 10A for converting the output of the first bandpass filter 9A into a digital signal, a second bandpass filter 9B for waveform-shaping the output of the second frequency converting circuit 3, a second A/D converter 10B for converting the output of the second bandpass filter 9B into a digital signal, a first frequency offset circuit 6D for removing a frequency offset contained in the output of the first A/D converter 10A, a second frequency offset circuit 6E for removing a frequency offset contained in the output of the second A/D converter 10B, the offset frequency generating circuit 7C for performing a minute frequency conversion to supply the offset amount to the respective frequency offset circuits 6D and 6E, a common wave extracting circuit 5C for extracting a component present in common in both the outputs of the first and second frequency offset circuits 6D and 6E, and a filter 8C for removing an unnecessary frequency component left in the output of the common wave extracting circuit 5C.

[0184] Secondly, a description will be made hereinbelow of an operation of the receiving circuit according to the fourth embodiment of this invention. In this embodiment, the process of performing a frequency offset in interchanged with the common wave extracting process as compared with the aforementioned third embodiment. The preceding execution of the frequency offset process allows the desired channel signal to directly constitute the base signal and permits a more stable extraction to be executable. In addition, digitalization allows the quadrature-demodulating function with a high accuracy, provides the suitability for integration, and causes the reduction of the power consumption.

[0185] Fifth Embodiment

[0186] FIG. 5 is an illustration of an arrangement of a receiving circuit according to a fifth embodiment of this invention. In FIG. 5, there are shown an antenna 1 for accepting a signal being received, first and second quadrature demodulating circuits 11 and 12 for receiving the received signal, a local frequency signal generating circuit 4 for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channels and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating circuit 11 while outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit 12, a first common wave extracting circuit 5D for extracting a component present in common in both the I outputs of the first and second quadrature demodulating circuits 11 and 12, a second common wave extracting circuit 5E for extracting a component present in common in both the Q output of the first quadrature demodulating circuit 11 and the Q output of the second quadrature demodulating circuit 12 reversed in polarity, a first frequency offset circuit 6F for removing a frequency offset remaining in the I side output extracted through the first common wave extracting circuit 5D, a second frequency offset circuit 6G for removing a frequency offset remaining in the Q side output extracted through the second common wave extracting circuit 5E, an offset frequency generating circuit 7D for performing a minute frequency conversion to supply the offset amount to the respective frequency offset circuits 6E and 6G, a first filter 8D for removing an unnecessary frequency component remaining in the output of the first frequency offset circuit 6F, and a second filter 8E for removing an unnecessary frequency component remaining in the output of the second frequency offset circuit 6G.

[0187] Secondly, a description will be made hereinbelow of an operation of the receiving circuit according to the fifth embodiment. In this embodiment, the concept of this invention is applied to a quaternary PSK, i.e., QPSK, of the digital modulation. According to the above description referring to the aforementioned equations, the received signal obtained from the antenna 1 is supplied to the first and second quadrature demodulating circuits 11 and 12, while the local frequency signal generating circuit 4A separately supplies two different frequencies, that is, the upside and downside frequencies comparable with the central value between the channels, to the first and second quadrature demodulating circuits 11 and 12, which produces four output signals in terms of each of three signals of the desired channel, the upside channel and the downside channel. According to the equation developments, the signal component present in common in both the first and second quadrature demodulating circuits 11 and 12 is only the desired channel signal, and the I-axis side is extractable as a balanced component, whereas the Q-axis side is extractable as a differential component. Accordingly, if the I-axis side balanced component is fed to the common wave extracting circuit 5D and the Q-axis side differential component is fed to the common wave extracting component 5E, the I-axis signal and Q-axis signal of the desired channel are obtainable. Since a frequency offset of ωo remains in the outputs of the common wave extracting circuits 5D and 5E, a minute frequency conversion is made in the offset frequency generating circuit 7D so that the offset amount is removed in the frequency offset circuits 6F and 6G. In addition, the unnecessary frequency component generated in this process is removed in the filters 8D and 8E, before the signal is supplied as a baseband signal to the baseband signal processing section.

[0188] Sixth Embodiment

[0189] FIG. 6 is an illustration of an arrangement of a receiving circuit according to a sixth embodiment of this invention. In FIG. 6, there are shown an antenna 1 for accepting a signal being received, first and second quadrature demodulating circuits 11 and 12 for receiving the received signal, a local frequency signal generating circuit 4A for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channels and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating circuit 11 while outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit 12, first and second frequency offset circuits 6H and 6I for removing a frequency offset contained in common in the I output and Q output of the respective quadrature demodulating circuits 11 and 12, an offset frequency generating circuit 7E for carrying out a minute frequency conversion to supply the offset amount to the respective frequency offset circuits 6H and 6I, a first common wave extracting circuit 5F for extracting a component present in common in both the I outputs of the first and second frequency offset circuits 6H and 6I, a second common wave extracting circuit 5G for extracting a component present in common in both the Q output of the first frequency offset circuit 6H and the Q output of the second frequency offset circuit 6I reversed in polarity, and first and second filters 8F and 8G for removing unnecessary frequency components remaining in the outputs of the respective common wave extracting circuits 5F and 5G.

[0190] Secondly, a description will be made hereinbelow of an operation of the receiving circuit of the sixth embodiment. In this embodiment, the common wave extracting process is interchanged with the frequency offset process as compared with the aforementioned fifth embodiment. When the frequency offset process is executed precedently, the desired channel signal directly becomes a base signal, which allows more stable extraction to be expectable.

[0191] Seventh Embodiment

[0192] FIG. 7 shows an arrangement of a receiving circuit according to a seventh embodiment of this invention. In FIG. 7, there are shown an antenna 1 for accepting a signal being received, first and second quadrature demodulating circuits 11 and 12 for receiving the received signal, a local frequency signal generating circuit 4 for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channels and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating circuit 11 while outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit 12, first and second bandpass filters 9C and 9D for waveform-shaping the I output and Q output of the first quadrature demodulating circuit 11, respectively, first and second A/D converters 10C and 10D for converting the outputs of the first and second bandpass filter 9C and 9D into digital signals, respectively, third and fourth band pass filters 9E and 9F for waveform-shaping the I output and Q output of the second quadrature demodulating circuit 12, respectively, third and fourth A/D converters 10E and 10F for converting the outputs of the third and fourth bandpass filters 9E and 9F into digital signals, respectively, a first common wave extracting circuit 5H for extracting a component present in common in the I outputs of the first and third A/D converters 10C and 10E, a second common wave extracting circuit 5I for extracting a component present in common in the Q output of the second A/D converter 10D and the Q output of the fourth A/D converter 10E reversed in polarity, a first frequency offset circuit 6J for removing a frequency offset remaining in the I side output extracted through the first common wave extracting circuit 5H, a second frequency offset circuit 6K for removing a frequency offset remaining in the Q side output extracted through the second common wave extracting circuit 5I, an offset frequency generating circuit 7F for carrying out a minute frequency conversion to supply the offset amount to the respective frequency offset circuits 6J and 6K, a first filter 8H for removing an unnecessary frequency component left in the output of the first frequency offset circuit 6J, and a second filter 8I for removing an unnecessary frequency component left in the output of the second frequency offset circuit 6K.

[0193] Secondly, a description will be made hereinbelow of an operation of the receiving circuit according to this seventh embodiment. In this seventh embodiment, the outputs of the same two quadrature demodulating circuits 11, 12 as in the aforementioned fifth embodiment are quantized with the A/D converters 10C to 10F so as to accomplish an operation equivalent to that of the fifth embodiment by means of the digital calculation, that is, to perform the common wave extraction, the frequency offset and the filtering. The common wave extraction and the filtering are possible by the digital filter technique, while the frequency offset is possible by the digital quadrature modulation.

[0194] Eighth Embodiment

[0195] FIG. 8 shows an arrangement of a receiving circuit according to an eighth embodiment of this invention. In FIG. 8, there are shown an antenna 1 for accepting a signal being received, first and second quadrature demodulating circuits 11 and 12 for receiving the received signal, a local frequency signal generating circuit 4 for generating middle frequencies between the radio carrier frequency of the received signal and the radio carrier frequencies of the adjacent upside and downside channels and for outputting the upside frequency of the upside and downside two wave frequencies as a conversion frequency input to the first quadrature demodulating circuit 11 while outputting the downside frequency as a conversion frequency input to the second quadrature demodulating circuit 12, first and second bandpass filters 9C and 9D for waveform-shaping the I output and Q output of the first quadrature demodulating circuit 11, respectively, first and second A/D converters 10C and 10D for converting the outputs of the first and second bandpass filter 9C and 9D into digital signals, respectively, third and fourth band pass filters 9E and 9F for waveform-shaping the I output and Q output of the second quadrature demodulating circuit 12, respectively, third and fourth A/D converters 10E and 10F for converting the outputs of the third and fourth bandpass filters 9E and 9F into digital signals, respectively, a frequency offset circuit 6L for removing frequency offsets remaining in the I outputs and Q outputs of the respective A/C converters 10C to 10F, an offset frequency generating circuit 7G for performing a minute frequency conversion to supply the offset amount to the respective frequency offset circuit 6L, a first common wave extracting circuit 5J for extracting a component present in common in the I outputs of the frequency offset circuits 6L, a second common wave extracting circuit 5K for extracting a component present in common in the Q output and the polarity-reversed Q output of the frequency offset circuit 6L, a first filter 8J for removing an unnecessary frequency component left in the I output extracted through the first common wave extracting circuit 5J, and a second filter 8K for removing an unnecessary frequency component left in the Q output of the second common wave extracting circuit 5K.

[0196] Secondly, a description will be made hereinbelow of an operation of the receiving circuit according to the eighth embodiment. In this embodiment, the common wave extraction process is interchanged with the frequency offset process as compared with the aforementioned seventh embodiment. That is, the preceded frequency offset process permits the desired channel signal to directly become a base signal, which makes more stable extraction expectable. In addition, the digitalization allows the quadrature-demodulating function with a high accuracy, provides the suitability for integration, and caused the reduction of the power consumption.

[0197] Ninth Embodiment

[0198] FIG. 9 is an illustration of an arrangement of a receiving circuit according to ninth embodiment of this invention. In this embodiment, a correlator 13 for calculating a cross-correlation is used in place of the common wave extracting circuit 5B in the third embodiment shown in FIG. 3. According to this embodiment, since the common wave extraction is effected by the digital filter technique, although the use of the correlator 13 causes the inversion of the polarity of the correlation coefficient if the components contained in common are different in polarity from each other, the correlator 13 can ensure its amplitude as it is. This feature of this embodiment is also applicable to the above-described fourth, seventh and eighth embodiments.

[0199] Tenth Embodiment

[0200] FIG. 10 is an illustration of an arrangement of a receiving circuit according to a tenth embodiment of this invention. In this embodiment, only the first frequency converting circuit 2 is provided for accepting the received signal from the antenna 1, as compared with the first embodiment shown in FIG. 1. After the frequency conversion by this first frequency converting circuit 2, a second frequency converting circuit 15, accepting a frequency corresponding to the inter-channel frequency 2ωo from the local frequency signal generating circuit 4B, provides a frequency-converted output in the side where the frequency conversion is not made by the first frequency converting circuit 2, thus ensuring two frequency-converted outputs necessary for the desired channel extraction by a common wave extracting circuit 5L.

[0201] Accordingly, in this embodiment, the outputs of the first and second frequency converting circuits 2 and 15 agree with the two outputs of the first and second frequency converting circuits 2 and 3 in the first embodiment, and hence the treatment of the carrier frequency is possible with only one high-frequency circuit set, with the result that not only the space necessary for the circuit but also the power consumption are reducible. The feature of this embodiment is also applicable to the second embodiment.

[0202] Eleventh Embodiment

[0203] FIG. 11 is an illustration of an arrangement of a receiving circuit according to an eleventh embodiment of this invention. In this embodiment, as compared with the third embodiment shown in FIG. 3, the frequency converting circuit for accepting the received signal is effected with only the first frequency converting circuit 2 and further the quantizing means is constructed with only a bandpass filter 9C and an A/D converter 10C responsive to the output of the first frequency converting circuit 2. After the A/D conversion by the A/D converter 10C, a digital frequency converting circuit 16, accepting a frequency corresponding to the inter-channel frequency 2ωo from a digital frequency generating circuit 17, carries a digital frequency conversion, thus providing a frequency-converted output of the side where the frequency conversion is not made by the first frequency converting circuit 2 so as to obtain two frequency-converted digital outputs necessary for the extraction of the desired channel by a common wave extracting circuit 5M.

[0204] Thus, according to this embodiment, the output of the first frequency converting circuit 2 A/D-converted through the A/D converter 10C and the output of the digital frequency converting circuit 16 with the two outputs of the first and second frequency converting circuits 2 and 3 in the above-mentioned third embodiment, with the result that the treatment of the carrier frequency is possible with only on a high-frequency circuit set, which makes it possible to reduce the space necessary for the circuit as well as the power consumption. The feature of this embodiment is also applicable to the fourth embodiment.

[0205] Twelfth Embodiment

[0206] FIG. 12 is an illustration of an arrangement of a receiving circuit according to a twelfth embodiment of this invention. A feature of this embodiment is that as compared with the fifth embodiment shown in FIG. 5 the quadrature demodulating means for accepting the received signal is constructed with only one quadrature demodulating circuit 11 and the two outputs of the quadrature demodulating circuit 11 are frequency-converted in a frequency converting circuit 15A which accepts a frequency corresponding to the inter-channel frequency 2ωo from a local frequency signal generating circuit 4B, whereby the frequency-converted output of the side which is not quadrature-demodulated is obtainable to obtain two quadrature-demodulated outputs necessary for the extraction of the desired channel by first and second common wave extracting circuits 5N and 5P.

[0207] Thus according to this embodiment, the outputs of the quadrature demodulating circuit 11 and the frequency converting circuit 15A coincide with the two quadrature-demodulated outputs of the first and second quadrature demodulating circuits 11 and 12 in the fifth embodiment, with the result that the treatment of the carrier frequency is possible with only on a high-frequency circuit set, which makes it possible to reduce the space necessary for the circuit as well as the power consumption. The feature of this embodiment is also applicable to the sixth embodiment.

[0208] Thirteenth Embodiment

[0209] FIG. 13 is an illustration of an arrangement of a receiving circuit according to a thirteenth embodiment of the invention. A feature of this embodiment is that as compared with the seventh embodiment shown in FIG. 7 the quadrature demodulating means for accepting the received signal is constructed with only one quadrature demodulating circuit 11 and the quantizing means is constructed with one bandpass filters 9D, 9E and one A/D converters 10D, 10E. After the A/D conversion by the A/D converters 10D and 10E, a digital conversion is made in a digital frequency conversing circuit 16A which accepts a frequency corresponding to the inter-channel frequency 2ωo from a digital frequency generating circuit 17A, whereby a frequency-converted output of the side where the quadrature modulation is not effected in the quadrature demodulating circuit 11 is obtainable to ensure two quadrature-demodulated outputs necessary for the extension of the desired channel by first and second common wave extracting circuits 5Q and 5R.

[0210] Thus, according to this embodiment, the outputs of the quadrature demodulating circuit 11 A/D converted by the A/D converters 10D and 10E and the output of the digital frequency converting circuit 16A agree with the two outputs of the first and second quadrature of demodulating circuits 11 and 12 in the seventh embodiment, with the result that the treatment of the carrier frequency is possible with only on a high-frequency circuit set, which makes it possible to reduce the space necessary for the circuit as well as the power consumption. The feature of this embodiment is also applicable to the eighth embodiment.

[0211] Fourteenth Embodiment

[0212] FIG. 14 is an illustration of an arrangement of a receiving circuit according to a fourteenth embodiment of this invention. This embodiment relates to an improvement of the common wave extracting circuit as employed in the above-described first embodiment. In FIG. 14, as well as the first embodiment, the receiving circuit includes an antenna 1, a first frequency converting circuit 2, a second frequency converting circuit 3, a local frequency signal generating circuit 4, a common wave extracting circuit 5, and a filter 8. Further included therein are a received signal input section 20, an input line 21 through which the output of the first frequency converting circuit 2 passes, an input line 22 through which the output of the second frequency converting circuit 3 passes, integrating circuits 23 and 24 also serving as low-pass filters for accepting the outputs of the first and second frequency converting circuits 2 and 3 through the input lines 22 and 22, respectively, first and second buffer amplifiers 25 and 26 for accepting the outputs of the integrating circuits 23 and 24, respectively, and first and second transformers 27 and 28 having primary coils whose one ends receive the outputs of the first and second buffer amplifiers 25 and 26. The other ends of the primary coils of the first and second transformers 27, 28 are alternating-grounded, and their secondary coils are coupled in parallel to each other in a state with the same polarity. A junction 29 between one ends of the secondary coils equal in polarity to the primary coil is used for its output, while the other ends are grounded. Further, included therein a third buffer amplifier 30 whose input terminal is connected to the junction 29. The output 31 of the third buffer amplifier 30 is supplied as the output of the common wave extracting circuit 5 to a next-stage frequency offset circuit 6.

[0213] Secondly, a description will be made hereinbelow of an operation of the common wave extracting circuit 5 in the fourteenth embodiment. A common wave e_Dof a desired wave signal component and an adjacent channel wave signal component e_Uare obtainable from the first and second frequency converting circuits 2 and 3. As to the adjacent channel wave signal component, the component obtainable from the first frequency converting circuit 2 and the component obtainable from the second frequency converting circuit 3 are different in central frequency from each other, and hence the component obtainable from the first frequency converting circuit 2 will be expressed by e_U1, while the component obtainable from the second frequency converting circuit 3 will be expressed by e_U2. That is, the signal from the first frequency converting circuit 2 assumes e_D+e_U1, whereas the signal from the second frequency converting circuit 3 takes e_D+e_U2. There is a possibility that, in addition to those signals, unnecessary high-frequency components are contained in the outputs of the first and second frequency converting circuits 2 and 3. These high-frequency components are lessened by means of the integrating circuits 23 and 24 also serving as low-press filters.

[0214] The signal e_D+e_U1from the frequency converting circuit 2 and the signal e_D+e_U2from the second frequency converting circuit 3, which are respectively passed through the integrating circuits 23 and 24, are fed to the corresponding buffer amplifiers 25 and 26, respectively. Each of the buffer amplifiers 25 and 26 has an output impedance set to a low value. After being passed through the buffer amplifiers 25 and 26, the signal e_D+e_U1from the first frequency converting circuit 2 and the signal e_D+e_U2from the second frequency converting circuit 3 are fed to the primary coils of the transformers 27 and 28, respectively. The ratio of the number of winding turns of the secondary coil to the number of winding turns of the primary coil of each of the transformers 27, 28 is 1, with the result that the signal e_D+e_U1from the first frequency converting circuit 2 and the signal e_D+e_U2from the second frequency converting circuit 3 correspondingly take place in the secondary coils of the transformers 27, 28.

[0215] In this instance, the secondary coils of the transformers 27, 28 are arranged such that their terminals are coupled in parallel in a state that their polarities are coincident with those of the primary coils. Accordingly, a signal corresponding to the common component, i.e., the desired wave signal component e_D, of the signal e_D+e_U1from the first frequency converting circuit 2 and the signal e_D+e_U2from the secondary frequency converting circuit 3 which are generated in the secondary coils is obtainable at the terminal with no troubles such as the collision of the outputs of the secondary coils.

[0216] On the other hand, since the component e_U1from the first frequency converting circuit 2 which arises in the secondary coil of the transformer 27 and the component e_U2from the second frequency converting circuit 3 which arises in the secondary coil of the transformant 28 are different in frequency from each other, the components are mutually introduced into the mating secondary coils. At this time, the input impedances of the transformers viewed from the secondary coil sides becomes equal to the output impedances of the signal sources coupled to the primary coils, meanwhile, because the output impedances of the buffer amplifiers 25, 26 being the signal sources are set to very low values as noted before, the components other than the common component, i.e., the component e_U1from the first frequency converting circuit 2 which arises in the secondary coil of the transformer 27 and the component e_U2from the second frequency converting circuit 3 which arises in the secondary coil of the transformer 28, are reduced due to the low impedances.

[0217] In general, the buffer amplifier can be realized with an emitter-follower constructed using transistors. In the case of using this device, the output impedance of the buffer amplifier based on the wiring shown in FIG. 14 is below several ohms. This principle will be described with reference to FIG. 15. In FIG. 15, a transformer comprises two coils L₁and L₂, and the coil L₁is used as a primary coil, while the coil L₂is used as a secondary coil. The voltages and currents at the respective coil terminals are set as follows. Let it be assumed that the primary current is taken to be I₁, the secondary current is taken to be I₂, the voltage developed between the terminals of the primary coil L₁is taken as V₁, the voltage developed between the terminals of the secondary coil L₂and the mutual inductance between the primary coil L₁and the secondary coil L₂is taken as M. In this case, if a load Z is coupled to the secondary coil L₂, the input impedance Z in viewed from the terminal of the primary coil L₁is given by the following equation. $\begin{array}{cc}\mathrm{Zin}=\frac{V_1}{I_1}=\mathrm{j\omega }{L}_1+\frac{{\left(\omega M\right)}^2}{\mathrm{j\omega }{L}_2+Z}& \left(46\right)\end{array}$

[0218] where ω denotes a angular frequency, and L₁L₂are almost equal to M².

[0219] If the load Z is in a short-circuited state, there is, if Z=0, the input impedance Zin becomes zero as follows. $\begin{array}{cc}\mathrm{Zin}=\mathrm{j\omega }{L}_1+\frac{{\left(\omega M\right)}^2}{\mathrm{j\omega }{L}_2+Z_{{\searrow }_0}}\to \mathrm{j\omega }{L}_1-\frac{\mathrm{j\omega }{M}^2}{L_2}\approx \mathrm{j\omega }{L}_1-\mathrm{j\omega }{L}_1=0& \left(47\right)\end{array}$

[0220] Subsequently, if the load Z is set to an open state, that is, if Z=∞, the input impedance Zin becomes as follows, with the result that the impedance merely depends upon the inductance of only the primary coil.

[0221] Accordingly, the input impedance of the primary coils of the transformers 27 and 28 are affected by the loads of the secondary coils.

[0222] Returning back to FIG. 14, when the principle described with reference to FIG. 15 is applied thereto, the buffer amplifiers 25, 26 loads are in the short-circuited states, and the respective primary coils of the transformers 27, 28 operate in a stat that the input impedances of the secondary coils of the transformers 27, 28 are zero (short-circuited state). Accordingly, in FIG. 14 the signal currents i_U1and i_U2do not induce a voltage between the terminals of each of the secondary coils. In general, when the transformer is driven, the signal for the transformer is treated as a current, and the magnetic flux proportional to the product of that current and the inductance of the primary coil breaks out within the magnetic core of the transformer. A voltage is induced in the secondary coil in accordance with the time-variation (differential coefficient) of this magnetic flux. Now, when a potential induced at the terminals of the secondary coil L₂is taken as e₂, this potential e₂can be defined as follows. $\begin{array}{cc}{e}_2=M\frac{d{I}_1}{dt}& \left(49\right)\end{array}$