Radio frequency transmitter and methods thereof

[0002] Modern systems enable radio transmitters to transmit at reduced power for long periods of time. The modulating signal of these transmissions may have large peak-to-minimum amplitude variations. Since the efficiency of power amplifiers is generally reduced at less-than-maximum power levels, these two factors may increase the average current consumption of power amplifiers in radio transmitters.

[0003] There is a continuing need to reduce the current consumption of power amplifiers in radio transmitters.

[0004] The subject matter regarded as the invention is particularly pointed out and distinctly claimed in the concluding portion of the specification. The invention, however, both as to organization and method of operation, together with objects, features and advantages thereof, may best be understood by reference to the following detailed description when read with the accompanying drawings in which:

[0005] FIG. 1 is a schematic block-diagram illustration of an exemplary radio frequency transmitter, according to an embodiment of the present invention;

[0006] FIGS. 2A and 2B are schematic illustrations of signal space diagrams, helpful in understanding the present invention;

[0007] FIG. 3 is a schematic block-diagram illustration of an exemplary up-conversion chain, according to an embodiment of the present invention;

[0008] FIGS. 4A, 4B and 4C are exemplary graphical illustrations of the instantaneous efficiency of the radio frequency transmitter of FIG. 1 and of a conventional class-B power amplifier as a function of the instantaneous output signal power due to the amplitude of the modulating signal;

[0009] FIG. 5 is a schematic block-diagram illustration of an exemplary radio frequency transmitter, according to another embodiment of the present invention;

[0010] FIG. 6A is an exemplary graphical illustration of the instantaneous efficiency of the radio frequency transmitter of FIG. 1 for a constant envelope signal as a function of the output signal power, according to another embodiment of the present invention; and

[0011] FIG. 6B is an exemplary graphical illustration of the instantaneous efficiency of the radio frequency transmitter of FIG. 1 for a non-constant envelope signal as a function of the output signal power, according to a further embodiment of the present invention.

[0012] It will be appreciated that for simplicity and clarity of illustration, elements shown in the figures have not necessarily been drawn to scale. For example, the dimensions of some of the elements may be exaggerated relative to other elements for clarity. Further, where considered appropriate, reference numerals may be repeated among the figures to indicate corresponding or analogous elements.

[0013] In the following detailed description, numerous specific details are set forth in order to provide a thorough understanding of the invention. However it will be understood by those of ordinary skill in the art that the present invention may be practiced without these specific details. In other instances, well-known methods, procedures, components and circuits have not been described in detail so as not to obscure the present invention.

[0014] It should be understood that the present invention may be used in a variety of applications. Although the present invention is not limited in this respect, the circuit disclosed herein may be used in many apparatuses such as in the transmitters of a radio system. Radio systems intended to be included within the scope of the present invention include, by way of example only, cellular radiotelephone communication systems, two-way radio communication systems, one-way pagers, two-way pagers, personal communication systems (PCS), and the like.

[0015] Types of cellular radiotelephone communication systems intended to be within the scope of the present invention include, although are not limited to, Direct Sequence-Code Division Multiple Access (DS-CDMA) cellular radiotelephone communication systems, Wideband CDMA (WBCDMA) and CDMA2000 cellular radiotelephone systems, Global System for Mobile Communications (GSM) cellular radiotelephone systems, North American Digital Cellular (NADC) cellular radiotelephone systems, Time Division Multiple Access (TDMA) systems, Enhanced Data for GSM Evolution (EDGE) and Universal Mobile Telecommunications Systems (UMTS).

[0016] Reference is now made to FIG. 1, in which an exemplary radio frequency (RF) transmitter in accordance with an embodiment of the present invention is described. An RF transmitter 100 may comprise a digital signal processor (DSP) 102, baseband (BB) to RF up-conversion chains 104 and 106, RF preamplifiers 108 and 110, a power amplifier 112, an antenna 114 and a controller 116.

[0017] A BB input signal 118 may be provided to DSP 102, which may convert it into two constant envelope vectors according to a method which will be described hereinbelow with respect to FIGS. 2A and 2B. For example, the first constant envelope vector may be represented by baseband signals I₁and Q₁, while the second constant envelope vector may be represented by baseband signals I₂and Q₂. Up-conversion chain 104 may convert signals I₁and Q₁into an RF signal RF₁; similarly up-conversion chain 106 may convert signals I₂and Q₂into an RF signal RF₂. RF signals RF₁and RF₂have a common carrier frequency. An exemplary embodiment of up-conversion chains 104 and 106 is described hereinbelow with respect to FIG. 3, although the present invention is in no way limited to this particular exemplary embodiment.

[0018] RF preamplifier 108, which has a variable gain, may amplify signal RF₁to produce a signal RF_IN-1; similarly RF preamplifier 110, which has a variable gain, may amplify signal RF₂to produce a signal RF_IN-2. Power amplifier 112, which may have reactive termination, may amplify and combine RF_IN-1and RF_IN-2to produce an output signal RF_OUTfor transmission by antenna 114.

[0019] Power amplifier 112 may comprise two branch amplifiers 120 and 122 connected in parallel, and shunt reactance elements 124 and 126 at the output of branch amplifiers 120 and 122, respectively. B_Sdenotes the shunt reactance of element 124 and -B_Sdenotes the shunt reactance of element 126. The efficiency of power amplifier 112 at a specific output signal power may be improved by adjusting the shunt reactance B_S. Power amplifier 112 may also comprise a transmission-line-coupler 128 for combining the outputs of branch amplifiers 120 and 122. Transmission-line-coupler 128 may comprise two transmission lines 130 and 132 connected to antenna 114 so that the sum of the branch currents goes through the load. Other combiner schemes yielding the same performance may be implemented instead, namely hybrid BALUN, center tap inductor, etc.

[0020] Controller 116 may receive as input a targeted average output signal power level P. Targeted average output signal power level P may be selected from a range of power levels or may be selected from a discrete set of at least two power levels. Controller 116 may provide data related to P to any of DSP 102, up-conversion chains 104 and 106, and RF preamplifiers 108 and 110, with the result that power amplifier 112 may produce an output signal whose average power is substantially equivalent to P. The operation of controller 116 and DSP 102 is better understood if reference is made additionally to FIGS. 2A and 2B, which are schematic illustrations of signal space diagrams. The horizontal axis of the diagram represents the real (in-phase) component of a signal vector, while the vertical axis of the diagram represents the imaginary (quadrature) component.

[0021] Three concentric circles, 202, 204 and 206, are shown in FIG. 2A. A vector 208 from the center of the diagram to the largest circle 206 represents the amplitude and phase of a BB signal, which after up-conversion and amplification may produce a signal having a maximal instantaneous output signal power. This maximal instantaneous output signal power may be determined both by the maximum amplitude A_MAXof input signal 118 (FIG. 1) and by the maximum average power P_MAXthat power amplifier 112 may be able to produce. Similarly, a vector 210 from the center of the diagram to circle 202 represents the amplitude and phase of a BB signal, which after up-conversion and amplification may produce a signal at an instantaneous output signal power that may be determined both by the minimum amplitude A_MINof input signal 118 and by an average output signal power level P_TH, the determination of which will be explained hereinbelow.

[0022] According to some embodiments of the present invention, when the targeted average output signal power level P is between P_MAXand P_TH, controller 116 may provide DSP 102 with ρ(P) so that DSP 102 may represent a baseband vector 212 by two constant envelope vectors 214 and 216. Since baseband vector 212 has an average amplitude controlled by ρ(P), baseband vector 212 may result, after up-conversion and amplification, in a signal at an average output signal power P and at an instantaneous output signal power determined both by the instantaneous amplitude A(t) of input signal 118 and by the targeted average output signal power level P.

[0023] The radius of circle 204 is predetermined both by the maximum amplitude A_MAXand by ρ(P_MAX). The data flow from controller 116 to DSP 102 is indicated in FIG. 1 by line 133, and constant envelope vectors 214 and 216 are represented by the signals I₁and Q₁, and I₂and Q₂, respectively.

[0024] If BB input signal 118 at time t is denoted s(t), with the real (in-phase) component denoted I(t) and the imaginary (quadrature) component denoted Q(t), then the following decomposition holds:

[0025] s(t)=I(t)+jQ(t).

[0026] The instantaneous amplitude A(t) of input signal 118 at time t is given as follows:

[0027] A(t)={square root over (I²(t)+Q²(t))}.

[0028] Signals I₁and Q₁are then given by Equations 1A and 1B, as follows: $\begin{array}{cc}{I}_1\left(t\right)=\sqrt{\rho \left(P\right)}\left(I\left(t\right)-Q\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)·A_{\mathrm{MAX}}^2}{\rho \left(P\right)·A^2\left(t\right)}-1}\right),& \left(\text{Eq. 1A}\right)\\ Q_1\left(t\right)=\sqrt{\rho \left(P\right)}\left(Q\left(t\right)+I\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)·A_{\mathrm{MAX}}^2}{\rho \left(P\right)·A^2\left(t\right)}-1}\right),& \left(\text{Eq. 1B}\right)\end{array}$

[0029] and signals I₂and Q₂are given by Equations 2A and 2B, as follows: $\begin{array}{cc}{I}_2\left(t\right)=\sqrt{\rho \left(P\right)}\left(I\left(t\right)+Q\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)·A_{\mathrm{MAX}}^2}{\rho \left(P\right)·A^2\left(t\right)}-1}\right),& \left(\text{Eq. 2A}\right)\\ Q_2\left(t\right)=\sqrt{\rho \left(P\right)}\left(Q\left(t\right)-I\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)·A_{\mathrm{MAX}}^2}{\rho \left(P\right)·A^2\left(t\right)}-1}\right).& \left(\text{Eq. 2B}\right)\end{array}$

[0030] It will be appreciated by persons of ordinary skill in the art from Equations 1A, 1B, 2A and 2B that the amplitude of the signal represented by I₁and Q₁, namely {square root over (I₁²+Q₁²)}, and the amplitude of the signal represented by I₂and Q₂, namely {square root over (I₂²+Q₂²)}, are both equal to {square root over (ρ(P_MAX))}·A_MAX. It will also be appreciated by persons of ordinary skill in the art that the relative phase differences of these signals are determined from the instantaneous amplitude of input signal 118 and from the targeted average output signal power level P. Clearly the present invention is not limited in any way to the exemplary equations given hereinabove in Equations 1A, 1B, 2A and 2B. Rather, any other set of equations yielding a constant envelope signal represented by signals I₁and Q₁, and a constant envelope signal represented by signals I₂and Q₂, is clearly also within the scope of the present invention.

[0031] According to some embodiments of the present invention, when the targeted average output signal power level P is between P_MAXand P_TH, controller 116 may provide predetermined, fixed values to any amplification elements of up-conversion chains 104 and 106 and to RF preamplifiers 108 and 110. The data flow from controller 116 to up-conversion chains 104 and 106 are indicated in FIG. 1 by lines 134 and 136, respectively. Lines 138 and 140 indicate the data flow from controller 116 to RF preamplifiers 108 and 110, respectively.

[0032] Three concentric circles, 202, 204 and 218, are shown in FIG. 2B. Circles 202 and 204 are the same or similar to those shown in FIG. 2A. A vector 220 from the center of the diagram to circle 218 represents the amplitude and phase of a BB signal, which after up-conversion and amplification may produce a signal at an instantaneous output signal power that may be determined both by the maximum amplitude A_MAXof input signal 118 (FIG. 1) and by the average output signal power level P_TH.

[0033] According to some embodiments of the present invention, when the targeted average output signal power level P is less than P_TH, controller 116 may provide DSP 102 with the power ρ(P_TH) so that DSP 102 may represent a baseband vector 222 by two constant envelope vectors 224 and 226, where the size of constant envelope 204 is the same or similar to that used in FIG. 2A. Constant envelope vectors 224 and 226 may be represented by signals I₁and Q₁, and I₂and Q₂, respectively, where Equations 1A, 1B, 2A and 2B are used with ρ(P_TH) in place of ρ(P). However, baseband vector 222, after up-conversion and amplification at fixed gain values, would produce an output signal at an average output signal power, which may be determined both by the instantaneous amplitude A(t) of input signal 118 and by the predetermined power level P_TH, and which is higher than the targeted average output signal power level P. Therefore, controller 116 may reduce the amplitudes of signals I₁and Q₁, and I₂and Q₂, or may reduce the gain of any of variable amplification elements in up-conversion chains 104 and 106 and RF preamplifiers 108 and 110, or a combination thereof, with the result that power amplifier 112 may produce an output signal whose average power is substantially equivalent to P.

[0034] The predetermined average output signal power level P_THmay act as a threshold between two modes of operation of the RF transmitter, according to some embodiments of the present invention. In one mode, the RF transmitter may control the instantaneous output signal power by combining constant envelope signals whose relative phase differences are determined from the instantaneous amplitude of a baseband input signal and from the targeted average output signal power level P, and by up-converting at a fixed gain. In another mode, the RF transmitter may control the instantaneous output signal power by combining constant envelope signals whose relative phase differences are determined from the instantaneous amplitude of the baseband input signal and from the predetermined average output signal power level P_TH, and by up-converting at a variable gain which is dependent on the targeted average output signal power level P and which is lower than the fixed gain of the first mode. Alternatively, in this other mode, the RF transmitter may control the instantaneous output signal power by combining constant envelope signals whose relative phase differences are determined from the instantaneous amplitude of the baseband input signal and from the predetermined average output signal power level P_TH, and whose amplitudes have been reduced in the baseband according to the targeted average output signal power level P, so that the average power of the output signal is substantially equivalent to the targeted average output signal power level P.

[0035] Reference is now made to FIG. 3, which is a schematic block-diagram illustration of an exemplary up-conversion chain, according to an embodiment of the present invention. The up-conversion chain may comprise an intermediate frequency (IF) local oscillator (LO) 300 and an RF local oscillator 302, IQ modulators 304 and 306, and phase lock loops (PLL) 308 and 310.

[0036] IQ modulator 304 may comprise mixers 312 and 314 and combiner 316. Mixer 312 may receive as input I₁and sin (ω_IFt), where ω_IFdenotes the frequency generated by IF LO 300 and t denotes time. Mixer 314 may receive as input Q₁and cos (ω_IFt). Combiner 316 may combine the outputs of mixers 312 and 314, and provides the combination to PLL 308. Similarly, IQ modulator 306 may comprise mixers 318 and 320 and combiner 322. Mixer 318 may receive as input I₂and sin (ω_IFt). Mixer 320 may receive as input Q₂and cos (ω_IFt). Combiner 322 may combine the outputs of mixers 318 and 320, and provides the combination to PLL 310.

[0037] PLL 308 may comprise a phase detector (PD) 324, a loop filter 326 and a voltage-controlled oscillator (VCO) 328. PLL 308 may also comprise a mixer 330, mixing the output of VCO 328 with the signal produced by RF LO 302, and providing an IF modulated signal to PD 324. Similarly, PLL 310 may comprise a PD 334, a loop filter 336 and a VCO 338. PLL 310 may also comprise a mixer 340, mixing the output of VCO 338 with the signal produced by RF LO 302, and providing an IF modulated signal to PD 334.

[0038] Alternatively, the up-conversion chain may comprise variable amplifiers (not shown) that amplify the input signals I₁and Q₁, and I₂and Q₂, prior to their modulation by IQ modulators 304 and 306, respectively. The gain of these variable amplifiers may be reduced by controller 116 (not shown) when the targeted average output signal power level P is less than the predetermined power level P_TH.

[0039] Reference is now made to FIGS. 4A, 4B and 4C, which are exemplary graphical illustrations of the instantaneous efficiency of the radio frequency transmitter of FIG. 1 (indicated by a solid line) and of a conventional class-B power amplifier (indicated by a dotted line) as a function of the output signal power. In FIG. 4A the average output signal power (indicated by a circle) is P_MAX, and the instantaneous output signal power (indicated by the solid and dotted lines) varies according to the amplitude of the input signal. In FIG. 4B the average output signal power is P_TH, and in FIG. 4C the average output signal power is less than P_TH. The average current consumption of the RF transmitter of FIG. 1 may be appreciably reduced with respect to that of class-B power amplifiers.

[0040] As shown in the exemplary graphical illustrations of FIGS. 4B and 4C, P_THis chosen to be the average output signal power at which the efficiency has a peak value. However, it will be appreciated that there are many other ways to select the threshold P_TH, all of which are included in the scope of the present invention. For example, the threshold P_THmay be chosen by minimizing the current consumption according to the output signal power probability distribution and the amplitude distribution of the baseband input signal.

[0041] Reference is now made to FIG. 5, which is a schematic block-diagram illustration of an exemplary radio frequency transmitter, according to another embodiment of the present invention.

[0042] An RF transmitter 500 may comprise DSP 102, RF preamplifiers 108 and 110, power amplifier 112, antenna 114 and controller 116. As in FIG. 1, BB input signal 118 may be provided to DSP 102. RF transmitter 500 may also comprise IF local oscillator 300, RF local oscillator 302, IQ modulators 304 and 306, and PLLs 308 and 310.

[0043] RF transmitter 500 may also comprise a feedback path to compensate for circuit imperfections that may occur in an open loop arrangement such as that of FIG. 1. In this embodiment, DSP 102 may comprise a compensation module 502. A small portion of the transmitted signal RF_OUTmay be taken through a directional coupler 504 via a step attenuator 506. The state of step attenuator 506 may be controlled by controller 116, as indicated by line 507, in order to divide the entire dynamic range into several smaller regions. The output of step attenuator 506 passes through an image rejection mixer (IRM) 508. IRM 508 down-converts the RF signal to IF. IRM 508 may receive as input, in addition to the RF signal, a signal from RF local oscillator 302. The IF signal produced by IRM 508 may be demodulated by an I/Q demodulator 510, which may receive as input a signal from IF local oscillator 300. I/Q demodulator 510 may produce feedback signals I_FBand Q_FB, which may be provided to DSP 102 through analog-to-digital converters (not shown).

[0044] As indicated by line 133, controller 116 may provide DSP 102 with a power level ρ. As explained hereinabove, when the targeted average output signal power level P is in a first range of average output signal power levels, i.e. between P_MAXand P_TH, then the power level ρ is related to the targeted average output signal power level P. When the targeted average output signal power level P is in a second range of average output signal power levels, i.e. less than P_TH, then the power level ρ is related to the predetermined average output signal power level P_TH.

[0045] Compensation module 502 may compare the input signal 118, the feedback signals I_FBand Q_FB, the power level ρ and the state 507 of step attenuator 506 to create the compensated baseband signals I₁and Q₁, and I₂and Q₂.

[0046] RF transmitter 500 may also comprise a power level measurement unit 512 that may take a small portion of the output of step attenuator 506 through a directional coupler 514. Power level measurement unit 512 may provide a measured power level P_FBto controller 116. Controller 116 may compare the targeted output signal power level with measured power level P_FBin order to set the targeted amplification values for RF preamplifiers 108 and 110 and for the amplification elements in the up-conversion chains.

[0047] In another embodiment of the present invention, signals I₁and Q₁are given by Equations 3A and 3B, as follows: $\begin{array}{cc}{I}_1\left(t\right)=\sqrt{\rho \left(P\right)}\left(I\left(t\right)-Q\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)}{\rho \left(P\right)}-1}\right),& \left(\text{Eq. 3A}\right)\\ Q_1\left(t\right)=\sqrt{\rho \left(P\right)}\left(Q\left(t\right)+I\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)}{\rho \left(P\right)}-1}\right),& \left(\text{Eq. 3B}\right)\end{array}$

[0048] and signals I₂and Q₂are given by Equations 4A and 4B, as follows: $\begin{array}{cc}{I}_2\left(t\right)=\sqrt{\rho \left(P\right)}\left(I\left(t\right)+Q\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)}{\rho \left(P\right)}-1}\right),& \left(\text{Eq. 4A}\right)\\ Q_2\left(t\right)=\sqrt{\rho \left(P\right)}\left(Q\left(t\right)-I\left(t\right)\sqrt{\frac{\rho \left(P_{\mathrm{MAX}}\right)}{\rho \left(P\right)}-1}\right).& \left(\text{Eq. 4B}\right)\end{array}$

[0049] It will be appreciated by persons of ordinary skill in the art from Equations 3A, 3B, 4A and 4B that the amplitude of the signal represented by I₁and Q₁, namely {square root over (I₁²+Q₁²)}, and the amplitude of the signal represented by I₂and Q₂, namely {square root over (I₂²+Q₂²)}, are both equal to {square root over (ρ(P_MAX)·(I²+Q²))}. That is, their amplitude depends on the amplitude of the input signal and the maximal average output signal power and does not depend on the average output signal power. These signals are constant envelope signals only if the input signal is a constant envelope signal. It will also be appreciated by persons of ordinary skill in the art that the relative phase differences of these signals are determined from the targeted average output signal power level P and not from the instantaneous amplitude of input signal 118.

[0050] According to this embodiment, if the amplitude of baseband input signal 118 is constant, an exemplary graphical illustration of the efficiency of the radio frequency transmitter of FIG. 1 as a function of the output signal power is shown in FIG. 6A, to which reference is now made. The efficiency has a peak at two output signal powers. The predetermined average output signal power level P_THmay be set to be close to the lower of these output signal powers having a peak efficiency.

[0051] If the amplitude of baseband input signal 118 is not constant, the amplitudes of baseband signals I₁and Q₁, and I₂and Q₂, are determined from the instantaneous amplitude A(t) of baseband input signal 118, and the relative phase differences of baseband signals I₁and Q₁, and I₂and Q₂, are determined from the targeted output signal power level P. The average efficiency of the radio frequency transmitter of FIG. 1 for this embodiment as a function of the output signal power is shown in FIG. 6B.

[0052] While certain features of the invention have been illustrated and described herein, many modifications, substitutions, changes, and equivalents will now occur to those of ordinary skill in the art. It is, therefore, to be understood that the appended claims are intended to cover all such modifications and changes as fall within the true spirit of the invention.