Multilevel DC link inverter

[0003] The field of the invention is electronic controls for motors, including brushless DC motors, synchronous ac motors and switched reluctance motors.

[0004] Permanent magnet motors are excellent candidates for traction drives in electric/hybrid-electric vehicle applications because of their higher efficiency. Due to a long effective air gap, PM motors tend to have low inductance. Recent design techniques for high power PM motors (rated at several tens of kWs) for electric vehicle and hybrid electrical vehicle propulsion have used an iron-less stator structure to eliminate stator iron loss. This design also reduces stator inductance well below 100 μH. While these types of very low inductance PM motors have the advantages of fast current control response and a linear relationship between current and its developed torque, they impose stringent current regulation demands for the inverter to obtain acceptable current ripple, which should be typically below 5%.

[0005] A PM motor can be excited in an ac synchronous mode or brushless dc (BLDC) mode using a dc link inverter to control three-phase switching of current in the windings of a 3-phase motor. The latter excitation provides a low cost drive system, which is well suited for PM motors having a trapezoidal back EMF. There is, however, in the known inverter technology, a problem of unacceptable current ripple with low inductance PM motors.

[0006] The maximum current ripple is inversely proportional to the motor inductance and the inverter switching frequency. A standard inverter with the most commonly used IGBT switching devices will produce an unacceptable maximum current ripple of 25% due to the IGBT's switching frequency, which is limited to 20 kHz. This high current ripple not only causes additional motor losses but also requires that the inverter be rated higher to handle the high peak current.

[0007] For a given switching frequency, the current ripple can be reduced by adding an external inductor in each phase to increase the inductance. However, inductors rated for high current rating are bulky and this produces a large inductance which undesirably slows current control response. Another method of reducing current ripple uses a step-down chopper to regulate the dc voltage applied to the inverter. This also requires an additional inductor. These approaches have not resulted in a control with the desired features of operation and construction.

[0008] In order to reduce current ripple and use permanent magnet (PM) motors having very low inductance, the invention provides a multilevel voltage source for the dc source of a multiple phase bridge inverter. The multilevel voltage source has multiple voltage-controlled cells cascaded to provide the necessary voltage to the motor. To regulate motor current with low ripple, the output voltage of the cells is adjusted through dc voltage level stepping and pulse width modulation (PWM) depending on the amplitude of the motor back electromotive force (EMF).

[0009] It is a primary object of the invention to reduce current ripple in a PM motor powered by a dc link inverter to below 5% over the full speed range of the motor.

[0010] It is a further object of the invention to provide controls for brushless DC motors, AC synchronous motors and switched reluctance motors having low inductance.

[0011] Other objects and advantages of the invention, besides those discussed above, will be apparent to those of ordinary skill in the art from the description of the preferred embodiments which follows. In the description reference is made to the accompanying drawings, which form a part hereof, and which illustrate examples of the invention. Such examples, however are not exhaustive of the various embodiments of the invention, and therefore reference is made to the claims which follow the description for determining the scope of the invention.

[0012] FIG. 1 is a block diagram of a motor control of the present invention connected to control a brushless dc motor;

[0013] FIG. 1ais a detail schematic of PWM inverter of the prior art;

[0014] FIG. 1bis a graph of trapezoidal back emf and switching conduction periods vs. time for the PWM inverter of FIG. 1a;

[0015] FIG. 1cis a schematic of an equivalent circuit for the PWM inverter and the PM motor of FIG. 1a;

[0016] FIG. 1dis a current waveform for current ripple in inverter of FIG. 1a;

[0017] FIG. 2 is a graph of current ripple vs. speed for a specific brushless dc motor using a motor control of the prior art;

[0018] FIG. 3 is an electrical schematic of a dc link inverter using additional inductors to control current ripple;

[0019] FIG. 4 is an electrical schematic of a dc link inverter using a step down chopper circuit to control current ripple;

[0020] FIG. 5ais a first electrical schematic representation of a dc link inverter using a multilevel dc voltage source of the present invention;

[0021] FIG. 5bis a second electrical schematic representation of a dc link inverter using a multilevel dc voltage source of the present invention;

[0022] FIG. 5cis an equivalent circuit schematic for the multilevel dc voltage source of FIGS. 5aand 5b;

[0023] FIG. 5dis a graph of trapezoidal back emf and switching conduction periods vs. time for the PWM inverter of FIGS. 5aand 5b;

[0024] FIG. 6ais a more specific example of the multilevel dc voltage source of the present invention;

[0025] FIG. 6bis a graph of current ripple vs. speed for a conventional inverted of the prior art and for a multilevel dc voltage source inverter of the present invention;

[0026] FIG. 6cis a graph of the bus voltage and phase currents in a conventional PWM inverter of the prior art, showing the current ripple components;

[0027] FIG. 6dis a graph of the bus voltage and phase currents in a multilevel dc voltage source inverter of the present invention, showing the current ripple components;

[0028] FIG. 7ais an electrical schematic of a second embodiment of the present invention applied to a switched reluctance motor;

[0029] FIG. 7bis a graph of trapezoidal motor inductance and switching conduction periods vs. time for the PWM inverter of FIG. 7a;

[0030] FIG. 8ais an electrical schematic of a third embodiment of the present invention using thyristors in the PWM inverter;

[0031] FIG. 8bis a graph of trapezoidal back emf and switching conduction periods vs. time for the PWM inverter of FIG. 8ain motoring mode;

[0032] FIG. 8cis a graph of trapezoidal back emf and switching conduction periods vs. time for the PWM inverter of FIG. 8ain regenerating mode;

[0033] FIG. 9ais an electrical schematic of a fourth embodiment of the present invention using a diode clamped multilevel dc voltage source inverter; and

[0034] FIG. 9bis a graph of trapezoidal back emf and switching conduction periods vs. time for the PWM inverter of FIG. 9a.

[0035] Referring to FIG. 1, the present invention is embodied in a motor control system 10 for controlling a motor 11, which may be a permanent magnet (PM) motor, such as a brushless dc motor or an ac synchronous motor, or in another embodiment described herein, may be a switched reluctance motor. In permanent magnet (PM) motors, the permanent magnets are positioned on the rotor, while the stator has windings for carrying the phase currents. The permanent magnets are typically formed of rare earth magnetic materials such as NdFeB (neodymium-iron-boron). When the rotor of these motors rotates, the rotating magnets induce a back EMF voltage in the stator. The magnitude of the back EMF increases with the speed of the rotor. The waveform of the back EMF can be either sinusoidal in the case of a permanent magnet synchronous motor or it can be trapezoidal in the case of a brushless dc motor.

[0036] As further seen in FIG. 1, the motor control system 10 includes an inverter 15 connected to three phases of the PM motor 11. The inverter 15 receives dc power from a dc power source 16 to be described in more detail below. A controller 14 transmits gating signals to control the firing of semiconductor switches in the inverter 15 for the three phases of the motor 11. The controller 14 senses the dc voltage through a dc voltage sensor 17. The controller 14 also receives rotor position information from one or more optional position sensors 12, which may be coupled to the motor output shaft, or positioned near the rotor as described below. Some brushless dc motor controls are classified as sensorless, meaning that they do not utilize position sensors. The controller 14 also receives current feedback for current sensors 13 in two of the three phase supply lines to the motor. From this current information, the controller can calculate current in the third phase.

[0037] The logic controller 14 preferably includes a microelectronic CPU and associated program memory and data memory. A program is stored in the program memory and is executed by the CPU to perform a current control loop and other basic control methods well known in the art of motor control.

[0038] Motor current ripple for inverter 15aand brushless PM motor 11a(FIG. 1a) can be analyzed by an equivalent circuit (FIG. 1c). FIG. 1aillustrates a typical PWM inverter 15aof the prior art with six IGBT's (insulated gate bipolar transistors) S1-S6 which switch at appropriate intervals. Bypass diodes are connected in parallel to and across the respective six IGBT's S1-S6 to allow transient currents associated with switching the IGBT's to bypass the IGBT's. Position sensors 12asuch as Hall-effect devices are positioned inside the motor to sense rotor position and signals are transmitted to a controller 14awhich calculates speed and generates gate signals through gate driver circuit 19ato the inverter 15a. Current sensors 13aprovide signals to the controller 14aas part of the current loop regulation.

[0039] FIG. 1bshows the three phase-to-phase back EMFs, a-b, b-c and c-a, which would be developed during operation. Only two of the three phase stator windings are excited at one time. To properly commutate the stator currents, rotor position information is detected with position sensors 12aas mentioned above. There are six combinations of the stator excitation, S3-S5, S1-S5, S1-S6, S2-S6, S2-S4 and S3-S4 over a fundamental cycle with each combination lasting for a phase period of π/3, as depicted in FIG. 1b. The corresponding two switches in each period can perform pulse width modulation to regulate the motor current. To reduce current ripple, it is, however, often useful to have one switch doing PWM while keeping the other switch conducting. For such a PWM scheme, an equivalent circuit is given in FIG. 1c, where R_mand L_mare the per-phase resistance and inductance including the mutual inductance between the two conducting windings, respectively, the PWM switch is S_Wand the diode D is the diode of the other switch in the same phase leg of the PWM switch . The commutation overlap during mode transition can be ignored for low inductance motors and is therefore not considered in the equivalent circuit.

[0040] Ignoring the stator coil resistance, current ripple, defined as the peak deviation from the average current as shown in FIG. 1d, at steady state and continuous conduction mode can be determined by the following equation. $\begin{array}{cc}{I}_{\mathrm{m\_ripple}}\approx \frac{1}{4L_m{f}_{\mathrm{SW}}}\left(1-\frac{E_{\mathrm{bemf\_peak}}}{V_{\mathrm{dc}}}\right)E_{\mathrm{bemf\_peak}}& \left(1\right)\end{array}$

[0041] where

[0042] f_sw: inverter switching frequency, f_sw=1/Tsw

[0043] V_dc: inverter dc link voltage,

[0044] E_bemfpeak: peak phase-to-phase back EMF.

[0045] Assuming the back EMF is linearly related to the motor speed, N, by E_bemfpeak=K_bemfN, where K_bemfis a constant determined by the motor, equation (1) can be rewritten as $\begin{array}{cc}{I}_{\mathrm{m\_ripple}}\approx \frac{1}{4L_m{f}_{\mathrm{SW}}}\left(1-\frac{K_{\mathrm{bemf}}N}{V_{\mathrm{dc}}}\right)K_{\mathrm{bemf}}N.& \left(2\right)\end{array}$

[0046] The maximum current ripple can be determined by $\begin{array}{cc}{I}_{\mathrm{m\_ripple}\left(\max \right)}\approx \frac{V_{\mathrm{dc}}}{16L_m{f}_{\mathrm{SW}}}\mathrm{at}N=\frac{V_{\mathrm{dc}}}{2K_{\mathrm{bemf}}}.& \left(3\right)\end{array}$

[0047] The maximum current ripple is inversely proportional to the motor inductance and the inverter switching frequency. A plot of current ripple as a percentage of the rated current vs. speed is shown in FIG. 2 for a 30 kW BLPM motor having a rated current of 110 amps and a phase inductance of Lm=37.5 μH with the inverter switching at 20 kHz and V_dc=325V. It shows that a standard inverter with the most commonly used IGBT switching devices will produce an unacceptable maximum current ripple of 25% due to the IGBT's limited switching frequency, which is limited to 20 kHz. This high ripple current percentage I_{ripple(peak)[%]} not only causes additional motor losses but also requires that the inverter be rated higher than otherwise would be to handle high peak current. For a given switching frequency, one can reduce the current ripple by adding external inductors, L₁, L₂, and L₃, to increase the inductance as shown in FIG. 3. However, inductors rated for high current rating are bulky and a large inductance undesirably slows current control response. FIG. 4 shows another way to reduce current ripple by using a step-down chopper including seventh IGBT S7, diodes D7 and D8 and inductor L₄to regulate the dc voltage. This requires the additional inductor, L₄.

[0048] FIG. 5ashows the motor control of the present invention, which comprises an incremental dc supply circuit 16eand a multiphase bridge inverter 15e. The multilevel dc supply circuit 16eis formed by connecting a number of cells #1, #2, . . . #n in series with each cell having a voltage source Vs controlled by two control switches Sa and Sb. The two switches, Sa and Sb, operate in a toggle fashion. The cell source is bypassed with Sa on and Sb off or adds to the dc link voltage by reversing the switches.

[0049] To control a PM motor 11ein BLDC mode, the bridge inverter 15eis used only to commutate the motor phase currents without doing PWM for current regulation. The current regulation is performed by the cell switches. For a given range of back EMF defined by its minimum E_bemfPeak(min) and maximum E_bemfPeak(max), k cells are active but only one of the k number of cells, performs PWM, as shown in FIG. 5b. The required number of active cells, k, is determined by $\begin{array}{cc}\frac{E_{\mathrm{bemf\_Peak}}\left(\max \right)}{V_s}<k<\frac{E_{\mathrm{bemf\_Peak}}\left(\min \right)}{V_s}+1.& \left(4\right)\end{array}$

[0050] where Vs is the source voltage of each cell. Since n cells cover the full voltage range, and with 1<=k<=n, the number of active cells increases with motor speed.

[0051] Alternatively, for a given number of cells, the controllable speed range is defined by $\begin{array}{cc}\frac{\left(k-1\right)V_s}{K_{\mathrm{bemf}}}<N<\frac{k{V}_s}{K_{\mathrm{bemf}}}.& \left(5\right)\end{array}$

[0052] The current ripple can be derived based on the equivalent circuit shown in FIG. 5cas follows. $\begin{array}{cc}{I}_{\mathrm{m\_ripple}}\approx \frac{k^2{V}_s}{4L_m{f}_{\mathrm{SW}}}\left(1-\frac{K_{\mathrm{bemf}}N}{k{V}_s}\right)\left(\frac{K_{\mathrm{bemf}}N}{k{V}_s}-\frac{k-1}{k}\right)& \left(6\right)\end{array}$

[0053] The maximum current ripple can be determined by $\begin{array}{cc}{I}_{\mathrm{m\_ripple}\left(\max \right)}\approx \frac{V_s}{16L_m{f}_{\mathrm{SW}}}\mathrm{at}N=\frac{\left(2k-1\right)V_s}{2K_{\mathrm{bemf}}}.& \left(7\right)\end{array}$

[0054] From equations (3) and (7), the maximum current ripple is reduced by a factor of Vdc/Vs, i.e. the number of cells. FIG. 5dshows the back EMF and switching conduction periods for the inverter of FIGS. 5aand 5b.

[0055] FIG. 6ashows an embodiment of the invention using power MOSFETS as switches 5aand 5bin each cell. IGBT's are used in the inverter 15f. Current and position signals are transmitted from current sensors 13fand position sensors 12fto controller 14f, which then calculates speed and current commands and transmits commutation signals through IGBT gate drivers 19f. The controller 14fcontrols the number of voltage cells 16fwhich are operated in series by transmitting signals to the cell controller 21f(Sa1-Sa5 and Sb1-Sb5) through MOSFET gate drivers 20f.

[0056] FIG. 6bshows curves of the calculated current ripple for this motor 11fwith the conventional inverter 15ashown in FIG. 1a, and then with the five-increment dc link inverter 15fshown in FIG. 6a. The maximum current ripple is reduced by a factor of 5. It is noted that since the cell voltage is low, the cell switches 21fcan be power MOSFETs as shown in FIG. 6a. This provides an additional option for ripple reduction by switching at a higher frequency, f_sw, as shown in equations (3) and (7). It is also useful to rotate the active cells so that the same amount of average power is drawn from each cell source. FIG. 6cshows the current ripple corresponding to the non-incremental dc voltage supply circuit 16ain FIG. 1aand FIG. 6dshows the current ripple for an incremental dc voltage supply circuit 16fof FIG. 6a.

[0057] While current ripple is best reduced when the voltage cells provide equal dc voltage, the invention would also reduce ripple in some measure where the voltage was divided into unequal increments.

[0058] The concept can also be applied to switched reluctance motor drives. FIG. 7ashows an example of the invention for switched reluctance motor drives, in which a five-increment dc supply circuit 16gand 21gis employed to power a three-phase reluctance motor 11grepresented by inductors La, Lb and Lc. FIG. 7billustrates operating waveforms and switch gating signals of the inverter 15g. Switches S1-S6 are used to commutate the stator currents according to the rotor position, θ and motor current regulation is accomplished by selecting an adequate number, k (where k=1, 2, 3, 4 or 5), of active cells according to the motor speed and by having one of the active cells performing pulse width modulation. The remaining inactive cells are bypassed by turning on switches Sa (k+1 to 5) while maintaining their counterparts Sb (k+1 to 5) in an off condition.

[0059] FIG. 8ashows an alternative configuration of a five-level dc supply circuit 16h, 21hemploying MOSFETs, Sa1-Sa5, Sb1-Sb5, Sc1-Sc5, Sd1-Sd5, and an inverter 15husing thyristors, T1-T6. FIG. 8billustrates back EMF, bus voltage and switch gating signal waveforms when the lower three voltage cells are active with the middle cell performing PWM and the motor 11his in motoring mode. The bus voltage Vbus is positive. FIG. 8cillustrates back EMF, bus voltage and switch gating signal waveforms when the lower three cells are active with the middle cell doing PWM but the motor 11his in a regeneration mode (the motor functions as a generator as power is fed back from the motor to the power supply). In this mode, the bus voltage Vbus is negative. The thyristor bridge inverter 15his used to commutate the stator current and the current control is accomplished with the selection of active cell number and the pulse width modulation of one of the active cells.

[0060] FIG. 9ashows an alternative configuration using a diode clamped multilevel dc voltage supply circuit 16i, 21i. The five capacitors, C1-C5 equally divide the dc source voltage, Vdc. FIG. 9billustrates back EMF, bus voltage and switch gating signal waveforms when the lower three capacitors, C1-C3 are active.

[0061] This has been a description of the preferred embodiments of the invention. The present invention is intended to encompass additional embodiments including modifications to the details described above which would nevertheless come within the scope of the following claims.