Threshold voltage-independent MOS current reference

[0002] (1) Field of the Invention

[0003] The invention relates to a current reference circuit, and more particularly, to a threshold voltage-independent MOS current reference circuit.

[0004] (2) Description of the Prior Art

[0005] Current and voltage reference circuits are widely used in analog designs. A particularly difficult problem encountered in MOS reference circuit designs is caused by the large variation in threshold voltage (V_th) that often occurs in CMOS processing. Since the voltage-to-current transfer response of the MOS transistor depends on the value of V_th, large variations in V_thcan cause large variations in the actual current or voltage output of the reference circuit. It is desirable, therefore to eliminate V_thdependence in the reference output.

[0006] However, prior art attempts to eliminate the V_thcomponent typically rely on complicated voltage addition techniques to create a V_x+V_th. These techniques create several problems due to the use of differing operation points, or modes, for different MOS devices. Therefore, mismatch problems are a major drawback.

[0007] Several prior art inventions describe voltage or current reference circuits. U.S. Pat. No. 5,739,682 to Kay describes a reference substantially independent of the threshold voltage of the transistor providing the reference. A pair of MOS transistors has gate voltages made equal. The current through the first transistor is very small. The current through the second transistor is equal to the first current multiplied by a scaling factor. Since the first current is so small, the second current through the second transistor is essentially not dependent upon the threshold voltage. U.S. Pat. No. 5,910,749 to Kimura teaches a current reference with no temperature dependence. Both bipolar and MOS embodiments are disclosed. U.S. Pat. No. 4,723,108 to Murphy et al describes a circuit to compensate for MOS transistor performance changing over temperature and manufacturing variation. Changing Vth, caused by temperature, is compensated by changing the mobility in the opposite direction. The gate drive of a MOS device is thereby compensated. U.S. Pat. No. 5,315,230 to Cordoba et al teaches a reference voltage generator circuit that compensates for temperature and VCC variation.

[0008] A principal object of the present invention is to provide an effective and very manufacturable current reference circuit.

[0009] A further object of the present invention is to provide a current reference circuit comprising MOS devices.

[0010] A still further object of the present invention is to provide an MOS current reference circuit that is independent of the threshold voltage to thereby reduce reference current variation due to processing variation.

[0011] Another still further object of the present invention is to provide a nearly zero temperature coefficient current references using this novel MOS current reference circuit.

[0012] In accordance with the objects of this invention, a new current reference circuit is achieved. This current reference circuit uses MOS transistors. However, the reference value does not depend upon the threshold voltage. The circuit comprises, first, a first MOS transistor having gate, drain, and source. A gate voltage value is coupled from the gate to the source. A second MOS transistor has gate, drain, and source. The second MOS transistor is of the same size and type as the first MOS transistor. The source is coupled to the first MOS transistor source. The gate voltage value plus a delta voltage value is coupled from the gate to the source. A means is provided for forcing a drain voltage value from the drain to the source of the first MOS transistor and from the drain to the source of the second MOS transistor. The first MOS transistor and the second MOS transistor conduct drain currents in the linear mode. Finally, a means is provided for subtracting the first MOS transistor drain current from the second MOS transistor drain current to thereby create a current reference value. The current reference value does not depend upon the threshold voltage of the first and second MOS transistors. The circuit may be further applied to create a nearly zero temperature coefficient current reference.

[0013] In the accompanying drawings forming a material part of this description, there is shown:

[0014] FIG. 1 illustrates a first preferred embodiment of the present invention.

[0015] FIG. 2 illustrates a first preferred embodiment of the present invention, using NMOS transistors, including a means of forcing a drain voltage and a means of subtracting the drain currents to thereby create the current reference.

[0016] FIG. 3 illustrates the second preferred embodiment of the present invention, using PMOS transistors, including a means of forcing a drain voltage and a means of subtracting the drain currents to thereby create the current reference.

[0017] FIG. 4 illustrates the application of the present invention in a nearly zero temperature coefficient current reference circuit.

[0018] FIG. 5 illustrates the current versus temperature performance of the nearly zero temperature coefficient current reference circuit.

[0019] FIG. 6 illustrates an exemplary circuit for creating a positive voltage coefficient voltage depending upon the thermal voltage (V_T).

[0020] The preferred embodiments disclose the novel current reference circuit of the present invention. In the first embodiment, a matched pair of NMOS transistors is used to create the threshold voltage-independent current reference. In the second preferred embodiment, a matched pair of PMOS transistors is used in an inverted version of the present invention. Finally, the invention is applied to a near zero temperature coefficient (TC) current reference. It should be clear to those experienced in the art that the present invention can be applied and extended without deviating from the scope of the present invention.

[0021] Referring now to FIG. 1, a first preferred embodiment of the present invention is illustrated. Several important features of the invention are shown. The circuit comprises a matching pair of MOS transistors, N110 and N214. Each transistor, N110 and N214, is of the same type and size, and more preferably, is oriented in the same layout direction. Even though the novel technique of the invention eliminates V_thfrom the current reference final value, other parameters, such as mobility, or μ_o, and gate capacitance, C_ox, should still be made to match as closely a possible between the two transistors. A significant advantage of the present invention is the elimination of the V_thdependence in the current reference. By comparison, μ_oand C_oxprocess variance is found to be much less than that of V_th.

[0022] The first MOS transistor, N110, has a gate voltage value, V₁26, coupled from the gate to the source. The second MOS transistor, N214, has the source coupled to the first MOS transistor source at the V_SSnode 42. A second gate voltage value, V₂30, is coupled from the gate to the source of N214. The second gate voltage value, V₂30, comprises the first gate voltage value, V₁26, plus a delta voltage value, ΔV.

[0023] A means is provided for forcing a drain voltage value, V_D34 and 38, from the drain to the source of the first MOS transistor, N110, and from the drain to the source of the second MOS transistor, N214. Most importantly, both transistors, 10 and 14, are biased to operate in the linear mode. To insure that both devices are in the linear mode, the gate voltages, V₁26 and V₂30, are much larger than the drain voltage, V_D34 and 38. In the linear mode, a direct relationship exists between the gate voltage and the drain current as given by:

[0024] I_D=(μ_oC_oxW/L)(V_G−V_th−V_D/2)V_D,

[0025] where W/L is the width to length ratio. In this mode, the gate voltages must be larger than the threshold voltage to insure that both transistors are in strong inversion. The first MOS transistor, N110, generates a current, I₁. The second MOS transistor, N214, generates a current, I₂.

[0026] Finally, a means, 18, is provided for subtracting the first MOS transistor N110 drain current I₁from the second MOS transistor N214 drain current I₂to thereby create a current reference value, I_REF. The subtracting means 18 creates the current reference output, I_REF, where I_REF=I₂−I₁.

[0027] Substituting the gate and drain voltage values into the linear mode drain current equation, we find:

[0028] I₁=(μ_oC_oxW/L)(V₁−V_th−V_D/2)V_D,

[0029] and

[0030] I₂=(μ_oC_oxW/L)(V₁+ΔV−V_th−V_D/2)V_D.

[0031] Since, I_REF=I₂−I₁, we can solve the drain equations for I_REF, yielding:

[0032] I_REF=(μ_oC_oxW/L)(ΔV)V_D.

[0033] We note from this result that the V_thterm has been canceled. Therefore, the resulting current reference value does not depend on the threshold voltage. Since the resulting reference does still depend upon both mobility and gate capacitance, I_REFis also called I_μCox.

[0034] Referring now to FIG. 2, the first preferred embodiment is illustrated in greater detail to show a realized circuit implementation of the invention concept. The matched NMOS transistor pair comprises N150 and N254. Once again, the sources of N1 and N2 are coupled together while the gates are coupled to V₁and V₁+ΔV such that the gate drive differs by the delta voltage, ΔV. The gate voltages, V₁and V₁+ΔV, are biased much higher than the drain voltage, V_D, so that the MOS devices are operating in the linear mode.

[0035] The means to force the drain voltage value, V_D34 and 38, from the drain to the source of both N1 and N214 is provided by two voltage followers comprising the operation amplifiers 74 and 78 and the output transistors, N366 and N470. Due to the large input impedance and the high gain of the operation amplifiers 74 and 78, the drain voltages, V_D1and V_D2are guaranteed to be driven to the reference drain voltage value, V_D82. Further, the voltage follower arrangement isolates the drain reference voltage, V_D, from the actual drains of the first and second MOS transistors, N150 and N254.

[0036] The means for subtracting the drain currents, I₁and I₂, is provided by the PMOS transistors, P190, P294, P398, and P4102. The gate and drain of P190 are coupled together and further coupled to the gate of P294 at the node A 106. P190 and P294 are the same type of device and are the same size. Further, the sources of P190 and P294 are coupled together at V_CC118. Therefore, P190 and P294 form a current mirror. Since P190 must conduct I₁, the mirror configuration causes P294 to likewise conduct a drain current of I₁.

[0037] MOS transistors P398 and P4102 form a second current mirror. Once again, the gate and drain of P398 are coupled together and further coupled to the gate of P4102. P398 and P4102 are another matched pair. Therefore, the drain current of P398 is mirrored by the drain current of P4102.

[0038] As an important feature, the drain of P398 is coupled to the drain of P294 at node B 110. As discussed above, the greater gate drive (V₁+ΔV) on N254 creates a drain current, I₂, which is larger than the drain current I₁of N150. Because P294 is biased to conduct only I₁, P398 will conduct the difference between I₁and I₂. Therefore, the P398 current is given by I₂-I₁. Finally, the P3 current is simply mirrored to the output current reference as I₂-I₁. As shown above, the subtraction of I₂from I₁effectively eliminates the V_thterm from the output current, I_μCox.

[0039] Referring now to FIG. 3, the second preferred embodiment of the present invention is illustrated. In this case, the circuit is inverted such that the main mirroring devices comprise the PMOS transistors P1216 and P2220. The analysis of operation of the circuit is the same as for the first embodiment of FIG. 2. In this second embodiment case of FIG. 3, the output current reference, I_μCox, is a sinking current rather than a sourcing current as in FIG. 2.

[0040] Referring now to FIG. 4, an important application of the voltage-threshold independent current reference of the present invention is illustrate. In this application, the novel circuit is used to create a nearly zero temperature coefficient (TC) current source.

[0041] First, a first voltage-threshold independent current reference 304 is used to form a positive temperature coefficient current reference circuit 304. The gate voltage for the voltage-threshold independent current reference 304 comprises a positive temperature coefficient value. The delta voltage value, ΔV 328, comprises a positive temperature coefficient value, mV_Twhere V_Tis the thermal voltage and m is a constant. The drain voltage value, V_D324, comprises another positive temperature coefficient value, kV_T, where k is another constant.

[0042] Once again, the output of the current reference 304 is given by:

[0043] I_REF=(μ_oC_oxW/L)(ΔV)V_D.

[0044] Since ΔV=mV_Tand V_D=mV_T, the reference current becomes:

[0045] I_REF=(μ_oC_oxW/L)mk(V_T)².

[0046] It is known that the mobility, μ_o, of the transistor varies as (T)^−3/2, where T is temperature. It is also known that V_Tvaries as (T)¹. Therefore, the reference current, I_PTC, for the positive current reference 304 varies as (T)^1/2.

[0047] Second, a second voltage-threshold independent current reference 300 is used to form a negative temperature coefficient, current reference circuit 300. The gate voltage for the voltage-threshold independent current reference 300 comprises a negative temperature coefficient value. The delta voltage value, ΔV 320, comprises a negative temperature coefficient value, V_BG/n, where V_BGis a bandgap voltage and n is a constant. The drain voltage value, V_D324, again comprises a positive temperature coefficient value, kV_T, where k is a constant. The current reference value output by the circuit 300 comprises a negative temperature coefficient current reference value, I_ZTC.

[0048] Referring again to the current relation, the output of the current reference 300 is given by:

[0049] I_REF=(μ_oC_oxW/L)(ΔV)V_D.

[0050] Since ΔV=(V_BG)/n and V_D=mV_T, the reference current becomes:

[0051] I_REF=(μ_oC_oxW/L)(V_BG)/n)(V_T).

[0052] Once again, the mobility, μ_o, of the transistor varies as (T)^−3/2, and V_Tvaries as (T)¹. However, the bandgap voltage, V_BG)/n does not significantly vary with T. Therefore, the reference current, I_NTC, for the negative current reference 300 varies as (T)^−1/2.

[0053] A means is provided for adding the positive temperature coefficient current reference value, T_PTC, and the negative temperature coefficient current reference value, I_NTC, to thereby obtain a nearly zero temperature coefficient current reference, I_ZTC. The adding means preferably comprises the current mirror circuit comprising the matching devices, N5308 and N6312. The gate and drain of N5308 are coupled together and further coupled to the gate of N6312 at the node C 332. The sources of N5308 and N6312 are coupled together such that a common gate-to-source voltage is obtained. The drain of N5308 is further coupled to the current reference outputs of the current reference circuits 300 and 304. The positive temperature coefficient current reference value, I_PTC, and the negative temperature coefficient current reference value, I_NTC, are added together to create the zero TC reference, I_ZTC, as the drain current of N5. This current, I_ZTC, is mirrored to the output, OUT 336, by N6.

[0054] Referring now to FIG. 5, the temperature performance of the nearly zero TC current reference of FIG. 4 is illustrated. Note, that the combined current, I_ZTC, is given by:

[0055] I_ZTC=I_PTC+I_NTC.

[0056] Further, substituting into the reference equation once again, the zero TC current is given by:

[0057] I_REF=(μ_oC_oxW/L)[mV_T+(V_BG)/n)](V_T).

[0058] Differentiating this equation with respect to temperature and setting the result to zero results in:

[0059] V_BG/n=mV_T,

[0060] where temperature is T_oat the zero slope point.

[0061] Referring again to FIG. 5, the response graph 350 shows how the output current source varies over temperature. The derivative zero indicates the point of zero slope at T_o. This is the desired operating point for the nearly zero TC circuit of FIG. 4. Further, this operating point can be selected at any fixed by setting the current and geometry of the MOS devices.

[0062] Referring now to FIG. 6, an exemplary circuit for deriving the mV_Tand kV_Tvoltages is illustrated. This circuit is well known in the art. The current mirror created by P1400 and P2404 is matched such that I₁is the drain current of both P1 and P2. N1412 and N2416 are operated in weak inversion such that the drain current is exponentially proportional to the drain voltage. Note that N2 is scaled from N1 at a ratio given by the constant C. The voltage drop across the first resistor, R1, is given by:

[0063] V_R1=ln(C)V_T.

[0064] Therefore, since P3408 is scaled from P2404 by the ratio given by the constant A, then the current flowing through the second resistor, R₂, is given by:

[0065] I₂=(Aln(C)V_T)/R₁.

[0066] Finally, since R₂is scaled from R₁by the constant B, then the voltage drop across the second resistor, R₂, is given by:

[0067] V_R2=ABln(C)V_T.

[0068] Therefore, V_R1and V_R2may be used for kV_Tand mkV_T.

[0069] The present invention provides a unique and advantageous current reference circuit. The unique configuration eliminates dependence on the threshold voltage to improve performance. Further, the simplicity of the scheme means that the circuits are stable, effective at low power levels, and space efficient. An effective and very manufacturable current reference circuit is achieved. The current reference circuit comprises all MOS devices. The MOS current reference circuit is not dependent upon the threshold voltage, and this reduces reference current variation due to processing variation. Finally, a nearly zero temperature coefficient current reference is achieved using this novel MOS current reference circuit.

[0070] As shown in the preferred embodiments, the novel current reference circuit provides an effective and manufacturable alternative to the prior art.

[0071] While the invention has been particularly shown and described with reference to the preferred embodiments thereof, it will be understood by those skilled in the art that various changes in form and details may be made without departing from the spirit and scope of the invention.