Switching power supply device for suppressing an increase in ripple output voltage

[0002] 1. Field of the Invention

[0003] The present invention relates to a switching power supply device.

[0004] 2. Description of the Prior Art

[0005] An example of a conventional switching power supply device is shown in FIG. 11. The switching power supply device includes a converter portion 100 and a controller portion 200. The switching power supply device converts a direct-current voltage fed in via an input terminal IN into a desired direct-current voltage, and then supplies the thus obtained direct-current voltage to a load resistor RL.

[0006] First, the configuration of the converter portion 100 will be described. The converter portion 100 includes a capacitor 1, an NPN-type transistor 2, a diode 3, a coil 4, and an output capacitor 5, which together constitute a step-down type DC—DC converter. The collector of the transistor 2 is connected to the input terminal IN and to one end of the capacitor 1. The emitter of the transistor 2 is connected to the cathode of the diode 3 and to one end of the coil 4.

[0007] The end of the coil 4 which is not connected to the transistor 2 is connected to the output capacitor 5, is connected also via an output terminal OUT to the load resistor RL, and is connected also via the output terminal OUT to the resistor R1 (described later) provided in the controller portion 200. The end of the capacitor 1 which is not connected to the transistor 2, the anode of the diode 3, the end of the output capacitor 5 which is not connected to the coil 4, and the end of the load resistor RL which is not connected to the output terminal OUT are each grounded.

[0008] Next, the configuration of the controller portion 200 will be described. The controller portion 200 includes an output voltage detection circuit 6, an error amplifier 7, a reference voltage source 8, an operational amplifier 9, an oscillator 10, and a driver circuit 11. The output voltage detection circuit 6 is composed of a resistor R1 and a resistor R2 that is connected in series with the resistor R1. One end of the resistor R1 is connected to the output terminal OUT, and the end of the resistor R2 which is not connected to the resistor R1 is grounded. The node between the resistors R1 and R2 is connected to the inverting input terminal of the error amplifier 7. The non-inverting input terminal of the error amplifier 7 is connected to the reference voltage source 8.

[0009] The output terminal of the error amplifier 7 is connected to the non-inverting input terminal of the operational amplifier 9. The inverting input terminal of the operational amplifier 9 is connected to the oscillator 10. The output terminal of the operational amplifier 9 is connected through the driver circuit 11 to the base of the transistor 2.

[0010] Next, the operation of the switching power supply device configured as described above will be described. The direct-current voltage fed in via the input terminal IN is first smoothed by the capacitor 1 so as to be formed into an input voltage V_IN, and is then converted into a pulse voltage by the switching operation of the transistor 2.

[0011] When the transistor 2 is in an on state, a current flows from the input terminal IN to the coil 4. As a result, energy is not only accumulated in the coil 4, but also supplied to the load resistor RL. On the other hand, when the transistor 2 is in an off state, the energy accumulated in the coil 4 is supplied through the diode 3 to the load resistor RL. Here, to the output terminal OUT is supplied an output voltage V_Osmoothed by the output capacitor 5, and this output voltage V_Ois applied to the load resistor RL.

[0012] The output voltage V_Oof the switching power supply device is fed via the output terminal OUT to the controller portion 200 so as to be subjected to feedback control performed by the controller portion 200. Specifically, according to the output voltage V_Oof the switching power supply device, the duty factor, i.e. the ratio of the on periods to the sum of the on and off periods, of the pulse voltage output from the transistor 2 is determined. The output voltage V_Oof the switching power supply device is first divided by the output voltage detection circuit 6. The thus divided voltage V_adjis then compared with a reference voltage V_ref(=1.25 V) output from the reference voltage source 8 by the error amplifier 7.

[0013] The error amplifier 7 amplifies the difference between the divided voltage V_adjand the reference voltage V_ref, and outputs an output voltage signal V_Ato the operational amplifier 9. The operational amplifier 9, in synchronism with the output voltage V_OSC(a triangular wave) of the oscillator 10, outputs a PWM signal V_PWMcorresponding to the output voltage signal V_A. Specifically, when the output voltage signal V_Afrom the error amplifier 7 is higher than the output voltage V_OSCfrom the oscillator 10, the operational amplifier 9 outputs a high level as the PWM signal V_PWM, and otherwise, i.e. when the output voltage signal V_Afrom the error amplifier 7 is not higher than the output voltage V_OSCfrom the oscillator 10, the operational amplifier 9 outputs a low level as the PWM signal V_PWM. Here, the frequency of the output voltage V_OSC(a triangular wave) oscillated by the oscillator 10 is set to be 100 kHz to prevent audible noise. Moreover, the maximum and minimum levels of the output voltage V_OSC(a triangular wave) oscillated by the oscillator 10 are set to be 1.75 V and 0.75 V, respectively.

[0014] The PWM signal V_PWMis fed to the driver circuit 11, and the driver circuit 11, according to the PWM signal V_PWM, supplies a current to the base of the transistor 2 and thereby controls the switching operation of the transistor 2. Specifically, when the driver circuit 11 receives a high level as the PWM signal V_PWMfrom the operational amplifier 9, it feeds a current I_Bto the base of the transistor 2 to bring the transistor 2 into an on state. On the other hand, when the driver circuit 11 receives a low level as the PWM signal V_PWM, it turns the current I_Bsupplied to the base of the transistor 2 to zero and thereby brings the transistor 2 into an off state. In this way, the ratio of the on periods t_ONto the off periods t_OFFof the transistor 2 is controlled in such a way that the output voltage V_Oof the switching power supply device which is supplied to the load resistor RL is stabilized at a predetermined level (5 V). The duty factor “duty” of the PWM signal V_PWMand of the transistor 2 is given by formula (1) below. $\begin{array}{cc}\mathrm{duty}=\frac{t_{\mathrm{ON}}}{t_{\mathrm{ON}}+t_{\mathrm{OFF}}}\times 100=\frac{V_O}{V_{\mathrm{IN}}}\times 100& \left(1\right)\end{array}$

[0015] In the on periods t_ON, in which the transistor 2 is in an on state, the gradient of the current I_Lthat flows through the coil 4 is positive, and, in the off periods t_OFF, in which the transistor 2 is in an off state, the gradient of the current I_Lthat flows through the coil 4 is negative.

[0016] To cope with this, as described earlier, a voltage smoothed by the output capacitor 5 is supplied as the output voltage V_Oto the load resistor RL. However, equivalent series resistance (hereinafter referred to as ESR) exists in the output capacitor 5, and therefore the output voltage Vo contains a ripple voltage V_rms, i.e. an alternating-current component. FIG. 12 shows a time chart of the output voltage signal V_Afrom the error amplifier 7, the output voltage V_OSCfrom the oscillator 10, and the PWM signal V_PWMas observed at room temperature (25° C.). The frequency of the PWM signal V_PWMis equal to that of the output voltage V_OSCfrom the oscillator 10, and therefore the switching frequency f₀of the transistor 2 is equal to the frequency of the output voltage V_OSCfrom the oscillator 10, i.e. 100 kHz.

[0017] However, as the temperature falls, the ESR of the output capacitor 5 increases, and thus, as shown in FIG. 13, the ripple voltage V_rmscontained in the output voltage V_Oincreases.

[0018] Here, as shown in FIG. 14, the gain G_AMPof the error amplifier 7 is fixed at 100 irrespective of the divided voltage V_adjfed to its inverting input terminal. Thus, the input-output characteristic of the error amplifier 7, i.e. the relationship between the divided voltage V_adjand the output voltage signal V_A, is as shown in FIG. 15. On the other hand, the input-output characteristic of the operational amplifier 9, i.e. the relationship between the output voltage signal V_Aand the duty factor “duty” of the PWM signal V_PWM, is as shown in FIG. 16.

[0019] In the switching power supply device having these characteristics, a malfunction in which the switching frequency f₀of the transistor 2 becomes equal to half the oscillation frequency of the oscillator 10 occurs when the ripple voltage V_adj1contained in the divided voltage V_adjfed to the error amplifier 7 and the duty factor “duty” has a relationship as shown in FIG. 17. In general, a switching power supply device provided with a step-down type DC—DC converter is used in the range of duty factors “duty” from 0% to 50%. Thus, as FIG. 17 clearly shows, the lower the duty factor “duty”, the lower the ripple voltage V_adj1at which the malfunction occurs. That is, the lower the duty factor “duty”, the more likely the malfunction occurs.

[0020] FIG. 18 shows a diagram in which the ripple voltage V_rmscontained in the output voltage V_Oshown in FIG. 13 is converted into the ripple voltage V_adj2contained in the divided voltage V_adj. It is to be noted that in FIG. 18 is indicated the ripple voltage V_10%, as calculated on the basis of FIG. 17, contained in the divided voltage V_adjwhen the malfunction occurs at a duty factor “duty” of 10%. As FIG. 18 clearly shows, at a duty factor “duty” of 10%, a malfunction in which the switching frequency f₀of the transistor 2 becomes equal to half the oscillation frequency of the oscillator 10 occurs at a temperature of −25° C. This is because, as shown in FIG. 19, the ripple voltage contained in the output voltage signal V_Afrom the error amplifier 7 becomes so high that the amplitude of the output voltage signal V_Abecomes greater than that of the output voltage V_OSCfrom the oscillator 10, with the result that the frequency of the PWM signal V_PWMbecomes equal to twice the frequency of the output voltage V_OSCfrom the oscillator 10.

[0021] The ripple voltage V_rmscontained in the output voltage V_Ois inversely proportional to the switching frequency f₀of the transistor 2, and therefore, when the switching frequency f₀of the transistor 2 becomes equal to half the oscillation frequency of the oscillator 10 as described above, the ripple voltage V_rmsbecomes twice as high, degrading the stability of the output voltage V_Oand increasing the ripple voltage contained therein.

[0022] An object of the present invention is to provide a switching power supply device of which the output voltage contains no higher ripple voltage even when the switching power supply device is used at low temperatures.

[0023] To achieve the above object, according to the present invention, a switching power supply device is provided with a DC—DC converter that receives an input voltage and outputs a varying voltage by varying the ratio of on periods to off periods of a switching transistor, an error amplifier that compares the output voltage of the DC—DC converter or a divided voltage thereof with a reference voltage and outputs an error voltage, an operator that produces a pulse signal according to the error voltage and controls the switching transistor with the pulse signal, and a gain control circuit that varies the gain of the error amplifier according to at least one of the duty factor of the pulse signal and the ambient temperature.

[0024] This and other objects and features of the present invention will become clear from the following description, taken in conjunction with the preferred embodiments with reference to the accompanying drawings in which:

[0025] FIG. 1 is a circuit block diagram of the switching power supply device of a first embodiment of the invention;

[0026] FIG. 2 is a circuit block diagram of the switching power supply device of a second embodiment of the invention;

[0027] FIG. 3 is a circuit block diagram of the switching power supply device of a third embodiment of the invention;

[0028] FIG. 4 is a graph showing the gain characteristic of the error amplifier provided in the switching power supply devices shown in FIGS. 1 and 2;

[0029] FIG. 5 is a graph showing the temperature characteristic of the ripple voltage contained in the divided voltage of the output voltage of the switching power supply devices shown in FIGS. 1 and 2;

[0030] FIG. 6 is a graph showing the gain characteristic of the error amplifier provided in the switching power supply device shown in FIG. 3;

[0031] FIG. 7 is a graph showing the temperature characteristic of the ripple voltage contained in the divided voltage of the output voltage of the switching power supply device shown in FIG. 3;

[0032] FIG. 8 is a diagram showing an example of the configuration of the gain control circuit provided in the switching power supply devices shown in FIGS. 1 to 3;

[0033] FIG. 9 is a diagram showing the configuration of the logarithmic amplifier;

[0034] FIG. 10 is a graph showing the input-output characteristic of the logarithmic amplifier shown in FIG. 9;

[0035] FIG. 11 is a circuit block diagram of a conventional switching power supply device;

[0036] FIG. 12 is a time chart showing the waveforms of relevant signals observed at room temperature in the switching power supply device shown in FIG. 11;

[0037] FIG. 13 is a graph showing the temperature characteristic of the ripple voltage contained in the output voltage of the switching power supply device shown in FIG. 11;

[0038] FIG. 14 is a graph showing the gain of the error amplifier provided in the switching power supply device shown in FIG. 11;

[0039] FIG. 15 is a graph showing the input-output characteristic of the error amplifier provided in the switching power supply device shown in FIG. 11;

[0040] FIG. 16 is a graph showing the input-output characteristic of the operational amplifier provided in the switching power supply device shown in FIG. 11;

[0041] FIG. 17 is a graph showing the relationship between the duty factor and the ripple voltage contained in the divided voltage of the output voltage when a malfunction occurs in the switching power supply device shown in FIG. 11;

[0042] FIG. 18 is a graph showing the temperature characteristic of the ripple voltage contained in the divided voltage of the output voltage of the switching power supply device shown in FIG. 11; and

[0043] FIG. 19 is a time chart showing the waveforms of relevant signals observed at low temperatures in the switching power supply device shown in FIG. 11.

[0044] First, the switching power supply device of a first embodiment of the invention will be described. FIG. 1 shows the switching power supply device of the first embodiment. In the following descriptions, such circuit elements and blocks as serve the same purposes as in FIG. 11 are identified with the same reference numerals or symbols, and their explanations will not be repeated. Here, the divided voltage V_adjof the output voltage V_Ofrom the output voltage detection circuit 6 is fed to a gain control circuit 12, and the gain control circuit 12, according to the divided voltage V_adj, varies the gain G_AMPof the error amplifier 7. Specifically, when the divided voltage V_adjis higher than 1.253 V, the gain control circuit 12 sets the gain G_AMPof the error amplifier 7 at 100 as in the conventional switching power supply device, and, when the divided voltage V_adjis equal to or lower than 1.253 V, the gain control circuit 12 sets the gain G_AMPof the error amplifier 7 at 50. Thus, since the divided voltage V_adjand the duty factor “duty” have a relationship as shown in FIGS. 15 and 16 described earlier, when the duty factor “duty” is higher than 20%, the gain G_AMPof the error amplifier 7 is set at 100, and, when the duty factor “duty” is equal to or lower than 20%, the gain G_AMPof the error amplifier 7 is set at 50, as shown in FIG. 4.

[0045] As a result of this operation, the ripple voltage contained in the divided voltage V_adjwhen a malfunction occurs at a duty factor “duty” of 10 % is now as indicated by V_10%′ in FIG. 5, i.e. twice as high as V_10%shown in FIG. 18. Thus, there no longer occurs a malfunction in which the switching frequency f₀of the transistor 2 becomes equal to half the oscillation frequency of the oscillator 10 even at a duty factor of 10% and at a low temperature (−25° C.).

[0046] Next, the switching power supply device of a second embodiment of the invention will be described. FIG. 2 shows the switching power supply device of the second embodiment. In the following descriptions, such circuit elements and blocks as serve the same purposes as in FIG. 11 are identified with the same reference numerals or symbols, and their explanations will not be repeated. Here, the input voltage V_INand the output voltage V_Oare fed to an input-output ratio detection circuit 13, which then feeds the input-output ratio V_IN/V_Oto the gain control circuit 12. The gain control circuit 12, according to the input-output ratio V_IN/V_O, varies the gain G_AMPof the error amplifier 7. Specifically, when the input-output ratio V_IN/V_Ois lower than 5, the gain control circuit 12 sets the gain G_AMPof the error amplifier 7 at 100 as in the conventional switching power supply device, and, when the input-output ratio V_IN/V_Ois equal to or higher than 5, the gain control circuit 12 sets the gain G_AMPof the error amplifier 7 at 50. Thus, since the input-output ratio V_IN/V_Oand the duty factor “duty” have a relationship as expressed by formula (1) described earlier, when the duty factor “duty” is higher than 20%, the gain G_AMPof the error amplifier 7 is set at 100, and, when the duty factor “duty” is equal to or lower than 20%, the gain G_AMPof the error amplifier 7 is set at 50, as shown in FIG. 4.

[0047] As a result of this operation, the ripple voltage contained in the divided voltage V_adjwhen a malfunction occurs at a duty factor “duty” of 10% is now as indicated by V_10%′ in FIG. 5, i.e. twice as high as V_10%shown in FIG. 18. Thus, there no longer occurs a malfunction in which the switching frequency f₀of the transistor 2 becomes equal to half the oscillation frequency of the oscillator 10 even at a duty factor of 10% and at a low temperature (−25° C.).

[0048] Now, an example of the configuration of the gain control circuit 12 provided in the switching power supply devices of the first and second embodiments will be described with reference to FIG. 8. The gain control circuit 12 is provided with resistors R3 to R5, an active device 15, and an active device control circuit 16. The inverting input terminal of the error amplifier 7 is connected through the resistor R4 to the output terminal of the error amplifier 7. The resistor R5 is connected, through the active device 15, in parallel with the resistor R4. Moreover, the resistor R3 is connected to the node between the resistor R4 and the inverting input terminal of the error amplifier 7. As the active device 15, it is possible to use a bipolar transistor, a MOS (metal-oxide semiconductor) transistor, or the like.

[0049] The active device control circuit 16 receives, in the case of the switching power supply device of the first embodiment, the divided voltage V_adjfrom the output voltage detection circuit 6 or, in the case of the switching power supply device of the second embodiment, the input-output ratio V_IN/V_Ofrom the input-output ratio detection circuit 13. The active device control circuit 16, according to the input signal thus fed thereto, controls the on/off state of the active device 15. When the active device 15 is in an on state, the gain G_AMPof the error amplifier 7 is given by formula (2) below, where R₃, R₄, and R₅represent the resistances of the resistors R3, R4, and R5, respectively. $\begin{array}{cc}{G}_{\mathrm{AMP}}=\frac{R_4\times R_5}{\left(R_4+R_5\right)\times R_3}& \left(2\right)\end{array}$

[0050] On the other hand, when the active device 15 is in an off state, the gain G_AMPof the error amplifier 7 equals R₄/R₃. Thus, by making the resistances of the resistors R4 and R5 equal, it is possible to obtain a gain characteristic as shown in FIG. 4.

[0051] The active device control circuit 16 may control the active device 15 not on an on/off basis but on a linear basis. By controlling the active device 15 linearly, it is possible to control the gain G_AMPof the error amplifier 7 linearly. Thus, by controlling the gain G_AMPof the error amplifier 7 so as to fulfill formula (6) or (8) described later, it is possible to make the total gain of the feedback system of the switching power supply device constant and thereby further enhance the stability thereof.

[0052] Now, to formulate the aforementioned condition (expressed by formula (6) or (8)) that makes the total gain of the feedback system of the switching power supply device constant, the amount of feedback ΔV_adj′ with respect to the variation ΔV_adjin the divided voltage will be determined. First, on the basis of formula (1), the output voltage V_Ois given by

[0053] V_O=V_IN×t_ON/T

[0054] where T represents the period of the PWM signal V_PWM. Hence, the variation ΔV_Oin the output voltage with respect to the variation ΔV_adjin the divided voltage is given by formula (3) below. $\begin{array}{cc}\begin{array}{c}\frac{\Delta V_O}{\Delta V_{\mathrm{adj}}}=\frac{\Delta V_O}{\Delta V_A}\times \frac{\Delta V_A}{\Delta V_{\mathrm{adj}}}\\ =\frac{V_{\mathrm{IN}}}{T}\times \frac{\Delta t_{\mathrm{ON}}}{\Delta V_A}\times \frac{\Delta V_A}{\Delta V_{\mathrm{adj}}}\end{array}& \left(3\right)\end{array}$

[0055] On the other hand, the amount of feedback ΔV_adj′ with respect to the variation ΔV_adjin the divided voltage is given by formula (4) below. $\begin{array}{cc}\frac{\Delta V_{\mathrm{adj}}^{\prime }}{\Delta V_{\mathrm{adj}}}=\frac{\Delta V_O\times \frac{V_{\mathrm{ref}}}{V_O}}{\Delta V_{\mathrm{adj}}}& \left(4\right)\end{array}$

[0056] With formulae (3) and (4) combined, the amount of feedback ΔV_adj′ with respect to the variation ΔV_adjin the divided voltage is given by formula (5) below. $\begin{array}{cc}\begin{array}{c}\frac{\Delta V_{\mathrm{adj}}^{\prime }}{\Delta V_{\mathrm{adj}}}=\frac{V_{\mathrm{IN}}}{T}\times \frac{\Delta t_{\mathrm{ON}}}{\Delta V_A}\times \frac{\Delta V_A}{\Delta V_{\mathrm{adj}}}\times \frac{V_{\mathrm{ref}}}{V_O}\\ =\frac{V_{\mathrm{IN}}}{V_O}\times \frac{1}{T}\times \frac{\Delta t_{\mathrm{ON}}}{\Delta V_A}\times G_{\mathrm{AMP}}\times V_{\mathrm{ref}}\end{array}& \left(5\right)\end{array}$

[0057] The total gain of the feedback system of the switching power supply device can be made constant by making constant the amount of feedback ΔV_adj′ with respect to the variation ΔV_adjin the divided voltage. Specifically, in formula (5), the terms 1/T, Δt_ON/ΔV_A, and V_refare all constant, and therefore, by controlling the active device 15 linearly so as to fulfill formula (6) below, it is possible to make the total gain of the feedback system of the switching power supply device constant. Here, K represents a constant. This control can be applied to the switching power supply device of the second embodiment. $\begin{array}{cc}\frac{V_{\mathrm{IN}}}{V_O}\times G_{\mathrm{AMP}}=K& \left(6\right)\end{array}$

[0058] Using the divided voltage V_adj, the term V_IN/V_Ocan be expressed as formula (7) below. $\begin{array}{cc}\frac{V_{\mathrm{IN}}}{V_O}=0.5-G_{\mathrm{AMP}}\times \left(V_{\mathrm{adj}}-V_{\mathrm{ref}}\right)\times \frac{\Delta \mathrm{duty}}{\Delta V_A}& \left(7\right)\end{array}$

[0059] Hence, formula (6) can be expressed as formula (8) below. By controlling the active device 15 linearly so as to fulfill formula (8), it is possible to make the total gain of the feedback system of the switching power supply device constant. This control can be applied to the switching power supply device of the first embodiment. $\begin{array}{cc}\left[0.5-G_{\mathrm{AMP}}\times \left(V_{\mathrm{adj}}-V_{\mathrm{ref}}\right)\times \frac{\Delta \mathrm{duty}}{\Delta V_A}\right]\times G_{\mathrm{AMP}}=K& \left(8\right)\end{array}$

[0060] Next, the switching power supply device of a third embodiment of the invention will be described. FIG. 3 shows the switching power supply device of the third embodiment. In the following descriptions, such circuit elements and blocks as serve the same purposes as in FIG. 11 are identified with the same reference numerals or symbols, and their explanations will not be repeated. Here, a temperature detection circuit 14 detects the ambient temperature, and feeds the resulting detection signal to the gain control circuit 12. The gain control circuit 12, according to the detection signal fed from the temperature detection circuit 14, varies the gain G_AMPof the error amplifier 7. Specifically, when the temperature T_adetected by the temperature detection circuit 14 is higher than 0° C., the gain control circuit 12 sets the gain G_AMPof the error amplifier 7 at 100 as in the conventional switching power supply device, and, when the temperature T_adetected by the temperature detection circuit 14 is equal to or lower than 0° C., the gain control circuit 12 sets the gain G_AMPof the error amplifier 7 at 50, as shown in FIG. 6.

[0061] As a result of this operation, the ripple voltage contained in the divided voltage V_adjwhen a malfunction occurs at a duty factor “duty” of 10% is now, in the range of temperatures of 0° C. and below, as indicated by V_10%″ in FIG. 7, i.e. twice as high as V_10%shown in FIG. 18. Thus, there no longer occurs a malfunction in which the switching frequency f₀of the transistor 2 becomes equal to half the oscillation frequency of the oscillator 10 even at a duty factor of 10% and at a low temperature (−25° C.).

[0062] In the switching power supply device of the first or second embodiment, it is possible to additionally provide a temperature detection circuit and vary the gain G_AMPof the error amplifier 7 according to the ambient temperature.

[0063] For example, in the switching power supply device of the first embodiment, the gain G_AMPof the error amplifier 7 is made lower when the ambient temperature is equal to or lower than 0° C. than when the ambient temperature is higher than 0° C. Moreover, the gain G_AMPof the error amplifier 7 is made lower when the ambient temperature is higher than 0° C. and in addition the divided voltage V_adjis equal to or lower than 1.253 V than when the ambient temperature is higher than 0° C. and in addition the divided voltage V_adjis higher than 1.253 V.

[0064] On the other hand, in the switching power supply device of the second embodiment, the gain G_AMPof the error amplifier 7 is made lower when the ambient temperature is equal to or lower than 0° C. than when the ambient temperature is higher than 0° C. Moreover, the gain G_AMPof the error amplifier 7 is made lower when the ambient temperature is higher than 0° C. and in addition the input-output ratio V_IN/V_Ois equal to or higher than 5 than when the ambient temperature is higher than 0° C. and in addition the input-output ratio V_IN/V_Ois lower than 5.

[0065] These configurations prevent the gain G_AMPof the error amplifier 7 from being reduced too much in normal operation in the range of temperatures from 25 to 80° C. This helps ensure the stability of the output voltage V_Oagainst variation in the input voltage V_IN.

[0066] In the first embodiment, as the error amplifier 7 provided with the gain control circuit 12, it is also possible to use a logarithmic amplifier 17 as shown in FIG. 9. The logarithmic amplifier 17 is provided with an error amplifier 7, a diode 18, a resistor R3, and a level shift amplifier 19. The error amplifier 7 has its inverting input terminal connected to the anode of the diode 18, and has its output terminal connected to the cathode of the diode 18. The node between the diode 18 and the inverting input terminal of the error amplifier 7 is connected to the resistor R3, and the node between the diode 18 and the output terminal of the error amplifier 7 is connected to the level shift amplifier 19. The level shift amplifier 19 adds a predetermined value to the signal fed thereto and then outputs the resulting signal as the output voltage signal V_A. Here, the divided voltage V_adjand the output voltage signal V_Ahave a relationship as shown in FIG. 10. Specifically, the higher the divided voltage V_adj, the lower the gain G_AMP. This makes it possible to realize the switching power supply device of the first embodiment with a cheaper and simpler configuration than that shown in FIG. 8. However, in this configuration, it is not possible to perform control so as to fulfill formula (8), and therefore it is not possible to make the total gain of the feedback system of the switching power supply device constant.

[0067] In the switching power supply devices of the first to third embodiments, a step-down type DC—DC converter is used as the DC—DC converter. However, the DC—DC converter used in the present invention is not limited to this specific type, but may be of any other type, such as a step-up or step-down/step-up type. For example, in general, a switching power supply device employing a step-up type DC—DC converter is used in the range of duty factors “duty” from 50% to 100%, and, the higher the duty factor “duty”, the lower the ripple voltage V_adj1at which a malfunction occurs. Therefore, it is advisable to lower the gain of the error amplifier at least when the duty factor of the PWM signal supplied to the switching transistor is high or when the ambient temperature is low.

[0068] In the switching power supply devices of the first to third embodiments, the error amplifier 7 is connected through the output voltage detection circuit 6 to the output terminal OUT so that the error amplifier 7 receives the divided voltage V_adjof the output voltage V_O. However, the present invention is not limited to this specific configuration, but may be so implemented that the error amplifier 7 is connected directly to the output terminal OUT so that the error amplifier 7 receives the output voltage V_O. In this case, it is necessary to set the reference voltage V_refoutput from the reference voltage source 8 higher than in the switching power supply devices of the first to third embodiments.