CMOS voltage controlled phase shift oscillator

[0002] The present invention relates generally to integrated circuits. In particular, the invention relates to CMOS voltage controlled phase shift oscillators.

[0003] Two types of oscillators are commonly used in CMOS integrated circuits to generate high frequency signals. A first type includes L-C oscillators. L-C oscillators with spiral inductors are used as voltage controlled oscillators (VCO's) in RF and wireless communication circuits. An example of this type of VCO's is described in an article by L. M. Burns, entitled, “Applications for GaAs and Silicon Integrated Circuits in next generation wireless communication systems,” IEEE J. Solid-State Circuits, Vol. 30, No. 10, pp. 1088-1095 October 1995. A second type of oscillators includes ring oscillators. Ring oscillators can be used to generate high frequency clocking voltages or signals in digital integrated circuits. An example of this type of oscillator is described in an article by L. Sun et al., entitled “Quadrature output voltage controlled ring oscillator based on three-stage sub-feedback loops,”IEEE Int. Symp. On Circuits and Systems, Orlando Fla., vol. II, pp. 176-170, May 1999. VCO's are also required in phase locked loops employed in communication systems and digital integrated circuits for signal and clock recovery. The use of VCO's in these types of applications are described in an article by W. Rhee, “Design of low-jitter 1-GHz phase-locked loops for digital clock generation,” IEEE Int. Symp. On Circuits and Systems, Orlando, Fla., vol.II, pp. 520-523, May 1999.

[0004] FIG. 1 shows a commonly employed CMOS VCO using resonant L-C circuits with spiral inductors (see J. Craninckx and M. Steyaert, “A 1.8 GHz low-phase-noise spiral-LC CMOS VCO,” Symp. on VLSI Circuits, 1996, pp. 30-31). One difficulty with this approach has been the fabrication of high Q inductors on the lossy silicon substrates. One solution to the above approach includes techniques for fabricating high Q inductors and VCO's without inductors or high Q active inductors using transistors to compensate for the losses in inductors or replace inductors are disclosed in copending applications by the same inventor (see P. Farrar and L. Forbes, “HIGH Q AIRBRIDGE INDUCTORS FOR SILICON CMOS RF INTEGRATED CIRCUITS,” U.S. Pat. No. 6,025,261, 15.

[0005] The second commonly used technique to generate high frequency signals in CMOS integrated circuits is a ring oscillator as shown in FIG. 2A (see L. Sun et al., “Quadrature output voltage controlled ring oscillator based on three-stage sub-feedback loops,” IEEE Int. Symp. On Circuits and Systems, Orlando, Fla., vol. II, pp. 176-170, May 1999). A ring oscillator uses an odd number of inverters in an unstable ring configuration which depends of the signal delay through each stage to generate a signal which returns to the input inverted to the original input but at a later time. A larger number of stages results in a lower frequency of oscillation as shown in FIG. 2B, where fo=1/(2 n tprop), where tprop is the propagation delay of the inverter stages used in the ring.

[0006] Phase shift oscillators are based on a 180 phase shift through a passive R-C network to generate a signal which appears back at the input of an inverting amplifier in phase with the original signal a shown in FIG. 3A. CMOS phase shift oscillators have been previously described. For a further description on the same, see M. Caughey et al., “Circuit simulation by computer,” Telesis (Bell-Northern Research), Vol. 2, no. 6, pp. 17-24, 1972; and G. W. Short, “CMOS phase shift oscillators,” New Electronics, vol. 13, no. 5, p. 64, Mar. 4, 1980. While passive tapered networks work well for low frequency oscillators they constitute a problem for high frequency oscillators in COOS technology. As shown in FIG. 3A, the resistances and impedance levels get successively large from one stage to the next so that the stages do not load the previous stage. Each can be treated as a separate stage each with 60 degrees phase shift, as shown in FIG. 3B. FIG. 3B is a phase shift diagram for a given stage in FIG. 3A which plots the imaginary phase component (Im) verses the real phase component (Re). As represented in FIG. 3A, the last capacitance in the sequence will be by necessity small, e.g. C/100, or have a large impedance, the input capacitance of the amplifier can and will be comparable loading the last stage and upsetting the correct phase relationships. FIG. 3C is a gain-phase shift diagram illustrating the gain and phase shift per stage in the conventional phase shift oscillator of FIG. 3A. FIG. 3D illustrates a circuit simulation expressed as voltage versus time for the conventional phase shift oscillator of FIG. 3A. With three passive stages, as illustrated in FIG. 3A, the gain of the amplifier must be about ten or larger so that the loop gain will be larger than one. Each R-C stage reduces the gain by about a factor of two. CMOS amplifiers typically have only limited gain bandwidth products and a gain of ten implies a low bandwidth and low frequency operation and oscillations.

[0007] Therefore, there is a need in the art to provide improved phase shift oscillators, or phase shift networks in which the oscillation frequency can be increased by increasing the number of stages and where each stage has a small gain and phase shift. It is also desirable to be able to control the phase shift by an externally applied voltage.

[0008] The above mentioned problems with oscillator circuits and other problems are addressed by the present invention and will be understood by reading and studying the following specification. Systems and methods are provided for CMOS voltage controlled phase shift oscillators.

[0009] In one embodiment of the present invention, a phase shift circuit is provided having any odd number of stages coupled in series between a circuit input and a circuit output. Each stage includes a CMOS amplifier coupled to the input. A phase shift network is coupled to the CMOS amplifier. The CMOS amplifier provides a gain and allows a small phase shift in each stage to eventually provide a signal which is 180 degrees out of phase with the input signal. In the CMOS amplifier, the PMOS transistor is a diode connected PMOS transistor which acts as a low valued load resistance. In the phase shift network, an NMOS transistor is used as a voltage variable resistor for providing a resistance value in the circuit. A source region of the phase shift network NMOS transistor is coupled to a gate bias supply, VGG, and a drain region of the phase shift network NMOS transistor is coupled to a gate of the NMOS transistor in the CMOS amplifier to provide a gate bias to the NMOS transistor in the CMOS amplifier.

[0010] These and other embodiments, aspects, advantages, and features of the present invention will be set forth in part in the description which follows, and in part will become apparent to those skilled in the art by reference to the following description of the invention and referenced drawings or by practice of the invention. The aspects, advantages, and features of the invention are realized and attained by means of the instrumentalities, procedures, and combinations particularly pointed out in the appended claims.

[0011] FIG. 1 shows a commonly employed CMOS VCO using resonant L-C circuits with spiral inductors.

[0012] FIG. 2A illustrates a conventional ring oscillator as used to generate high frequency signals in CMOS integrated circuits.

[0013] FIG. 2B illustrates the manner in which a larger number of stages results in a lower frequency of oscillation, where fo=1/(2 n tprop) and where tprop is the propagation delay of the inverter stages used in the ring oscillator of FIG. 2A.

[0014] FIG. 3A illustrates a conventional phase shift oscillator which is based on a 180 phase shift through a passive R-C network to generate a signal which appears back at the input of an inverting amplifier in phase with the original signal.

[0015] FIG. 3B, illustrates that the resistances and impedance levels get successively large from one stage to the next in the circuit of FIG. 3A with the effect that the stages do not load the previous stage and each can be treated as a separate stage each with 60 degrees phase shift.

[0016] FIG. 3C is a gain-phase shift diagram illustrating the gain and phase shift per stage in the conventional phase shift oscillator of FIG. 3A.

[0017] FIG. 3D illustrates a circuit simulation expressed as voltage versus time for the conventional phase shift oscillator of FIG. 3A.

[0018] FIG. 4A illustrates an embodiment of a CMOS phase shift oscillator, phase shift network, or phase shift circuit according to the teachings of the present invention.

[0019] FIG. 4B illustrates in more detail the make up of a stage of the phase shift circuit of FIG. 4A according to the teachings of the present invention.

[0020] FIG. 4C is another representation of the stage shown in FIG. 4B in which the diode connected PMOS transistor is schematically illustrated.

[0021] FIG. 4D is a gain-phase shift diagram illustrating the gain and phase shift per stage in an example embodiment of a five stage embodiment for the circuit shown in FIGS. 4A-4C.

[0022] FIG. 4E illustrates a circuit simulation expressed as voltage versus time for the example embodiment of a five stage circuit shown in FIGS. 4A-4C.

[0023] FIG. 5 is a graph of the oscillation frequency (fo) versus the number of stages according to the teachings of the present invention.

[0024] FIG. 6A is a schematic illustration representative of any given stage, according to the teachings of the present invention, taking account of second order effects.

[0025] FIG. 6B is a gain-phase shift diagram illustrating the gain and phase shift per stage, according to the teachings of the present invention, taking into account second order effects.

[0026] FIG. 7A provides another embodiment of the invention, illustrating a circuit having seven stages in which three stages have been designed, or configured, such as by adjustment of the resistor value R, provided by the NMOS transistor, or voltage variable resistor, in the phase shift network to have a first degree phase shift and another four stages have been configured by like methods to have a second degree phase shift.

[0027] FIG. 7B illustrates that, according to the circuit embodiments of the present invention, multiples of two resistors in a given circuit can be varied, one becoming larger increasing the phase shift on one stage and one resistor made small decreasing the phase shift on another stage but leaving the overall phase shift and oscillation frequency of the novel circuit fixed.

[0028] In the following detailed description of the invention, reference is made to the accompanying drawings which form a part hereof, and in which is shown, by way of illustration, specific embodiments in which the invention may be practiced. The embodiments are intended to describe aspects of the invention in sufficient detail to enable those skilled in the art to practice the invention. Other embodiments may be utilized and changes may be made without departing from the scope of the present invention. The following detailed description is not to be taken in a limiting sense, and the scope of the present invention is defined only by the appended claims.

[0029] We here describe CMOS phase shift oscillators which employ a CMOS amplifier and phase shift network in each stage. These do not utilize L-C circuits nor resonant circuit concepts, and they do not utilize the signal delay concepts employed in ring oscillators. They utilize the phase shift in successive stages to generate an unstable feedback circuit. Here, however, unlike in simple phase shift oscillators a transistor amplifier and phase shift network is used in each stage. Using an amplifier in each stage provides gain in each stage and allows a small phase shift in each stage to eventually provide a signal which is 180 degrees out of phase with the input signal. Unlike simple CMOS phase shift oscillators with three passive R-C phase shift networks here there can be any odd number of amplifier and phase shift networks connected in series.

[0030] FIG. 4A illustrates an embodiment of a CMOS phase shift oscillator, phase shift network, or phase shift circuit 400 according to the teachings of the present invention. As shown in FIG. 4A, the phase shift circuit includes a number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N. The phase shift circuit includes an input 402 for receiving an input signal and an output 403. As shown in FIG. 4A, a feedback line 405 is provided which couples the output 403 to the input 402. In the embodiment shown in FIG. 4A only five stages are shown. As one of ordinary skill in the art will understand upon reading this disclosure, the invention is not limited to any fixed number of stages. This will further be understood from the description provided below. By way of example, this circuit 400 is understood to oscillate at 1.0 Ghz. According to the teachings of the present invention, the circuit 400 depends on a phase shift in each stage to contribute to the required total phase shift. Also, according to the teachings of the present invention, a larger number of stages requires a smaller phase shift in each stage which is realized at a higher frequency, consequently a larger number of stage results in a higher oscillation frequency. By contrast, In the ring oscillators, described above, a larger number of stages results in a lower frequency.

[0031] As one of ordinary skill in the art will understand upon reading this disclosure, for oscillation the series or number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, between the input 402 and the output 403 of circuit 400 must produce a phase shift of 180 degrees. Thus, in the embodiment shown in FIG. 4A, each stage, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, has a phase shift of 180/5=36 degrees.

[0032] FIG. 4B illustrates in more detail the make up of stage 401-1. Stage 401-1 is a representative sample of any one of the stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, in circuit 400 according to the teachings of the present invention. According to the teachings of the present invention, the oscillator circuit 400 includes any odd number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, connected in series. As shown in FIG. 4B, each stage, e.g. 401-1, has an input (Vin) and an output (Vo). Each stage further includes an amplifier 407 and a phase shift network 409 coupled to the amplifier 407. According to the teachings of the present invention, the amplifier 407 includes a CMOS amplifier 407. As one of ordinary skill in the art will understand from reading this disclosure, the CMOS amplifier provides a gain and allows a small phase shift in each stage, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, to eventually provide an output signal at the output 403 of circuit 400 which is 180 degrees out of phase with an input signal received at the input 402 of the circuit 400. As shown in FIG. 4B, the amplifier 407 includes an NMOS transistor 411 and a diode connected PMOS transistor 413 acting as a load transistor. In FIG. 4B the diode connected PMOS transistor 413 is illustrated having a source region 415 coupled to a power supply (VDD) and a drain region 417 coupled to the output (Vo) of the stage 401-1, a drain region 419 of the NMOS transistor 411, and to a gate 421 of the diode connected PMOS transistor 413. The input (Vin) of the stage 401-1 is coupled to a gate 423 of the NMOS transistor 411. Vin is similarly coupled to the phase shift network 409 of the stage 401-1. According to the teachings of the present invention, the phase shift network 409 includes an NMOS transistor 425 which can be designed to have a variable resistance, R. The drain region 427 for the phase shift network 409 NMOS transistor 425 is coupled to the gate 423 of the NMOS transistor in the amplifier 407. The source region 429 for the phase shift network 409 NMOS transistor 425 is connected to a gate bias supply (VGG). A source region for the NMOS transistor 411 is coupled to a low potential, e.g. ground.

[0033] FIG. 4C is another representation of the stage 401-1 shown in FIG. 4B. In FIG. 4C, the diode connected PMOS transistor 413 is schematically illustrated as a diode 413. According to the teachings of the present invention, an amplifier 407 input capacitance is much smaller than the capacitance, C, in the phase shift network 409. Likewise, the phase shift network resistance, R, is much larger than the load resistor, formed by the diode connected PMOS transistor 413, of the amplifier 407. The load resistance for the amplifier can be expressed as RL=1/gd, where gd is the conductance looking back into the drain 417 of the diode connected PMOS transistor 413. A diode connected PMOS transistor 413 has a relatively low impedance, or acts as a relatively low valued load resistance, e.g. RL=1/gd. This allows an analysis of the circuit 400 which can be regarded as being composed of the phase shift network resistance, R, and the capacitance, C, in the phase shift network 409. In other words, in contrast to the conventionally employed CMOS VCO circuit using resonant L-C circuits with spiral inductors the circuit 400 of the present invention provides an output signal without the use of any L-C circuits. The circuit 400 thus provides an output signal without utilizing signal delay concepts employed by conventional ring oscillators. Only a low valued load resistance is required since the gain of the amplifier 407 needs only to be slightly greater than one according to the teachings of the present invention. In one embodiment, the overall gain of the stage 401-1, including amplifier 407 plus phase shift network 409, can be expressed as about, 0.9 gm RL, where (gm) is the transconductance and RL is the load resistance of the amplifier 407. Thus, as one of ordinary skill in the art will understand according to the teachings of the present invention, if the product (gm RL) is slightly larger than one the overall gain will be greater than one and the condition for oscillation is satisfied. For oscillation the series of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, must produce a phase shift of 180 degrees and a gain greater than one. The frequency of oscillation for the circuit 400 using this analysis can be expressed as, fo=n/(2 pi×pi×R×C) Hz, where n=number of stages and pi=3.14 radians. As one of ordinary skill in the art will understand from reading this disclosure, a larger number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, will result in a higher frequency of oscillation for the circuit 400 and requires a lower gain from the amplifier 407 per stage, 401-1, 401-2, 401-3, 401-4, . . . , 401-N.

[0034] FIG. 4D is a gain-phase shift diagram illustrating the gain and phase shift per stage, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, for the five stage embodiment shown in FIGS. 4A-4C. As shown in FIG. 4D, the gain (H) for any given stage can be expressed as H(s)=R/(R+(1/sC)=sRC/(1+sRC)=1/(1+(1/sRC)) where s=j ω. The corner frequency ωc attributable to phase shift network can be express as ωc =1/RC. And, the angular frequency at the input to the amplifier 407 is shown as ωo. Again, for the embodiment shown in FIG. 4A, having five stages, each phase shift network is shown having a phase shift of approximately 36 degrees and an overall gain, including amplifier 407 plus phase shift network 409, expressed as about, 0.9 gm RL, here (gm) is the transconductance and RL is the load resistance of the amplifier 407. If the product (gm RL) is slightly larger than one the overall gain will be greater than one and the condition for oscillation is satisfied.

[0035] FIG. 4E gives a circuit simulation expressed as voltage versus time for the example five stage circuit embodiment shown in FIG. 4A. The embodiments shown in FIGS. 4A-4E demonstrate operation of the phase shift oscillator circuit 400. The novel circuit 400 is achieved using nominal CMOS transistors but ones which have finite input capacitances, Cin, and utilize an amplifier stage 407 with a low gain, wide bandwidth and low output impedance. The output impedance (Zo) of the amplifier is Zo=1/gd. Since the gain requirement is low the load resistance RL can be small and the output impedance low. A low output impedance drives the phase shift network of the next stage without significant loading effects. According to the teachings of the present invention, tuning of the circuit 400, or varying the frequency (fo) is accomplished by using the NMOS transistor 425 as a voltage variable resistor for the phase shift network resistance, R, in the phase shift network 409. As one of ordinary skill in the art will understand upon reading this disclosure, changing the value of R changes the oscillation frequency, fo, in the above shown oscillation frequency formula. As long as the overall gain (H) remains greater than one as the phase shift network resistance, R, varies the circuit will still oscillate but now at a new frequency. The phase shift network resistance, R, provided by the NMOS transistor 425, e.g. voltage variable resistor, also serves to provide gate bias to the gate 423 of the NMOS transistor 411 in the amplifier circuit 407. As stated above a source region 429 for the NMOS transistor 425 in the phase shift network 409 is connected to a gate bias supply, VGG.

[0036] The above Figures further demonstrate how the phase shift and gain (H) vary with R and C. Larger values of the phase shift network resistance, R, and/or the capacitance, C, in the phase shift network 409 result in a smaller phase shift per stage at any particular frequency, e.g. ωo=1/RC. Again, the number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, can be any odd number of stages. If on the other hand, the required phase shift is fixed by the number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, then larger values of the phase shift network resistance, R, will lower the frequencies of oscillation. If the number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, is fixed then the frequency (fo) can be varied by varying the phase shift network resistance, R, provided by the NMOS transistor 425, e.g. voltage variable resistor, in the phase shift network 409, for all stages or only some stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N.

[0037] FIG. 5 is a graph of the oscillation frequency (fo) versus the number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, according to the teachings of the present invention. FIG. 5 shows the results of circuit simulations for different numbers of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, and different values of the phase shift capacitance, C, of the phase shift network 409. As shown in FIG. 5, the oscillation frequency increases with the number of stages and increases as the R-C time constant decreases or the value of phase shift capacitance, C, of the phase shift network 409 decreases. Thus, in the embodiment of five stages, as shown in connection with FIGS. 4A-4E, a low phase shift capacitance, C=20 femtoFarads (fF), of the phase shift network 409 produces an oscillation frequency (fo) of approximately 1.0 Ghz.

[0038] In a practical oscillator circuit it may be necessary and/or desirable to incorporate the effects of a finite value of input capacitance, Cin, of the amplifier 407, which in the above example has been taken to be much smaller that the capacitance, C, in the phase shift network, 409 from FIG. 4B. Similarly it may be necessary and/or desirable to incorporate the fact that the phase shift network resistance, R, provided by the NMOS transistor 425, e.g. voltage variable resistor, in the phase shift network 409, may not be many times larger than load resistor provided by the diode connected PMOS transistor 413 in the amplifier 407. In the discussion connected with FIGS. 4A-4E it was assumed that the phase shift network resistance, R, was much larger than the load resistor, RL, of the amplifier 407. Accordingly, these effects have been included in circuit simulations, and while the simple formula for the oscillation frequency, fo, predicts the correct trend, the actual oscillation frequencies may be different than those given by the simple formula fo=n/(2 pi×pi×R×C) Hz.

[0039] FIGS. 6A-6B shows a more detailed analysis including these second order effects. FIG. 6A is a schematic illustration representative of any given stage, e.g. 401-1, in the novel circuit 400 taking account of these second order effects, e.g. the finite input capacitances (Cin) of the amplifier 407. FIG. 6B is a gain-phase shift diagram illustrating the gain and phase shift per stage, 401-1, 401-2, 401-3, 401-4, . . . , 401-N taking into account these second order effects. As one of ordinary skill in the art will understand upon reading this disclosure and viewing FIG. 6B, the effect of Cin, or input capacitance, of the amplifier 407 is to reduce the oscillation frequency (fo). The effect of a limited value for the phase shift network resistance, R, provided by the NMOS transistor 425, e.g. voltage variable resistor, in the phase shift network 409, which may not be many times larger than load resistor (RL) provided by the diode connected PMOS transistor 413 in the amplifier 407, is to require an increased gain (H) for the amplifier 407.

[0040] As stated above, any odd number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, can be utilized in the phase shift circuit 400. It should further be noted that while the total phase shift has to be 180 degrees, each stage, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, does not have to have the same phase shift. In other words, the 180 degree phase shift can be divided up between the amplifier 407 and phase shift networks 409 for different ones of the number of stages, 401-1, 401-2, 401-3, 401-4, . . . , 401-N, in unequal portions.

[0041] To further illustrate this point, FIG. 7A provides another embodiment of the invention. FIG. 7A illustrates a circuit 700 having seven stages formed according to the teachings of the present invention. In the embodiment of FIG. 7A, three stages, 701-5, 701-6, and 701-7, have been designed, or configured, such as by adjustment of the phase shift network resistance, R, provided by the NMOS transistor 425, e.g. voltage variable resistor, in the phase shift network 409, to have a 30 degree phase shift. In FIG. 7A, another four stages, 701-1, 701-2, 701-3, and 701-4, have been configured by like methods to have 22.5 degree phase shifts. As one of ordinary skill in the art will understand upon reading this disclosure, this embodiment of the invention will allow two quadrature signals to be obtained or two signals at the same frequency but with a 90 degree phase relationship. Such quadrature signals may be useful in signal processing circuits and phase-locked loops.

[0042] One of ordinary skill in the art will further understand upon reading this disclosure and referring to FIG. 7B, that according to the present invention two phase shift network resistances, e.g. R1 and R2, provided by the NMOS transistors 425 in a given circuit 700 can be varied, one becoming larger increasing the phase shift on one stage and one resistor made small decreasing the phase shift on another stage but leaving the overall phase shift and oscillation frequency fixed. Again, here the reference is being made to the phase shift network resistance, R, provided by the NMOS transistor 425, e.g. voltage variable resistor, in the phase shift network 409 described in connection with FIG. 4B. To illustrate, in the embodiment shown in FIG. 7B two resistors, R1 and R2, in a given circuit 700 are varied, one becoming larger (R2) increasing the phase shift on one stage, e.g. 701-4 and one resistor (RI) made small decreasing the phase shift on another stage, e.g. 701-2, but leaving the overall phase shift and oscillation frequency fixed. In such a manner phase modulated signals can be produced for communication systems.

[0043] We have described here a CMOS voltage controlled phase shift oscillator for use in communication and digital integrated circuits, timing circuits, and signal and clock recovery circuits. The CMOS phase shift voltage controlled ring oscillator, or phase shift circuit, includes any odd number of stages coupled in series. Each stage includes a CMOS amplifier. A phase shift network is coupled to the CMOS amplifier. The CMOS amplifier provides a gain and allows a small phase shift in each stage to eventually provide a signal which is 180 degrees out of phase with the input signal. In the CMOS amplifier, the PMOS transistor is a diode connected PMOS transistor which acts as a low valued load resistance. In the phase shift network, an NMOS transistor is used as a voltage variable resistor for providing a resistance value in the circuit. The oscillation frequency can be increased, unlike a conventional inverter ring oscillator, by increasing the number of stages. Further, unlike a conventional phase shift oscillator, each stage has a small gain and phase shift. The ability to control the phase shift by an externally applied voltage through the NMOS transistor serving as a voltage variable resistor provides a wide range of potential applications.