Linear amplifier using frequency 2nd order intermodulation feed-forwarding method

[0002] 1. Field of the Invention

[0003] The present invention relates to a linear amplifier, and more particularly, to a linear amplifier using a low frequency 2^ndorder intermodulation feed-forwarding method.

[0004] 2. Description of the Related Art

[0005] When a mobile terminal or a mobile base station power-amplifies and transmits a signal, the receiving sensitivity of a receiving side is degraded by distortion due to the interference between a fundamental signal to be transmitted and its intermodulation component.

[0006] In order to solve the above problem, conventional linear amplifiers shown in FIGS. 1A, 1B and 2 are used. Each of the conventional linear amplifiers roughly includes an input impedance matching unit (Z_s) 20 for matching the output impedance of a signal source Vs with the input impedance of the input port of a transistor 12, 14 or 24, and an output impedance matching unit (Z_L) 29 impedance-matching the output port of the transistor 12, 14 or 24 with an output load. An amplifier shown in FIG. 1A tunes to a distortion component signal using a source harmonic tuning method and reflects an intermodulation signal reflected by the gate of the transistor 12 back to the gate using a reflection coefficient control unit 17 having a variable capacitor or a stub at the gate port of the transistor 12, thereby offsetting a distortion component. In FIG. 1B, a reflection coefficient control unit 16 is installed to the rear of the transistor 14, and tunes a harmonic output from the rear of the transistor 14 and reflects the harmonic back to the gate using a load harmonic tuning method, thereby offsetting a 3^rdorder harmonic component.

[0007] FIG. 2 shows a linear amplifier using an intermodulation signal feed-backwarding method. The linear amplifier of FIG. 2 includes an amplification transistor 24, filters 25 and 28 for filtering a 2^ndorder harmonic component from an input signal, a phase shifting unit 26 for shifting the phase of the 2^ndorder harmonic component, and an amplifier 27 for amplifying the phase-shifted 2^ndorder harmonic component to a predetermined value. The phase-shifted 2^ndorder intermodulation signal is fed backward to the input port of the amplification transistor 24, thereby offsetting a 3^rdorder intermodulation component.

[0008] However, these conventional intermodulation signal tuning methods can offset only several dB of 3^rdorder harmonic component. In the intermodulation signal feed-backwarding method, 10 dB to 15 dB of intermodulation component can be offset, but the characteristics is unstable and the band is narrowed.

[0009] An objective of the present invention is to provide a linear amplifier using a low frequency 2^ndorder intermodulation feed-forwarding method for offsetting a 3^rdintermodulation component while maintaining the stable characteristics of a linear amplification circuit obtained by feed-forwarding.

[0010] To achieve the above objective, the present invention provides a linear amplifier using a low frequency 2^ndintermodulation feed-forwarding method to offset the 3^rdintermodulation component of a fundamental signal, the device including: a transistor for amplifying the power of the fundamental signal; an input impedance matching unit connected to the gate of the transistor for impedance-matching a source of the received fundamental signal with the gate of the transistor; a first filter unit, one end of which is connected to the gate of the transistor, for filtering a 2^ndorder intermodulation signal from the fundamental signal; a low frequency amplifier connected to the other end of the first filter unit, for amplifying the 2^ndorder intermodulation signal with a predetermined voltage gain; a second filter unit connected in series between the output port of the low frequency amplifier and the drain output port of the transistor, for filtering the amplified 2^ndintermodulation signal and feeding the same to the output port of the transistor; and an output impedance matching unit connected to the output port of the transistor for impedance-matching the output port of the transistor with an external load.

[0011] The voltage gain (β) of the low frequency amplifier $-\frac{{\mathrm{gm}}_3·C_0}{\mathrm{gmd}·C_1}.$

[0012] The gm₃and gmd are the coefficients of terms v_gs(t)³and V_gs(t)v_ds(t), respectively, of Equation:

[0013] i_ds[v_gs,v_ds]=gm·v_gs(t)+gd·v_gs(t)+gm₂·

[0014] v_gs(t)²+gmd·v_gs(t)v_ds(t)+

[0015] gd₂·v_ds(t)²+gm₃·v_gs(t)³+gm₂d·

[0016] v_gs(t)²·v_ds(t)+

[0017] gmd₂·v_gs(t)·v_ds(t)²+gd₃·v_ds(t)³+. . .

[0018] which is a Taylor series extension formula of the drain-source non-linear current i_ds[v_gs,v_ds] of the transistor.

[0019] Also, C0 and C1 in the voltage gain are the coefficients of the constant term and term v_gs(t) in Equation C_gs[v_gs(t)]=C₀+C₁·v_gs(t)+C₂·v_gs(t)²+ . . . , which is a power series of a gate source capacitor of the transistor.

[0020] The above objective and advantage of the present invention will become more apparent by describing in detail a preferred embodiment thereof with reference to the attached drawings in which:

[0021] FIGS. 1A and 1B are block diagrams of conventional linear amplifiers each including a harmonic tuning circuit;

[0022] FIG. 2 is a block diagram of a conventional linear amplifier including a circuit for feeding an intermodulation component backward using a 2^ndorder intermodulation component;

[0023] FIG. 3 is a circuit diagram of a linear power amplification circuit using a 2^ndorder intermodulation feed-forwarding method according to the present invention;

[0024] FIG. 4A is an equivalent circuit diagram of the circuit of FIG. 3 with respect to a high frequency;

[0025] FIG. 4B is an equivalent circuit diagram of the circuit of FIG. 3 with respect to a low frequency;

[0026] FIG. 5 shows output signals depending on whether a low frequency 2^ndorder intermodulation feed-forwarding circuit is adopted, when 2-ton signals f₁and f₂are input to the circuit of FIG. 3; and

[0027] FIG. 6 shows output signals depending on whether a low frequency 2^ndorder intermodulation feed-forwarding circuit is adopted, when a code division multiple access (CDMA) signal is input to the circuit of FIG. 3.

[0028] FIG. 3 shows a low frequency 2^ndorder intermodulation feed-forwarding amplifier according to the present invention. FIG. 4A shows the equivalent circuit of FIG. 3 which is applied to a fundamental signal, that is, a high frequency signal, and FIG. 4B shows the equivalent circuit of FIG. 3 which is applied to a 2^ndorder harmonic component, that is, a low frequency signal.

[0029] In the fundamental signal processing shown in FIG. 3, a fundamental signal from a signal source V_Sis applied to the gate of a transistor 30 via an input impedance matching unit 32 and power-amplified and output to the drain. An output impedance matching unit 34 impedance-matches the amplified signal with an output load. A first direct current blocking filter 36, an amplifier 38 and a second direct current blocking filter 37 are serially connected to each other between the gate, which is an input port, and the drain, which is an output port, such that a 2^ndintermodulation signal is fed forward.

[0030] In FIGS. 4A and 4B, reference character Rs denotes the gate resistance of the transistor 30, reference character Rd denotes the drain resistance of the transistor 30, reference character Ri denotes an input resistance, and reference character Rs is a source resistance. An impedance component due to capacitance is particularly greater than other shown parameters. Accordingly, when Rs, Rd, Ri and Rg are set to be 0, a drain-source nonlinear channel current i_ds[v_gs, v_ds] is obtained by Taylor series expension from Equation 1:

[0031] i_ds[v_gs, v_ds]=gm·v_gs(t)+

[0032] gd·v_ds(t)+gm₂·v_gs(t)²+

[0033] gmd·v_gs(t)v_ds(t)+

[0034] gd₂·v_ds(t)²+gm₃·v_gs(t)³+gm₂d·v_gs(t)

[0035] 2·v_ds(t)+

[0036] gmd₂·v_gs(t)·v_ds(t)²+

[0037] gd₃·v_ds(t)³+ . . .   (1)

[0038] wherein a first order coefficient gm denotes trans conductance, gd denotes output conductance, and v_gsand v_dsdenote the alternating current (AC) voltage of a gate voltage and that of a drain voltage, respectively. When 2 tone input signals f₁and f₂are applied to the gate of the transistor 30, Equation 2 is established: $\begin{array}{cc}\begin{array}{c}{v}_{\mathrm{gs}}\left(t\right)=A_1·\cos \left(2\pi f_{1}t\right)+A_2·\cos \left(2\pi f_{2}t\right)\\ =\frac{1}{2}\left[A_1·\exp \left(j2\pi f_{1}t\right)+A_1·\exp \left(-j2\pi f_{1}t\right)\right]+\\ \frac{1}{2}\left[A_2·\exp \left(j2\pi f_{2}t\right)+A_2·\exp \left(-j2\pi f_{2}t\right)\right]\\ =\frac{1}{2}\left[V_{\mathrm{gs}-f_1}+V_{\mathrm{gs}-f_1}^{*}\right]+\frac{1}{2}\left[V_{\mathrm{gs}-f_2}+V_{\mathrm{gs}-f_2}^{*}\right]\end{array}& \left(2\right)\end{array}$

[0039] wherein V_gs−fis equal to A₁exp(j2nf₁), V_gs−fis equal to A₂exp(j2nf₂), V*_gs−fis equal to A₁exp(−j2nf₁), V*_gs−fis equal to A₂exp(−j2nf₂), 2-ton input signals f₁and f₂are signals within a bandwidth in a fundamental signal, and A₁and A₂denote the magnitudes of signals f₁and f₂, respectively.

[0040] When Equation 2 is substituted for Equation 1, the current of a main 3^rdintermodulation component (2f₂−f₁) is expressed as a term combined with a 3^rdcoefficient gm₃in Equation 1, as in Equation 3:

[0041] i_ds(t)|_2f−f^gm=¼gm₃·A₂²·A₁·cos [2π(2f₂−f₁)t]  (3)

[0042] The harmonic component i_gs(t) of a gate current due to the non-linearity of a gate-source capacitor (C_gs) can be expressed by the input voltage v_gsof the transistor 30, as in Equation 4: $\begin{array}{cc}{i}_{\mathrm{ds}}\left(t\right)=\left[C_0+C_1·v_{\mathrm{gs}}\left(t\right)+C_2·v_{\mathrm{gs}}{\left(t\right)}^2+\dots \right]\frac{d{v}_{\mathrm{ds}}\left(t\right)}{dt}& \left(4\right)\end{array}$

[0043] Since C_gs[v_gs(t)] is equal to C₀+C₁·v_gs(t)+C₂·v_gs(t)²+ . . . a low frequency 2^ndorder intermodulation component in Equation 4 can be expressed as in Equation 5:

[0044] i_ds(t)|_f−fj½C₁(f₂−f₁)[V_gs−f·V*_gs−f−V_gs−f·V*_gs−f]  (5)

[0045] A gate voltage for a 2^ndintermodulation component can be obtained using Equation 5 as in Equation 6: $\begin{array}{cc}\begin{array}{c}{{v_{\mathrm{gs}}\left(t\right)|}_{f_2-f_1}\approx \frac{1}{C_0}\int i_{\mathrm{gs}}\left(t\right)|}_{f_2-f_1}dt\\ =\frac{C_1}{4C_0}\left[V_{\mathrm{gs}-f_2}·V_{\mathrm{gs}-f_1}^{*}+V_{\mathrm{gs}-f_1}·V_{\mathrm{gs}-f_2}^{*}\right]\\ =\frac{C_1}{2C_0}{A}_2·A_1·\cos \left[2\pi \left(f_2-f_1\right)t\right]\end{array}& \left(6\right)\end{array}$

[0046] When the 2^ndorder intermodulation component f₂−f₁is amplified by a low frequency amplifier 38 and applied to the drain port, a drain voltage is calculated by Equation 7:

[0047] {circumflex over (v)}_ds(t)=v_ds(t)+β·v_ds(t)|_f−f  (7)

[0048] wherein β denotes the voltage gain of the low frequency amplifier 38, and v_ds(t) denotes a drain voltage when feed-forwarding is not performed.

[0049] When Equation 7 is substituted for Equation 1, a new 3^rdorder intermodulation current can be obtained. The new 3^rdintermodulation component is obtained by the interaction between an amplified 2^ndorder intermodulation component and a 1^storder main signal, and is expressed as the product of gmd and gd₂, which are 2^ndorder coefficients. However, since generally gmd>>gd₂, a main distortion component due to gmd is expressed as in Equation 8: $\begin{array}{cc}{{\hat{i}}_{\mathrm{ds}}\left(t\right)|}_{2f_{2-}{f}_1}^{{\mathrm{gm}}_d}=\frac{\mathrm{gmd}·\beta ·C_1·A_2^2·A_1}{4C_0}\cos \left[2\pi \left(2f_2-f_1\right)t\right]& \left(8\right)\end{array}$

[0050] A main 3^rdorder intermodulation component output when, not amplified using a feed-forwarding method from Equation 3, can be offset by a 3^rdorder intermodulation component generated from the amplified 2^ndintermodulation component from Equation 8. Hence, under an ideal condition in which the sum of Equations 3 and 8 is set to be 0, the voltage gain (β) of the low frequency amplifier 38 is calculated by Equation 9: $\begin{array}{cc}\beta =-\frac{{\mathrm{gm}}_3·C_0}{\mathrm{gmd}·C_1}& \left(9\right)\end{array}$

[0051] As shown in Equation 9, the voltage gain (β) of the low frequency amplifier 38, which is a pure real number value, is a constant which has no dependency on A₁and A₂which are the magnitudes of the input voltages of 2-tone signals f₁and f₂, respectively. Thus, even if the input voltages of the 2-tone signals f₁and f₂are changed to some extent, the linearity of a linear amplifier using a feed-forwarding method according to the present invention is considerably stably maintained, and a phase change in a significantly low frequency is so negligible that there is no need to adjust the phase.

[0052] In FIG. 5, when 2 ton frequency f₁and f₂included within a band are applied to a linear amplifier according to the present invention at a central frequency of 2.15 GHz, a 3^rdintermodulation component has magnitudes a and b in case that a 2^ndintermodulation component is not fed forward, and the 3^rdintermodulation component has magnitudes c and d in case that the 2^ndintermodulation component is fed forward. It can be seen from FIG. 5 that the 3^rdorder intermodulation component is improved by 18 dB in case that the 2^ndintermodulation component is fed forward.

[0053] FIG. 6 shows output signals with respect to an input code division multiple access (CDMA) signal, depending on whether the circuit of FIG. 3 adopts a low frequency 2^ndorder intermodulation feed-forwarding method. It can be seen from FIG. 6 that an adjacent channel power ratio (ACPR) is improved by 10 dB.

[0054] In the above embodiment of the present invention, an amplifier has been limited to a transistor. However, it is apparent that the parameters for the transistor can also be adopted in general power amplification transistors by one of ordinary skill in the art.

[0055] A linear amplifier according to the present invention feeds the low frequency 2^ndorder intermodulation component of a fundamental signal forward to the output port of a power amplification transistor, thereby maintaining the stable characteristics of a circuit adopting a feed-forwarding method and favorably offsetting a 3^rdintermodulation component that is adjacent to the frequency of the fundamental signal.