Broadband antenna incorporating both electric and magnetic dipole radiators

[0002] This application claims priority from U.S. Provisional Patent Application Serial No. 60/105,612, filed Oct. 26, 1998, and entitled “Broadband Antenna Incorporating Both Electric and Magnetic Dipole Radiators.”

[0003] The present invention relates generally to the field of broadband, reduced-size antennas for use in, e.g., HF and VHF communications, electromagnetic compatibility testing, electronic warfare, and ultrawideband and ground penetrating RADAR.

[0004] For most applications, including both communications and electromagnetic compatibility testing, it is generally desirable for antennas to be as small as possible for reasons of convenience, durability, and aesthetics. In the case of military communications, it is also often necessary for antennas to exhibit low observability (LO). In the HF (3-30 MHz) and VHF (30-300 MHz) bands for which wavelengths are on the order of meters to tens of meters, it is thus necessary to utilize electrically-small antennas, that is, antennas with geometrical dimensions which are small compared to the wavelengths of the electromagnetic fields they radiate. Quantitatively, electrically-small antennas are generally defined as antennas which fit inside a so-called radiansphere, that is a sphere with a radius, r=λ/2π, where λ is the wavelength of the electromagnetic energy radiated.

[0005] Electrically-small antennas exhibit large radiation quality factors Q; that is, they store (on time average) much more energy than they radiate. This leads to input impedances which are predominantly reactive, and, as a result, the antennas can be impedance-matched only over narrow bandwidths. Furthermore, because of the large radiation quality factors, the presence of even small resistive losses leads to very low radiation efficiencies. According to known quantitative predictions of the limits on the radiation Q of electrically small antennas, the minimum attainable radiation Q for any linearly polarized antenna which fits inside a spherical volume of radius a can be computed exactly, according to the equation: $\begin{array}{cc}Q=\frac{1}{\mathrm{ka}}+\frac{1}{k^3{a}^3}& \text{(Equ. 1)}\end{array}$

[0006] where k=1/λ, the wavenumber associated with the electromagnetic radiation. The available theories can be succinctly summarized by stating that the radiation Q of an electrically small antenna is roughly proportional to the inverse of its electrical volume. Furthermore, the radiation Q is essentially inversely proportional to the antenna bandwidth. Therefore, in order to achieve relatively broad bandwidth and high efficiency with a single-element electrically-small antenna of a given size, it is necessary to utilize as much as possible of the entire volume that an antenna occupies.

[0007] In order to achieve an antenna having this fundamental limit on radiation Q given in Equation 1, an antenna would have to excite only the TM₀₁or TE₀₁mode outside the enclosing spherical surface and store no electric or magnetic energy inside the spherical surface. So while, a Hertzian (short) dipole excites the TM₀₁mode, it does not satisfy the criterion of storing no energy within the sphere and thus will exhibit a higher radiation Q (and hence narrower bandwidth) than that predicted by Equation 1.

[0008] In general, all antennas which radiate dipolar fields, such as wire dipoles and loops, are limited by the constraint given in Equation 1. Some broadband dipole designs have been successfully implemented and approach the limit given in Equation 1. However, it is not possible to construct a linearly-polarized, isotropic antenna which exhibits a radiation Q less than that predicted by Equation 1.

[0009] While Equation 1 represents the fundamental limit on the radiation Q for a linearly polarized, omni-directional antenna, it is not the global lower limit on radiation Q. Instead, an antenna which radiates equal power into the TM₀₁and TE₀₁modes can (in principle) achieve a radiation Q of: $\begin{array}{cc}Q=\frac{1}{2}\left[\frac{1}{{\left(\mathrm{ka}\right)}^3}+\frac{2}{\mathrm{ka}}\right]& \text{(Equ. 2)}\end{array}$

[0010] A quality factor for an antenna which meets this characteristic is roughly half of that of an antenna which radiates only TM₀₁or TE₀₁, alone. As a result, the attainable impedance bandwidth of the antenna is nearly doubled. While an equipartition of radiated power in the two modes is required to achieve the radiation Q given in Equation 2, the polarization state and radiation pattern of the modes do not need to match, and instead can take on different forms depending on the relative phases and orientations of the modes. Although prior analysis has been performed on a very general class of antennas with equal electric and magnetic multipole moments, no specific antenna designs having these characteristics have been presented.

[0011] Ideal antennas having a pair of infinitesimally small, co-located, electric and magnetic dipoles oriented to provide orthogonal dipole moments have been theoretically and numerically examined previously and found to provide several useful features. Examples of such ideal antennas 10, 20 are shown in FIGS. 1 and 2. The antennas 10, 20 include an infinitesimal magnetic dipole loop 11, 21 with an associated feed 12, 22 and an infinitesimal electric (wire) dipole 13, 23 with an associated feed 14, 24. As can be appreciated, because the antenna elements are infinitesimally small, the shape of the loop is not crucial. Thus, the square loop 21 in FIG. 2 functions essentially equivalently to the circular loop 11 in FIG. 1.

[0012] For the theoretically-examined co-located pair antenna described above, the electric field, in the far field region, is given by the equations: $\begin{array}{cc}{E}_{\theta }\left[\frac{A}{\eta }\sin \theta +B\sin \phi \right]\frac{e^{-j\mathrm{kr}}}{r},\mathrm{and}& \text{(Equ. 3)}\\ E\phi =B\left[\cos \mathrm{\theta cos\phi }\right]\frac{e^{-j\mathrm{kr}}}{r}& \text{(Equ. 4)}\end{array}$

[0013] where A and B are weighting coefficients of the TM₀₁and TE₁₁modes respectively, and r, θ, and φ constitute a standard right-hand spherical coordinate system. If A=ηB then the directional gain of the antenna is given by the equation: $\begin{array}{cc}G\left(\theta ,\phi \right)=\frac{3\left[{\left(\sin \theta +\sin \phi \right)}^2+{\cos }^2{\mathrm{\theta cos}}^2\phi \right]}{4}& \text{(Equ. 5)}\end{array}$

[0014] and cardioid patterns with linear polarization are provided in the θ=90 plane and the φ=90 plane. FIG. 3 is a graph of the farfield gain pattern. As can be seen, a maximum gain of G_max=3.0 (4.77 dBi) is achieved at θ=90 and φ=90.

[0015] However, if A=jηB, the directional gain is: $\begin{array}{cc}G\left(\theta ,\phi \right)=\frac{3\left[{\left(\sin \theta \right)}^2+{\left(\sin \phi \right)}^2\right]}{4}& \text{(Equ. 6)}\end{array}$

[0016] The farfield gain pattern of such an antenna is depicted in FIG. 4. The maximum gain still occurs at θ=90. However, for this configuration, the maximum gain value _Gmaxis now only 1.5 (1.77 dBi). Therefore, as can be appreciated by one of skill in the art, the combination of an electric and a magnetic dipole with proper orientation, amplitude ratio, and relative phase results in a radiator with roughly half the radiation Q and as much as 3 dB more gain than an isolated dipole.

[0017] Another useful aspect of including both electric and magnetic dipole modes in an antenna is that the maximum power output (as limited by electric field breakdown in the nearfield) is improved. It can be seen physically and has been shown mathmatically, that, for purposes of producing maximum radiated power before electric field breakdown in the nearfield, the TE (magnetic multipole) modes and in particular, the TE₀₁mode are better. This is because the nearfield energy is magnetic as opposed to electric. Thus any admixture of TE modes is an improvement over a simple dipole antenna.

[0018] Previous work in this area has been limited primarily to theoretical and numerical investigations of co-located pairs of infinitesimal electric and magnetic dipoles (as well as ensembles of higher-order multipoles). While, as discussed above, the co-located pair of infinitesimal electric and magnetic dipoles has been shown to possess many valuable attributes, it is not a practical radiator. First, co-location is impossible when finite-sized elements are used. Furthermore, unless the elements have some appreciable electrical size, while still remaining electrically-small, broad band operation is impossible. Therefore, for an antenna to achieve multi-octave bandwidths, it is necessary for it to be electrically-small at the lower end of its operating frequency range, but only slightly so. In other words, the enclosing spherical surface has a radius of approximately λ/2π. This requirement is in stark contrast to “infinitesimally” small radiators having a radius on the order of λ/100.

[0019] In addition, the feed network for the electric and magnetic dipole combination is difficult to implement. Although possible feed networks have been previously suggested, none of the presently known designs suggest operate effectively over a broad frequency range. Thus, use of these designs negates any improvements in bandwidth provided by the lower radiation Q of the radiator.

[0020] In order to provide broadband operation, it is necessary that the relative amplitudes and phases of the electric and magnetic dipole radiation be maintained over the operating frequency range to within some finite tolerance. Having done this it is necessary to effectively impedance match the resulting antenna system to RF source. This is a particularly difficult problem due to the resonant nature of the combined electric and magnetic dipole radiator. To date, while extensive analyses extolling the desirable characteristics of idealized radiators combining electric and magnetic dipole radiation have been published, no practical systems have been implemented.

[0021] Accordingly, it would be advantageous to provide a practical antenna design which combines electric and magnetic dipole radiators to provide an antenna with a small quality factor Q.

[0022] It would be further advantageous if such an antenna had a broad bandwidth of operation and, in particular, maintained modal amplitude and phase matching of the electric and magnetic radiation as well as impedance matching over a wide range of frequencies.

[0023] According to the invention, a novel antenna design is presented which includes a broadband, electrically-small radiating element containing an electric dipole and a magnetic loop dipole oriented so that their dipole moments are orthogonal. A physical connection is provided between the electric and magnetic dipoles, which connection is displaced from the feed point of the antenna. By physically connecting a broadband electric dipole and a broadband loop much broader bandwidth is achieved than can be provided by a wire dipole and loop combination. In a particular embodiment, the antenna comprises a capacitively loaded bow-tie dipole antenna coupled to a dual-loop structure which is attached near the outer corners of the bow-tie dipole and operates in conjunction with the bow-tie dipole to form a magnetic dipole antenna. The new antenna configuration combines electric and magnetic dipole radiators in a single package and solves the above mentioned problems concerning maintaining modal amplitudes and phases as well as impedance matching.

[0024] The foregoing and other features of the present invention will be more readily apparent from the following detailed description and drawings of illustrative embodiments of the invention in which:

[0025] FIGS. 1 and 2 are illustrations a conventional co-located infinitesimal electric and magnetic dipole pairs;

[0026] FIG. 3 is a graph of the cardioid elevation pattern produced by an electric and magnetic dipole pair when equipartition of power is maintained and modal phase is 90 degrees;

[0027] FIG. 4 is a graph of the elevation pattern produced by an electric and magnetic dipole pair when equipartition of power is maintained but modal phase is zero degrees;

[0028] FIG. 5 is an illustration of an antenna according to the invention which incorporates electric and magnetic dipole radiation;

[0029] FIG. 6 is an exploded view of the antenna of FIG. 5;

[0030] FIG. 7 is an illustration of the magnetic and electric dipole components of the antenna of FIG. 5;

[0031] FIG. 8 is an illustration of the antenna of FIG. 5 formed using conductive sheet or mesh;

[0032] FIG. 9 is an illustration of the antenna of FIG. 5, further including interior support elements;

[0033] FIG. 10 is an illustration of the antenna of FIG. 9 formed using a combination of conductive frames and conductive sheet or mesh;

[0034] FIG. 11 is an illustration of the antenna of FIG. 5 including L-shaped top loading elements;

[0035] FIG. 12 is an illustration of the antenna of FIG. 5 including curved loop elements;

[0036] FIG. 13 is an illustration of an antenna according to a second embodiment of the invention which incorporates electric and magnetic dipole radiation;

[0037] FIG. 14 is an illustration of the antenna of FIG. 5 combined with a log periodic dipole array;

[0038] FIG. 15 is an illustration of the antenna of FIG. 14 formed using conductive sheet or mesh; and

[0039] FIG. 16 is a graph of gain vs. frequency of antenna of FIG. 5.

[0040] Turning now to FIG. 5, there is shown a compact broadband antenna 50 according to the invention which combines electric and magnetic dipole radiators. The construction of the antenna 50 may be more clearly understood using the exploded view of FIG. 6. Referring to FIGS. 5 and 6, the antenna 50 comprises a bow-tie dipole or tapered feed element 100 and illustrated here as a pair of triangular elements 100a, 100blying in the same plane. The bow-tie dipole 100 has a pair of central feeds 60a, 60b. The use of a tapered geometry greatly enhances radiation at the higher end of the operating range of the antenna 50. Although a planar, triangular bow-tie structure 100 is disclosed, it is understood that biconical or other tapered antenna elements may be used instead and the terms bow-tie and tapered feed will be used interchangeably throughout the following discussion.

[0041] Preferably, a pair of parallel U-shaped elements 101a, 101bextend substantially perpendicularly from the respective ends 104a, 104bof the bow-tie dipole 100 and provide top hat or capacitive loading of the bow-tie element 101 in order to lower its fundamental resonance and hence enhance its performance at the lower end of its opening frequency range. The pair of parallel elements 101a, 101btogether with the bow-tie dipole 100 generally form a tapered inverted-L dipole antenna. In addition, a pair of loops 102a, 102bare attached generally between the top outer comers 106a, 106band bottom outer corners 108a, 108b, respectively, of bow-tie dipole 100. Preferably, as illustrated, the loops 102a, 102bare parallel to each other and extend from the bow-tie 100 in an opposite direction from the capacitive loading conductors 101a, 101b.

[0042] FIG. 7 is an illustration of the magnetic and electric dipole components of the antenna formed by the antenna of FIG. 5. For clarity, the bow-tie element 100 is illustrated twice. As shown, an electric dipole antenna 110 is formed by the capacitively loaded bow-tie 100. In addition, the loops 102a, 102boperate in conjunction with the bow-tie dipole 100 to form a magnetic dipole 112. The electric and magnetic dipoles 110, 112 are merged in the antenna of FIG. 5 and are analogous to the idealized co-located electric and magnetic dipoles in FIGS. 1 and 2. However, unlike the idealized infinitesimal dipoles examined in theory, the antenna of the invention is a physically practical form.

[0043] The elements comprising the antenna embodiment 50 of FIG. 5 generally take the form of conductive frames. The conductive frames may be formed from any conductive material or combination of materials which may be shaped to form the elements shown. In a preferred embodiment, the conductive material is aluminum. In this and other embodiments of the antenna 50 recited herein, the various antenna elements may also be formed from conductive mesh, conductive sheets, or a combination thereof. In addition, it is understood that when a conductive mesh or sheet is used to form a section of the antenna, the frame for that section need not be conducting, although the use of a conducting frame is preferred.

[0044] For example, and with reference to FIG. 8, conductive sheet or mesh 114 may be used to “fill in” one or more of (a) the triangles formed by tapered feed elements 100a, 100b, (b) the rectangles formed by capacitive loading elements 101a, 101b, and (c) the area between the loops 102a, 102b. In embodiments of the antenna 50 which have conductive frames elements, the frames may contain interior elements which may be conductive or non-conductive. For example, FIG. 9 illustrates an embodiment of antenna 50 having interior support elements 116a, 116bplaced between the magnetic loop elements 102a, 102b. FIG. 10 illustrates the antenna of FIG. 9 having conductive frame elements and further including a conductive mesh or sheet 114 only between the loop portions 102a, 102b.

[0045] Various combinations of frames, mesh, and/or conductive sheets may be used to form an antenna according to the invention. The specific combinations used are dependent on design aspects, such as various mechanical design considerations. In addition, because of the electrically-small nature of the antenna 50, the exact shapes of the component elements are not critical. For example, the capacitive loading plates 101a, 101bneed not be exactly parallel to each other. Nor do they need to be exactly rectangular, but can have other regular or irregular shapes. Thus, as shown in FIG. 11, the ends 111a, 111bof the loading plates 101a, 101bmay be bent inwards, forming a pair of L-shaped elements having increased loading capacitance. The shape of the loop elements 102a, 102bcan also be distorted with minimal impact on the performance of the antenna. For example, as shown in FIG. 12, loop elements 102a, 102bmay be curved, rather than U-shaped.

[0046] In addition, the connection points of the pair of loops 102a, 102bto the bow-tie element can be moved closer together vertically along the opposed ends 104a, 104bsuch that the separation between the loops is reduced. It should be noted, that while parallel loop elements are preferred, the elements need not be parallel to each other and can also be tilted with respect to the horizontal plane. Further, the connection points of each of the loop elements to the bow-tie feed 100 can be moved inwards along the tapered edges of elements 100a, 100b, respectively, toward the feed points 60a, 60b, such that the connecting ends of the loop elements are closer together, resulting in a “tighter” loop. Preferably, the connections of the loops to the bow-tie feed 100 are displaced from the feed points 60a, 60bat least an electrically significant amount. Such a displacement modifies the input impedance of the antenna in such a way as to reduce the overall impedance level, especially in the vicinity of the first and second parallel resonances.

[0047] While the antenna 50 has been discussed above as having a pair of loop elements 102a, 102b, preferably attached between the top and bottom outer comers of the bow-tie element, as discussed, the loops need not be connected to the outermost comers of feed elements 100. Further, the number of loops may be varied, from a single loop to multiple loops. As shown in FIG. 13, a single loop 102cis connected between elements 100aand 100bat points 109a, 109b. The position of the connection points 109a, 109bcan vary in a manner similar to that discussed above with respect to a dual loop embodiment. Although this design may be somewhat lighter and more compact than embodiments for which the loop portion has a height comparable to that of the bow-tie feed element 100, i.e., as achieved by the use of two loops connecting the upper and lower corners, respectively, of the bow-tie elements 100a, 100b, such a single-loop embodiment may have a somewhat reduced bandwidth compared to those having a greater height.

[0048] In addition, as will be appreciated by those of skill in the art, if feed elements 100a, 100bare formed using conductive mesh or sheet, loop elements 102 could also connect at points within the perimeters of feed elements 100. Support elements within the interior of elements 100 may be needed to realize such a connection mechanically. Preferably, however, the loop elements are connected at or near the outermost points of the feed elements as shown and described above.

[0049] The broadband antenna 50 described above can be combined with a log periodic dipole array (LPDA) 120 to produce a very broadband directional antenna. Hybrid combinations of LPDAs and broadband, electrically-small radiating elements (such as a bowtie or biconical dipole) are sometimes constructed in order to augment the performance of the LPDA at the lower end of its operating range. The antenna described herein is particularly useful for such a system because its directional gain (4.77 dBi) approaches that of the low-gain LPDAs often used in such hybrid systems. Such a combination is shown in FIG. 14. A balun 121 is used to connected the LDPA 120 to feeds 60a, 60bof the antenna 50. A dielectric support assembly 122 is also provided to support cable 123 used to connect to the antenna 50. FIG. 15 illustrates the hybrid antenna formed with conductive mesh, similar to the antenna embodiment illustrated in FIG. 8. To provide for a convenient feed line connection, a portion 124 of the conductive mesh or screen 114 may be removed.

[0050] FIG. 16 is a graph of the forward gain vs. frequency of the antenna 50 calculated using Numerical Electromagnetics Code. A conventional broadband dipole provides only 1.7-2.1 DBi of directional gain. As can be seen, the new low-frequency antenna element disclosed herein exhibits much higher forward directional gain.

[0051] It should be noted that the input impedance of the antenna 50 constructed in accordance with the invention may vary significantly over its operating frequency range. Thus, it may be advantageous to use a matching transformer at the input to the antenna. When combining the antenna with an LPDA, it is generally advantageous to place this transformer between the LPDA and the broadband element 50. The selection of a suitable matching transformer is dependent on the geometries of the specific antenna configuration at issue and other factors known to those of skill in the art.