Optical control device for electronic scanning antenna

[0002] This application relates to French Patent Application 98-07240 incorporated by reference herein.

[0003] 1. Field of the Invention

[0004] This invention regards an optical control device for broadband radar transmission and reception. It is applicable to the control of broadband electronic scanning antennas to ensure both the formation of a beam for transmission and the reception of a beam reflected by a target.

[0005] 2. Discussion of the Background

[0006] An electric scanning antenna comprises many radiating elements that ensure both the transmission and the reception of an ultrahigh frequency signal. A transmission or reception beam is formed by all the signals transmitted or received by each element. To orient a beam in a given direction θ, it is necessary to create time delays between signals transmitted or received by the various radiating elements. To obtain an analogous effect, it is known how to create a phase delay between these signals. The phase difference φ1−φ2 between the signals transmitted or received by two radiating elements is given by the following equation: $\begin{array}{cc}{\Phi }_1-{\Phi }_2=\frac{d\sin \theta }{c}\times 2\pi f& \left(1\right)\end{array}$

[0007] where d represents the distance between the two radiating elements, f represents the frequency of the signals and c represents the speed of the light, the time delay created being $T_1-T_2=\frac{d\sin \theta }{c}.$

[0008] The phase difference φ1−φ2 is equal to 2πf(T₁−T₂).

[0009] The preceding equation (1) highlights a major disadvantage residing in the fact that the phase difference depends on the frequency. Consequently, if the frequency varies, the sighting angle varies as well. This method for orienting a beam is therefore not suited to broadband radar. However ultrahigh frequency techniques do not allow us to create a time delay between the signals other than through the creation of the preceding phase difference, except to implement a device that is prohibitive from a size and cost standpoint. Effectively, a theoretically simpler solution would be to create a delay directly between the signals supplied to the different radiating elements, but that would require cumbersome and costly ultrahigh frequency circuits, due more particularly to the unavoidable dimensions imposed by the wavelengths in question.

[0010] The use of optical techniques allows us to overcome the aforementioned disadvantage by controlling the radiating elements directly through time delays, without requiring the artifice of phase differences, these delays being created in the optical domain. To that effect, optical control solutions for electronic scanning antennas have already been implemented. With regard to transmission, numerous optical control architectures have already been proposed in order to control the radiation pattern during transmission. An example of optical architecture is given in French patent No. 90 03386.

[0011] With regard to reception, beam formation using time delays requires a very significant dynamic of all the delays, still inaccessible to the optical components. A direct architecture based on the bi-directional operation of the control developed for transmission therefore does not seem possible in the short- or intermediate-term. To mitigate this disadvantage, a correlation architecture was defined in particular in accordance with the description of French patent No. 94 11498. However, this type of architecture is restricted to radars with a small bandwidth, typically 10 MHz.

[0012] The disadvantage of a correlation architecture stems particularly from the fact that the use of complementary delays is incompatible with local oscillator signal frequencies and remote transmission signal frequencies, for example 500 MHz, which characterize broadband radar. This frequency difference is unavoidable for the proper operation of a radar system, particularly in order to avoid problems linked to aliasing.

[0013] One goal of the invention is more particularly to allow an architecture of the aforementioned type to function for a radar with a large bandwidth. To this end, the aim of the invention is an optical control device for electronic scanning antenna comprising radiating elements for controlling this device comprising a set of optical circuits for creating delays, each receiving a first light beam, polarized according to a first direction and having a first wavelength, this first beam being affected by an appropriate delay; and a second light beam, polarized according to a second direction, with a second wavelength. Each optical delay circuit induces complementary delays with respect to a determined time value on the light of the first and second beams that it receives, a chromatic separator is situated at the output of each delay circuit and separates the light with the first wavelength from the light with the second wavelength, and each radiating element of the antenna is coupled to the output of a delay circuit by a first photodetector. As the two beams are modulated at the transmission frequency, for each receiving signal of a radiating element, the local oscillator is supplied at the output of a first ultrahigh frequency mixer by mixing the transmission frequency and an intermediate frequency, then the frequency of the receiving signal supplied to the radar processing means, at an intermediate frequency augmented by the Doppler frequency of the signal received, is obtained at the output of a second ultrahigh frequency mixer by mixing the local oscillator frequency with the frequency of the signal received.

[0014] In an embodiment variation of a device according to the invention, as the two beams are modulated at the transmission frequency, for each receiving signal of a radiating element the local oscillator frequency is supplied at the output of a photo-mixer by mixing the frequency of the signal received carried by the optical wave and an intermediate frequency, then the frequency of the receiving signal supplied to the radar processing means, at an intermediate frequency augmented by the Doppler frequency of the signal received, is obtained at the output of an ultrahigh frequency mixer by mixing the local oscillator frequency with the transmission frequency.

[0015] The main advantages of the invention are that the invention makes it possible to avoid the transposition of the receiving signal on an optical carrier, while benefiting from the broadband processing offered by a time-delay architecture, and that it is simple to implement.

[0016] Other features and advantages of the invention will become clear in connection with the description that follows given in relation to the appended drawings, which represent:

[0017] FIG. 1, is a block diagram of a correlation optical control device;

[0018] FIG. 2, is a schematic illustration of an antenna configuration with, by way of example, two radiating elements;

[0019] FIG. 3, is a schematic illustration of a part of a correlation optical control example, showing mixers introduced behind each radiating element;

[0020] FIG. 4, is a block diagram of an example of a possible architecture present in a device according to the invention;

[0021] FIG. 5, is a schematic illustration of the delays applied to the radiating elements and of the wave thus transmitted toward a target;

[0022] FIGS. 6 and 7, are block diagram of other possible examples of embodiment of a device according to the invention.

[0023] FIG. 1 is an optical control device that functions for transmission and reception of the type, for example, described by French patent No. 94 11498. The signals emanating from the device are used for transmission, the supply of active modules or radiating elements and, for reception and for the generation of a local oscillator adapted in frequency and direction.

[0024] In this system, the light from beams F1, F2 is modulated in frequency. A first source L1 emits a single frequency light beam F1 with a wavelength λ1 (pulsation ω1 ). A translator at frequency T1 receives this light and transmits light at ω1 and light at ω1+2πfe modulated using a signal at frequency fe. The translator at frequency T1 is, for example, a Bragg acoustical-optical cell for frequencies roughly less than or equal to 5 GHz or an integrated optical device for higher frequencies.

[0025] A second light source transmits another single frequency light beam F2 with a wavelength λ2 (pulsation ω2). A T2 frequency translator receives this light and transmits light at ω2 and light at ω2+2π(fe+fo) modulated by a signal at frequency fe+fo.

[0026] In one application for controlling an electronic scanning antenna, the frequency fe is situated in the range of ultrahigh frequencies and corresponds to the transmission frequency of the antenna. The frequency fo is substituted for a local oscillator frequency for the antenna's receiving mode in the remainder of the description.

[0027] The light transmitted by the translator T1 is polarized according to a determined direction. The light transmitted by the translator T2 is polarized according to a direction perpendicular to that transmitted by T1.

[0028] An optical mixer system ME superimposes the light emanating from translator T1 onto that from translator T2. The resulting beam therefore comprises light polarized according to two orthogonal directions, as symbolized in FIG. 1, and at two different frequencies emanating from translators T1 and T2.

[0029] The resulting beam is stretched by a beam separator SE so that it is distributed over the different inputs of a set of delay circuits DCR.

[0030] This set of delay circuits DCR can, for example, be embodied like the set described in French patent No. 92 34 467.

[0031] Each delay circuit delays the light from source L1 and the light from source L2 differently. More precisely, according to an example of an embodiment, if T is the maximum delay induced by a delay circuit, the light from source L1 is delayed by a time ti and the light from source L2 is delayed by a time T−ti complementary to time T. The times T of the various delay circuits are equal, for example.

[0032] For example, the delay circuits DCR comprise a set of spatial light modulators comprising p×p pixels (same number of pixels as antenna radiating elements) and allow one to control the phase difference and the delay assigned to each of the p×p channels thus cut out. The delay circuits DCR supply delays in geometric progression so that N spatial modulators suffice to obtain 2N distinct delay values for each of the p×p channels of the architecture. Switching of delays is based on controlled rotation, owing to the spatial light modulators, of the polarization of the beams. In order to obtain a local oscillator that is directionally adapted, one exploits the property of the DCR, which is to generate, on each channel, complementary delays for combined input polarization states. Effectively, if the beam emanating from L1 undergoes a delay in channel i, then the beam emanating from L2 undergoes a delay T−ti,T being the transit time of the DCR.

[0033] Each Sd output of a delay circuit supplies light with a wavelength λ1 modulated at the frequency fe and light with a wavelength λ2 modulated at the frequency fe+fo.

[0034] Detection circuits PDRi and PDRn are connected to the Sd outputs, for example by optical fibers. These circuits are, for example, embodied as represented at the bottom right of FIG. 1. Each circuit comprises a chromatic separator MD that separates the light with a wavelength λ1 from the light with a wavelength λ2.

[0035] The light with a wavelength λ1 is transmitted to a photodetector PDRi,l, which transmits a photocurrent at frequency fe toward a radiating element ED1.

[0036] This photocurrent results from the interference between the light at ω1 and the light at ω1+2πfe.

[0037] The photocurrent transmitted is amplified by an amplifier so as to be compatible with the radiated power necessary for the transmission of the radiating element of the radar.

[0038] By allowing for suitable delays in the various delay circuits, one controls the antenna's radiation pattern. The antenna's transmission orientation is thus controlled optically.

[0039] Additionally, the light with a wavelength λ2 is transmitted to another photodetector PDRi,2 by the chromatic separator. This separator transmits a photocurrent resulting from the interference between the light at ω2 and ω2+2π(fe+fo). This photocurrent is applied to an ultrahigh frequency mixer Mk that also receives a signal received by an antenna element. It should be noted that a directional coupler CD permits, on the one hand, the coupling of the photocurrent of PDRi,1 to an antenna element in the direction of transmission and, on the other hand, the coupling of a detection current from an antenna element (in the direction of reception) to the frequency mixer Mk.

[0040] All the signals emanating from the photodetectors PDRi,2 in fact make up a local oscillator (homodyne or heterodyne) adapted to the antenna's direction of transmission.

[0041] Thus, the signal received by an antenna element EDk is amplified and applied jointly with the signal from PDRk2 to an ultrahigh frequency mixer Mk. In effect, if the signal transmitted by the antenna element EDk has the form S(t−τk), the same element receives a signal R(t′+τk), which must therefore be mixed with a local oscillator S′(t′+T+τk).

[0042] The low frequency signals from the mixers are processed according to two possibilities:

[0043] digitization at the level of each antenna element and summation of all these signals in a standard digital processor for fine beam formation through the FFC calculation. This processor may also be remote with respect to the receiving antenna by means of a reduced number of wavelength division multiplexed digital fiber optic links.

[0044] excitation of the p×p pixels of a bidimensional spatial light modulator using these low frequency p×p signals to implement coherent optical processing of the reverse channel.

[0045] With regard to the needs of a radar system, and assuming a system with two radiating elements powered by connections of variable lengths that introduce delays, one arrives for transmission at a configuration such as the one illustrated by FIG. 2. By way of example, this figure presents a system made up of two radiating elements S₁and S₂separated by a distance d and powered by variable delays l₁and l₂. An angle θ represents the sighting angle or boresight angle of the beam. A plane 21, perpendicular to the boresight angle 22 referenced as θ represents an equiphase plane, that is, a plane in which all the signals have the same phase. For both signals from the elements S₁and S₂to radiate in phase in the direction θ, the delays must have the following relation: $\begin{array}{cc}{l}_1-l_2=\frac{d\sin \theta }{c}& \left(2\right)\end{array}$

[0046] In a direct architecture, the receiving signals are transposed on an optical carrier. Since the system is reciprocal, the condition for the delays to be introduced to capture a wave in the direction θ is, in this case, strictly identical.

[0047] In a correlation architecture of the type shown in FIG. 1, mixers are introduced behind each radiating element as illustrated in FIG. 3. The mixers M1 and M2 situated behind each radiating element receive the signal to be received and the OL signal from the local oscillator having experienced a delay l₁′ and l₂′. The intermediate frequency signals Fi1 and Fi2 result from mixing the signal received at the transmission frequency RF and the signal OL of the local oscillator. This mixture is expressed by subtracting the frequencies and the phases of the signals. The phases ψ(Fi1) and ψ(Fi2) of the intermediate frequency signals Fi1 and Fi2 confirm the following equations: $\begin{array}{cc}\varphi \left(\mathrm{Fi1}\right)=-\left[\left({\omega }_{\mathrm{RF}}-{\omega }_{\mathrm{OL}}\right)t+{\Phi }_{\mathrm{r1}}+\frac{{\omega }_{\mathrm{OL}}}{c}\right]l_1^{\prime }& \left(3\right)\\ \varphi \left(\mathrm{Fi2}\right)=-\left[\left({\omega }_{\mathrm{RF}}-{\omega }_{\mathrm{OL}}\right)t+{\Phi }_{\mathrm{r2}}+\frac{{\omega }_{\mathrm{OL}}}{c}\right]l_2^{\prime }& \left(4\right)\end{array}$

[0048] where ω_RFand ω_OLrepresent respectively the pulsations of the receiving signals and of the local oscillator, and c represents the speed of the light.

[0049] Phases φ_r1and φ_r2represent the phases received on the dipoles.

[0050] For the wave coming from the direction θ to be received and summed, it is necessary that these phases confirm the following equation: $\begin{array}{cc}{\Phi }_{\mathrm{r2}}={\Phi }_{\mathrm{r1}}-\frac{d\sin \theta }{c}{\omega }_{\mathrm{RF}}& \left(5\right)\end{array}$

[0051] The following equation results:

[0052] ω_OL(l₂′−l₁′)=ω_RF(l₁−l₂)  (6)

[0053] By applying complementary delays to the receiving and ω_RFpulsation signals and to the signals from the local oscillator ω_OL, the paths of travel confirm that l₁+l₁′=L, L being a constant length and i being equal to 1 or 2, or more when the system comprises more than two radiating elements, as is generally the case. In particular, l₁+l₁′=l₂+l₂′. Then, according to the equation (6): $\begin{array}{cc}\left(l_1-l_2\right)=\frac{{\omega }_{\mathrm{OL}}}{{\omega }_{\mathrm{RF}}}\left(l_1-l_2\right)& \left(7\right)\end{array}$

[0054] It therefore appears that the condition of operation is verified only if the pulsations ω_RFand ω_OLare relatively close to one another.

[0055] Now, in the receiving mode of a standard radar system, the frequency of the local oscillator is chosen outside of the agility band of the radar system in order, more particularly, to avoid problems linked to aliasing. Consequently, the use of this type of architecture in receiving mode is restricted to radar systems with narrow bandwidths, for example approximately 10 MHz. It therefore appears necessary to make modifications to the operation, particularly of an architecture of the type shown in FIG. 1, so as to retain the broadband property for architectures with optical control using time delays.

[0056] One disadvantage of the correlation architecture therefore stems from the fact that the use of complementary delays is incompatible with frequencies f_oand f_eof the local oscillator and transmission signals, usually separated by 500 MHz for broadband radar. This frequency difference between the local oscillator and the transmission or receiving signal is necessary for the proper operation of the radar system in order to avoid problems linked to aliasing as indicated above.

[0057] To mitigate the aforementioned disadvantage, according to the invention, complementary delays are imprinted on the same frequency f_e, the transmission frequency, then the local oscillator OL with frequency f_OLis formed by mixing the transmission frequency with an intermediate frequency f_iproduced, for example, by a frequency generator common to all the channels. In this case, the two beams F1 and F2 are both modulated at the same frequency f_e, which is the transmission frequency. This dual mixing solution is illustrated by FIG. 4. This figure shows the circuits coupled with a radiating element EDk, of order k. The light polarized (41) at the output of the optical mixer ME undergoes a delay (42) by passing, for example, through the beam separator SE and the delay circuit DCR as illustrated by FIG. 1. The direct and complementary delays are imprinted on the same frequency f_e. To this end, a chromatic separator MD is situated at the output of the delay elements 42, more particularly on each of the outputs of the delay circuit DCR. This separator MD separates the light with wavelength λ₁from the light with wavelength λ₂. The light with wavelength λ₁is transmitted to a first photodetector PD1 that transmits a photocurrent with the transmission frequency f_etoward the radiating element EDk. A directional coupler CD is incorporated between this first photodetector PD1 and the radiating element EDk.

[0058] The light with wavelength λ₂is transmitted to a second photodetector PD2 that transmits a photocurrent with the transmission frequency f_etoward an input of a first ultrahigh frequency mixer Mk1. The other input of this mixer receives the aforementioned intermediate frequency f_i. The output of the first mixer Mk1 gives a frequency signal f_e+f_i. This signal acts as the local oscillator signal, the aforementioned frequency f_OLbeing equal to f_e+f_i. The output of the first mixer Mk1 is connected to the input of a second mixer Mk2, which therefore receives the frequency signal f_e+f_i. The other input of the second mixer Mk2 is connected to an output of the directional coupler CD, in the knowledge that one of its inputs is connected to the output of the first photodetector PD1 and that the other input/output is connected to the radiating element ED1. This directional coupler therefore allows one, on the one hand, to couple, in the direction of transmission, the photocurrent created by the first photodetector PD1 to the radiating element EDk, and, on the other hand, to couple, in the direction of reception, the radiating element EDk with the second mixer Mk2. The receiving signal supplied to this second mixer Mk2, via the directional coupler CD, has a frequency equal to that of the transmission frequency f_eaugmented by a Doppler frequency f_D. The receiving signal that enters the second mixer therefore has a frequency equal to f_e+f_D. The output signal from the second mixer consequently has a frequency equal to f_i+f_D, i.e., a frequency equal to the sum of the intermediate frequency and the Doppler frequency. In other words, one thus recovers at the output of the second mixer a signal at the intermediate frequency offset by the Doppler frequency. This signal is then processed by standard processing means for radar operations.

[0059] FIG. 5 shows an equiphase plane 51 of a wave transmitted from radiating elements EDk directed toward a target 52. Each radiating element is affected by a delay τ_k, produced in accordance with FIGS. 1 and 4. In this case, each radiating element of order k EDk transmits a signal whose phase φ_E(k) is defined by the following equation:

[0060] φ_E(k)=2πf_e(t−τ_k)  (8)

[0061] where τ_krepresents the delay on the channel of the radiating element EDk. During reception, this channel receives a signal of phase φ_R(k) defined by the following equation:

[0062] φ_R(k)=2πf_r(t−T−τ+τ_k)  (9)

[0063] where the receiving frequency f_r=f_e+f_D, f_Dbeing the Doppler offset frequency cited previously. The value T represents the round trip time for a signal transmitted to the target 53, more particularly for a signal transmitted from a first group of radiating elements ED1, as illustrated in FIG. 5.

[0064] The phases φ_OL, formed on the complementary channels are defined by the following equation:

[0065] φ_OL(k)=2πf_e(t−τ+τ_k)  (10)

[0066] The inclined local oscillator is obtained, at the output of the first mixer Mk1, by mixing the complementary channels with an intermediate frequency f_iwith a pure phase φ_fi=2π(f_i)t. The inclined local oscillator phase φ_OL′ is therefore given by the following equation:

[0067] φ_OL′(k)=2π(f_ef_i)t−2πf_e(τ−τ_k)  (11)

[0068] The second term 2πf_e(τ−τ_k) of the equation (11) represents a phase gradient at the transmission frequency, which causes this phase to follow an inclination law according to the transmission frequency f_e.

[0069] By once again mixing the inclined local oscillator signal with the signal received by means of the second mixer, a signal around the intermediate signal is obtained whose phase φ_fi(k) satisfies the following equation:

[0070] φ_fi(k)=2π(f_i+f_D)t+2πf_D(τ−τ_k)+2π(f_e+f_D)T  (12)

[0071] Considering the order of values in question, where notably f_Dis approximately 10³Hz and where τ and τ_kare approximately 10⁻⁸sec., the term f_D(τ−τ_k) is negligible. The signals on the different channels coupled with the various radiating elements EDk can therefore be summed in phase since there are no more terms that depend on the delays τ_k. The signal thus formed on a channel has a phase φ_signaldefined as follows:

[0072] φ_signal(k)=2π(f_i+f_D)t+2π(f_e+f_D)T  (13)

[0073] The first term of the equation (13) 2π(f₁+f_D) provides information about the speed of the target and the second term 2π(f_e+f_D)T, which is constant compared to the time t, provides information concerning the distance of the target, more specifically by calculation of the value T from which the distance is deduced. The equation (13) therefore shows that the signals from the various radiating elements EDk can be summed in phase without limiting the bandwidth. The very broadband property permitted by optical control is thus retained for reception.

[0074] In an optical control device with a correlation architecture, the dynamic constraint on the optical links is replaced by a stability constraint on the stability signal of the local oscillator. A dual mixture architecture according to the invention makes it possible to avoid the transposition of the receiving signal on an optical carrier while taking advantage of the broadband processing offered by an optical architecture with time delays.

[0075] FIG. 6 shows an embodiment variation of a device according to the invention. To economize one mixer per antenna element, for example, the first mixture is, for example, accomplished by the second photodetector PD2. Therefore in this case, this detector serves as both photodetector and ultrahigh frequency mixer. In this way, the number of mixers is reduced by half for all of the antenna.

[0076] In another embodiment variation, generation of the complementary delays necessary for the inclined local oscillator can also be obtained by doubling the number of pixels of the delay circuit DCR. In this case, p×p pixels are, for example, assigned to generate signals to be transmitted and other p×p pixels are each used to generate the local oscillator assigned to each radiating element. This embodiment variation has the advantage of greater flexibility of use. More particularly, it allows one to obtain different transmission and reception patterns. The delay law applied to the local oscillator is totally independent of the law applied to the signal transmitted. If τ_k^eand τ_j³represent the delays applied to the transmission respectively in channel k and channel j on the one hand, and if τ_k^OLand τ_j^OLrepresent the delays applied respectively to channel k and to channel j of the local oscillator on the other hand, then the construction of a suitable local oscillator is accomplished for a given frequency f_eof the band of the radar system by producing the following equation:

[0077] (τ_k^OL−τ_j^OL)f_OL=(τ_k^e−τ_j^e)f_e  (14)

[0078] where f_eand f_OLrepresent the frequencies of the local oscillator and of the transmission signal.

[0079] FIG. 7 shows a third variation of embodiment for a device according to the invention. In this variation, the functions of both mixers Mk1 and Mk2 are inverted The intermediate frequency signal f_lis mixed with the receiving signal at frequency f_e+f_Dby the second mixer Mk2 to form a local oscillator signal received at the output of this second mixer. This local oscillator signal received is then mixed with the local oscillator transmission signal, whose frequency is, in fact, the transmission frequency f_e, by the first mixer Mk1. This second mix gives a signal with frequency f_l+f_D, i.e., an intermediate frequency augmented by the Doppler frequency of the signal received. This second mix may also be accomplished directly by the second photodetector PD2 in accordance with the first variation presented in relation to FIG. 6.

[0080] Numerous modifications and variations are possible in light of the above teachings. It is therefore to be understood that within the scope of the appended claims, the present invention may be practiced other than as specifically described herein.