Operational filter building block

[0002] This is a continuation-in-part of copending, commonly assigned U.S. patent application Ser. No. 09/145,536, filed Sep. 2, 1998.

[0003] This invention relates to electrical filters. More particularly, this invention relates to 2nd order cascadable active-RC filters.

[0004] Electrical filters receive signals that typically oscillate between a maximum value and a minimum value (e.g., a sinusoidal signal). These signals are known as AC, or alternating current, signals. (In contrast, signals that maintain a substantially steady value are known as DC, or direct current, signals.) Each oscillation between a maximum and minimum value is a cycle, and the number of cycles per second is the frequency, which is measured in Hertz (one Hertz is one cycle per second). AC signals typically have more than one frequency component. These components can range from low frequencies to high frequencies (e.g., 100 Hz to 100k Hz).

[0005] Electrical filters attenuate, or filter out, one or more undesired frequency components from an AC signal, while permitting other frequency components of the signal to pass through. Depending on the undesired frequencies, different types of filters are used. For example, a low pass filter permits only frequencies below a cutoff frequency to pass through, while frequencies above the cutoff frequency are filtered out. Conversely, a high pass filter permits only frequencies above a cutoff frequency to pass through, while frequencies below the cutoff frequency are filtered out. Band-pass filters permit a range, or band, of frequencies (or only a single frequency) to pass through, while frequencies below a lower bandwidth-edge frequency and above an upper bandwidth-edge frequency are filtered out. Conversely, band-reject or notch filters permit all frequencies except a band of frequencies (or only a single frequency) to pass through. The frequencies allowed to pass through the filter are said to be in the passband, while the filtered out frequencies are said to be in the stopband.

[0006] Filters can be of different “orders.” For example, filters can be 2nd order low pass filters, 5th order low pass filters, 6th order band-pass filters, and 8th order high pass filters, among many others. The filter order relates mathematically to the transfer function of the filter. The filter transfer function is a ratio of the filter output to the filter input. Typically, this ratio is a function of signal frequency and phase. Filters of the 2nd order are useful because they can be cascaded to form higher order filters. Cascading is the coupling of filters into a series such that the output of one becomes the input of the next.

[0007] As is well known in the art, simple circuits including capacitors, inductors, and resistors can be used to construct low pass, high pass, band-pass, and notch “passive” filters (e.g., RLC filters). Passive filters provide no signal gain. As such, they are of limited value in many practical applications because signal gain is often required. Furthermore, inductors are generally avoided (particularly at low frequencies) because they have wide tolerances and are bulky, heavy, and non-linear.

[0008] “Active” filters provide signal gain and include passive elements and one or more active elements (e.g., transistor devices). Active elements have frequency dependent characteristics and are usually devices that are voltage-dependent or current-dependent. As is known in the art, active filters can be constructed with off-the-shelf operational amplifiers (op amps). However, such op amps usually require numerous external precision components, thus consuming large amounts of circuit board space. Moreover, precision components can be expensive.

[0009] Active filters of the 2nd order are characterized by various filter parameters, including center frequency (f_O), quality factor (Q), and filter gain. Cutoff frequencies, mentioned above with respect to low pass and high pass filters, are functions of the center frequency and quality factor. Furthermore, the center frequency, quality factor, and filter gain are functions of the various filter circuit elements, and can be calculated accordingly with known filter equations.

[0010] Active filters are typically either available as standard off-the-shelf (usually discrete) circuit devices with fixed filter functions and parameters, or are custom designed as either discrete or integrated circuit devices. In either case, such filters usually cannot be easily modified or adjusted to meet application requirements other than those they were originally designed for. In other words, filter functions and parameters usually cannot be easily modified or adjusted once the filter is manufactured, because doing so usually requires either adding additional components and elements, replacing one or more existing circuit elements with different elements (e.g., replacing a resistor with a capacitor), replacing one or more existing elements with elements of different value (e.g., replacing a 10 k ohm resistor with a 150 k ohm resistor), or all of the above.

[0011] For example, FIG. 1A shows a known 2nd order filter that provides low pass and band-pass frequency responses. Filter 100 includes op amps 103, 113, and 123; resistors 101, 107, 109, 111, 117, and 119; and capacitors 105 and 115. Band-pass response V₁is available at node 121, while low pass frequency responses V₂and V₃are respectively available at nodes 125 and 127. To customize filter 100 to particular filter parameters, values for each of the numerous circuit elements are determined based on a cumbersome series of known design equations.

[0012] To subsequently use filter 100 for another application requiring different filter parameters, the circuit element values again need to be determined. This will probably result in one or more of these elements requiring replacement. To replace such elements, sufficient access to and appropriate means of replacing them are required. Such a process is often impractical even if filter 100 is a discrete device, and is more likely impossible if filter 100 is an integrated circuit.

[0013] Similarly, modifying filter 100 to perform other filtering functions can be equally difficult. For example, to modify low pass filter 100 to provide 2nd order high pass filtering, the following circuit component and elements should be coupled to filter 100, as shown in FIG. 1B: op amp 139 and resistors 131, 133, 135, and 137. Additional calculations need to be performed to determine the values of resistors 131, 133, 135, and 137, and sufficient space needs to be available to add these parts. High pass frequency response V_HPis then available at node 138. However, depending on the specified filter parameters, the values of the other circuit elements of filter 130 may also need to be recalculated. This probably will require that one of more of these elements be replaced. Again, this process often is impractical if not impossible.

[0014] Known notch filters, such as filters 160 and 190, shown respectively in FIGS. 1C and 1D, also cannot be easily modified or adjusted once constructed. Furthermore, constructing notch filters 160 and 190 with filters 100 or 130 is typically cumbersome and impractical.

[0015] Notch filter 160 includes op amps 163 and 181, integrators 171 and 173, and resistors 161, 165, 167, 169, 175, 177, and 179. Notch response V_N1is available at node 180 and, as shown in FIG. 1C., is obtained by summing high pass response V_HPat node 164 with low pass response V_LPat node 174. The notch frequency f_N(i.e., the signal frequency filtered out) is equal to the following: $f_N=f_O\sqrt{\frac{R_{169}{R}_{177}}{R_{165}{R}_{175}}}\mathrm{where}{f}_O=\frac{1}{2\pi \mathrm{RC}}\sqrt{\frac{R_{165}}{R_{169}}}$

[0016] and R and C are the combined internal resistance and capacitance of integrators 171 and 173. The values of resistors 165, 169, 175, and 177, and the RC value of integrators 171 and 173 accordingly determine the notch frequency, which can be higher, lower, or equal to the center frequency of notch filter 160. Modifying the notch frequency will require replacement of one or more of these circuit elements, which again is often impractical even if filter 160 is a discrete device, and is more likely impossible if filter 160 is an integrated circuit.

[0017] Known notch filter 190 provides notch frequency response V_N2by summing input signal V_INwith inverting band-pass output response V_BPas shown in FIG. 1D. Notch filter 190 includes 2nd order inverting band-pass filter 191 (of which many circuit configurations are known), op amp 199, and resistors 193, 195, and 197. Although notch filter 190 requires only a few external components, filter 190 is limited to a notch frequency equaling the center As frequency. Another disadvantage of notch filter 190 is that if precision components are not used, the notch frequency will not be completely filtered out, which will result in some signal gain at that frequency.

[0018] In sum, users are left with few choices for meeting particular filter applications. For example, users can search for an available off-the-shelf filter; laboriously modify, if possible, an existing filter; or custom design a new filter.

[0019] In view of the foregoing, it would be desirable to provide a filter circuit that can be easily configured to provide 2nd order low pass, band-pass, notch, and high pass frequency responses.

[0020] It would also be desirable to provide a filter circuit that can be easily configured to provide a selectable center frequency, quality factor, and gain.

[0021] It would further be desirable to provide a plurality of filter circuits that can be easily configured to construct various types of active filters of 2nd order or higher.

[0022] It would still further be desirable to provide a plurality of filter circuits that can be fabricated on a single integrated circuit chip.

[0023] It is an object of this invention to provide a filter circuit that can be easily configured to provide 2nd order low pass, band-pass, notch, and high pass frequency responses.

[0024] It is also an object of this invention to provide a filter circuit that can be easily configured to provide a selectable center frequency, quality factor, and gain.

[0025] It is a further object of this invention to provide a plurality of filter circuits that can be easily configured to construct various types of active filters of 2nd order or higher.

[0026] It is a still further object of this invention to provide a plurality of filter circuits that can be fabricated on a single integrated circuit chip.

[0027] In accordance with this invention, a 3-terminal operational filter circuit is provided. The filter circuit includes an input terminal for receiving an input signal, a first output terminal for providing a first frequency response, and a second output terminal for providing a second frequency response. The filter circuit also includes an operational amplifier circuit, a capacitor, and a non-inverting integrator. The operational amplifier circuit has an inverting input, a non-inverting input and an amplifier output. The inverting input is coupled to the input terminal, the non-inverting input is coupled to ground, and the amplifier output is coupled to the second output terminal. The capacitor is coupled between the inverting input and the amplifier output. The input of the non-inverting integrator is coupled to the amplifier output, and the integrator output is coupled to the first output terminal.

[0028] The filter circuit can be advantageously configured to provide either 2nd order low pass and band-pass frequency responses, 3rd order low pass or high pass frequency responses, or 2nd order band-pass and high pass frequency responses. The filter circuit can be configured to provide these responses by coupling at least three additional circuit elements (e.g., resistors) to the three terminals of the filter circuit. Moreover, the filter circuit can be manufactured to operate within a selected range of center frequencies. A user then can select a particular center frequency, quality factor, and gain by simply selecting particular values for the additional circuit elements. Furthermore, various types of higher order filters can be constructed by cascading filter circuits.

[0029] The filter circuit can be also configured to provide notch responses (i.e., frequencies at which signals are substantially filtered out). In one embodiment of the invention, the filter circuit can provide a notch at a frequency less than the center frequency. In another embodiment, the filter circuit can provide a notch at a frequency greater than the center frequency. In still another embodiment, the filter circuit can provide a notch at a frequency less than, equal to, or greater than the center frequency.

[0030] The invention also provides a configurable operational filter that includes a plurality of 3-terminal operational filter circuits fabricated preferably as an integrated circuit chip. The configurable filter can easily be configured to construct various types of active filters by simply coupling a few additional circuit elements (e.g., resistors and capacitors) to the configurable filter.

[0031] The above and other objects and advantages of the invention will be apparent upon consideration of the following detailed description, taken in conjunction with the accompanying drawings, in which like reference characters refer to like parts throughout, and in which:

[0032] FIG. 1A-D are block diagrams of known 2nd order filters;

[0033] FIG. 2 is a block diagram of an exemplary embodiment of a 3-terminal operational filter circuit according to the print invention;

[0034] FIG. 3 is a block diagram of an exemplary embodiment of the non-inverting integrator of the filter circuit of FIG. 2 according to the present invention;

[0035] FIG. 4 is a block diagram of the filter circuit of FIG. 2 configured to provide 2nd order low pass and band-pass frequency responses according to the present invention;

[0036] FIG. 5 is a block diagram of two of the filter circuits of FIG. 2 cascaded to provide a higher order response according to the present invention;

[0037] FIG. 6 is a block diagram of the filter circuit of FIG. 2 configured to provide 2nd order band-pass and high pass frequency responses according to the present invention;

[0038] FIG. 7 is a block diagram of an exemplary embodiment of a configurable operational filter constructed with the filter circuit of FIG. 2 according to the present invention;

[0039] FIG. 8 is a circuit diagram of an exemplary embodiment of the bias generator of the configurable filter of FIG. 7 according to the present invention;

[0040] FIG. 9 is a circuit diagram of an exemplary embodiment of the operational amplifier of the filter circuit of FIG. 2 and the operational amplifier and inverter of the non-inverting integrator of FIG. 3 according to the present invention;

[0041] FIG. 10 is a block diagram of an exemplary embodiment of a dual 4th order low pass filter constructed with the configurable filter of FIG. 7 according to the present invention;

[0042] FIG. 11 is a block diagram of an exemplary embodiment of an 8th order low pass elliptic filter constructed with the configurable filter of FIG. 7 according to the present invention;

[0043] FIG. 12 is a block diagram of an exemplary embodiment of an 8th order high pass filter constructed with the configurable filter of FIG. 7 according to the present invention;

[0044] FIG. 13 is a block diagram of an exemplary embodiment of a quadruple 3rd order low pass filter constructed with the configurable filter of FIG. 7 according to the present invention;

[0045] FIG. 14A is a block diagram of an exemplary embodiment of a notch filter constructed with the filter circuit of FIG. 2 according to the present invention;

[0046] FIG. 14B is a block diagram of an exemplary embodiment of a high order cascaded notch filter constructed with the filter circuit of FIG. 2 according to the present invention;

[0047] FIG. 15 is a general illustration of a gain response of the notch filter of FIG. 14 according to the present invention;

[0048] FIG. 16 is a general illustration of another gain response of the notch filter of FIG. 14 according to the present invention;

[0049] FIG. 17 is a block diagram of another exemplary embodiment of a notch filter constructed with the filter circuit of FIG. 2 according to the present invention;

[0050] FIG. 18 is a general illustration of a gain response of the notch filter of FIG. 17 according to the present invention;

[0051] FIG. 19 is a block diagram of still another exemplary embodiment of a notch filter constructed with the filter circuit of FIG. 2 according to the present invention; and

[0052] FIG. 20 is a block diagram of yet another exemplary embodiment of a high order notch filter constructed with the filter circuit of FIG. 2 according to the present invention.

[0053] The present invention provides a building block active-RC filter circuit that can be used to construct various types of active filters of 2nd order or higher. The present invention advantageously requires few precision components, and can be used in many applications, including high resolution systems, anti-aliasing filters, smoothing or reconstruction filters, data communications, equalizers, dual I-and-Q channels, linear phase filtering, and replacement of LC filter modules.

[0054] FIG. 2 shows an exemplary embodiment of a 3-terminal operational filter circuit 200 constructed in accordance with the present invention. Filter circuit 200 is preferably fabricated as an integrated circuit. However, filter circuit 200 can alternatively be packaged as a 3-terminal device with discrete components.

[0055] Filter circuit 200 includes three terminals, input terminal 204, which receives input signals; output terminal 202, which provides a first frequency response; and output terminal 206, which provides a second frequency response. Filter circuit 200 further includes operational amplifier (op amp) circuit 208, which can be a commercially available part; filter capacitor 210, which is preferably an internal precision component and preferably variable; and non-inverting integrator 212. To power op amp circuit 208 and non-inverting integrator 212 (as is known in the art), filter circuit 200 also includes at least two terminals (not shown in FIG. 2) for coupling to power and ground.

[0056] Non-inverting integrator 212 performs the following function: $\frac{1}{s\left(R_{\mathrm{INTERNAL}}\right)\left(C_{\mathrm{INTERNAL}}\right)}$

[0057] where R_INTERNALand C_INTERNALare the respective internal resistance and capacitance of filter circuit 200. The operator “s” is equal to jω, where j is the imaginary number {square root over (−1+L )}, and ω is the angular frequency, which is equal to 2πf (f is frequency).

[0058] While non-inverting integrator 212 can be constructed in any known manner, an exemplary embodiment of non-inverting integrator 212 constructed in accordance with the present invention is shown in FIG. 3. Integrator 300 includes inverter 314, filter resistor 316, op amp circuit 318, and filter capacitor 320. Preferably, resistor 316 and capacitor 320 are each a precision element and preferably each are variable.

[0059] Note that filter circuit 200 represents a portion of a 2nd order continuous-time filter. This advantageously permits filter circuit 200 to be easily configured to provide, for example, 2nd order low pass and band-pass frequency responses, 3rd order low pass or high pass frequency responses, or 2nd order band-pass and high pass frequency responses. For example, to provide 2nd order low pass and band-pass frequency responses, resistors 401, 403, and 405 are coupled to filter circuit 200 in accordance with the present invention as shown in FIG. 4. Low pass frequency response V_LPis provided at output terminal 202, and band-pass frequency response V_BPis provided at output terminal 206.

[0060] In contrast to known active-RC filters, which are typically manufactured with fixed center frequencies, filter circuit 200 can be advantageously manufactured with different ranges of center frequencies. As shown below, center frequency f_Ois a function of C_INTERNAL, R_INTERNAL, and resistor 403. $f_O=\frac{1}{2\pi C_{\mathrm{INTERNAL}}\sqrt{R_{\mathrm{INTERNAL}}{R}_{403}}}$

[0061] Filter circuit 200 can therefore be manufactured with selected values for R_INTERNALand C_INTERNAL, which in the embodiments shown in FIGS. 2 and 3, are as follows:

[0062] C_INTERNAL=C₂₁₀C₃₂₀

[0063] R_INTERNAL=R₃₁₆

[0064] A user can then fine tune filter circuit 200 to a particular center frequency within that range by selecting a particular value for resistor 403.

[0065] To be able to select various values for resistor 316 and capacitors 210 and 320, resistor 316 and capacitors 210 and 320 can each be fabricated as a parallel arrangement of fixed value elements and fusible metal links or switches (which is known in the art and not shown). Once the particular values of resistor 316 and capacitors 210 and 320 are determined, either the appropriate switches are closed or the appropriate links are blown by laser to set the determined values.

[0066] The range of center frequencies is limited mainly by the practical magnitude limits of resistor 403 and their effects on filter circuit 200. For example, if the value of resistor 316 is set to about 10 k ohms and the value of C_INTERNALis set to about 159 pF, the resulting center frequency range is about 10 k Hz to 150 k Hz. This range corresponds to an upper resistor 403 limit of about 1M ohm (f_O=10 k Hz), and a lower resistor 403 limit of about 4.4k ohms (f^O=150 k Hz). However, because of increased output loading and increased total-harmonic-distortion effects that result from such a low valued resistor, the lower limit of resistor 403 is preferably at least 5 k ohms.

[0067] Continuing with the above example, if a user requires a center frequency of about 100 k Hz, solving for resistor 403 in the center frequency equation yields a value of about 10 k ohms. Similarly, if another user required a center frequency of 50 k Hz, resistor 403 should be about 40 k ohms. This feature of filter circuit 200 advantageously accommodates users with differing filter requirements.

[0068] Returning to FIG. 4, the transfer functions of the low pass and band-pass responses of filter 400 are, respectively, $\begin{array}{c}{H}_{\mathrm{LP}}\left(s\right)=\frac{-H_L{\omega }_O^2}{s^2+\left({\omega }_O/Q\right)s+{\omega }_O^2}\\ H_{\mathrm{BP}}\left(s\right)=\frac{-H_B\left({\omega }_O/Q\right)s}{s^2+\left({\omega }_O/Q\right)s+{\omega }_O^2}\end{array}$

[0069] where: $\begin{array}{c}{H}_L\left(\mathrm{low}\mathrm{pass}\mathrm{gain}\right)=\frac{R_{403}}{R_{401}}\\ H_B\text{(band-pass gain)}=\frac{R_{405}}{R_{401}}\\ Q=\frac{R_{405}}{\sqrt{R_{316}{R}_{403}}}\end{array}$

[0070] and

[0071] ω_O=2πf_O

[0072] Thus, as shown above, the center frequency, quality factor, and gains of filter 400 can be easily set by selecting particular values for resistors 401, 403, and 405. Advantageously, no clocks, phase lock loops, digital signal processing, or tuning cycles are required to set these filter parameters.

[0073] FIG. 5 shows two filter circuits 200 cascaded to advantageously construct a filter for higher order band-pass responses in accordance with the present invention. As shown, resistor 511 is simply coupled between the terminal 206A (the band-pass output) and input terminal 204B to cascade filter circuits 200A and 200B. Advantageously, filter 500 can be easily converted to a higher order low pass filter by simply coupling resistor 511 between terminals 202A and 204B instead of between terminals 206A and 204B.

[0074] The input of each filter circuit 200 is a virtual-ground summing point for a corresponding filter circuit 200. This gives each filter circuit 200 the capability to perform analog operations such as gain (preamplification), summing and weighting of multiple inputs, and accepting current or charge signals directly. Note that the transfer functions of filter 400 are inverting (indicated by the minus sign). Thus, when two filter circuits 200 are cascaded as shown in FIG. 5, these phase inversions cancel, and the resulting filter 500 is a non-inverting filter.

[0075] To provide 2nd order band-pass and high pass frequency responses, capacitor 601 and resistors 603 and 605 are coupled to filter circuit 200 in accordance with the present invention as shown in FIG. 6. Note that filter 200 easily changes from a low pass filter to a high pass filter by simply replacing resistor 401 with capacitor 601. High pass frequency response V_HPis provided at output terminal 206, and band-pass frequency response V_BPis now provided at output terminal 202.

[0076] The band-pass transfer function of filter 600 is the same as the band-pass transfer function of filter 400. The high pass transfer function of filter 600 is as follows: $\begin{array}{c}{H}_{\mathrm{HP}}\left(s\right)=\frac{-H_H{s}^2}{s^2+\left({\omega }_O/Q\right)s+{\omega }_O^2}\\ \begin{array}{c}\text{where:}\\ \begin{array}{c}{H}_H\left(\mathrm{high}\mathrm{pass}\mathrm{gain}\right)=\frac{C_{601}}{C_{\mathrm{INTERNAL}}}\\ Q=\frac{R_{605}}{\sqrt{R_{316}{R}_{603}}}\\ {\omega }_O=2\pi f_O\\ \mathrm{and}\\ f_O=\frac{1}{2\pi C_{\mathrm{INTERNAL}}\sqrt{R_{\mathrm{INTERNAL}}{R}_{603}}}\end{array}\end{array}\end{array}$

[0077] The replacement of resistor 401 with capacitor 601 modifies the band-pass gain of filter 600 as follows: $H_B\text{(band-pass gain)}=\frac{R_{605}}{R_{316}}\frac{C_{601}}{C_{\mathrm{INTERNAL}}}$

[0078] Again, C_INTERNALis the internal capacitance of filter 200, which in the embodiments of filter circuit 200 and integrator 300 shown respectively in FIGS. 2 and 3 is

[0079] C_INTERNAL=C₂₁₀C₃₂₀

[0080] Note that the replacement of resistor 401 with capacitor 601 affects only the gains. Thus, if resistors 603 and 605 are respectively equal to resistors 403 and 405, the center frequencies and quality factors of filters 600 and 400 are equal.

[0081] To configure higher order high pass filters, filter circuit 600 can be similarly cascaded as filter circuit 500, except that resistors 501 and 511 are replaced with capacitors.

[0082] FIG. 7 shows an exemplary embodiment of a configurable operational filter constructed with a plurality of filter circuits 200 in accordance with the present invention. Configurable filter 700 is preferably fabricated as an integrated circuit chip. However, configurable filter 700 can be alternatively constructed as a discrete component device.

[0083] Configurable filter 700 includes filter circuits 200A-D having input terminals 704A-D and output terminals 702A-D and12th order (i.e., twelve poles, another mathematical concept related to the transfer function) can be configured with configurable filter 700. Note that while four filter circuits 200 are shown in FIG. 7 for illustrative purposes, other numbers of filter circuits 200 can be used to construct configurable filter 700.

[0084] Configurable filter 700 also includes voltage input terminal 722 for preferably receiving a positive voltage V+, voltage input terminal 724 for preferably receiving a negative voltage V−, and ground <terminal 726 for coupling to a DC or AC ground. Preferably, voltage input terminals 722 and 724 are each bypassed with 0.1 μF capacitors to AC ground (as shown in FIGS. 10-13).

[0085] Ground terminal 726 is coupled to an internal node of configurable filter 700 (not shown), which is the midpoint of a resistive voltage divider between voltage input terminals 722 and 724. This results in a half-supply voltage that serves as an internal ground reference that advantageously permits configurable filter 700 to operate with single or dual supply voltages. For dual supply operation, ground terminal 726 is coupled to an AC or DC ground. For single supply operation, voltage input terminal 724 (V−) is coupled to system ground (the ground of the system using configurable filter 700). Ground terminal 726 is bypassed with at least a 0.1 μF capacitor to the system ground at voltage input terminal 724 (V−). The internal ground reference establishes a reference voltage for the input and output signals in single supply operation.

[0086] Configurable filter 700 preferably includes shutdown terminal 728 and shutdown circuitry 730. Shutdown circuitry 730 is coupled to shutdown terminal 728, voltage terminals 722 and 724, and filter circuits 200A-D. (For clarity, these connections are not shown in FIG. 7; furthermore, because shutdown circuitry 730 can be implemented by persons of ordinary skill in the art in accordance with the following description, shutdown circuitry 730 is not shown). Shutdown circuitry 730 drives configurable filter 700 into a shutdown state when shutdown terminal 728 is not receiving a logical 0 signal. In other words, if shutdown terminal 728 receives a logical 1 signal, or is left open-circuited, shutdown circuitry 730 generates two disable signals OFF and OFFB that disable bias generator 732 and filter circuits 200A-D. This causes ground terminal 726 and output terminals 702A-D and 706A-D to enter a high impedance state. A small pull-up current source at shutdown terminal 728 defaults configurable filter 700 to the shutdown state when shutdown terminal 728 is left open-circuited.

[0087] Configurable filter 700 also preferably includes bias generator 732, which biases filter circuits 200A-D (for clarity, connections to and from bias generator 732 are not shown in FIG. 7). FIG. 8 shows an exemplary embodiment of bias generator 732 in accordance with the present invention. Bias generator 800 preferably includes a 5-bit programmable binary resistor array 840 to control the supply current of configurable filter 700. Resistor array 840 adjusts for process variations and advantageously permits fabrication of lower power versions of configurable filter 700. Resistor array 840 includes weighted resistors 841-846 and fuses 847-851. Resistors 841-846 are preferable weighted as follows: resistor 842 is 2× resistor 843, resistor 843 is 2 resistor 844, resistor 844 is 2 resistor 845, and resistor 845 is 2 resistor 846. Fuses 847-851 are preferably laser-fusible.

[0088] Bias generator 800 also includes PMOS transistor 852, which is preferably a long channel device that preferably provides a few microamperes of startup current to enable bias generator 800 to be DC self-starting under all conditions. Bias generator 800 further includes PMOS transistors 853-857 and 859-862; NPN transistors 863, 864, and 868; NMOS transistors 865-867; and capacitor 858. Output nodes 869-872 are respectively coupled to filter circuits 200A-D, and outputs OUT0-3 respectively supply bias current to filter circuits 200A-D. Signals OFFB and OFF received at respective nodes 873,874 are generated by shutdown circuitry 730.

[0089] FIG. 9 shows an exemplary embodiment of operational amplifier circuitry preferably used to construct inverter 314 and operational amplifiers 208 and 318 in accordance with the present invention. Operational amplifier (op-amp) circuit 900 includes output node 957, inverting input node 958, and non-inverting input node 959. Nodes 960 and 961 respectively receive signals OFFB and OFF generated by shutdown circuitry 730.

[0090] Op amp circuit 900 is preferably an N-well CMOS circuit that advantageously does not use NMOS devices in the high gain portion of the signal path because of their inferior 1/f noise. Instead, N-well isolated NPN transistors 941,942 form a folded-cascode stage feeding high impedance PMOS current mirror 943,944,963,964,965. Resistors 945,946 behave like current sources and are not a significant source of noise. PMOS transistor 947 is a level shift device that permits a convenient DC level at the first stage output (the collector of NPN transistor 942), and NPN transistor 948 is a clamp for rapid overload recovery driven by signal BIAS1.

[0091] Signals BIAS1 and BIAS2 are outputs of circuitry described in commonly-owned U.S. patent application Ser. No. 09/113,618, filed Jul. 10, 1998, which is hereby incorporated by reference herein in its entirety.

[0092] The output stage includes PMOS transistors 949 and 951 and NMOS transistors 950, 952, and 953. NMOS transistor 950 and PMOS transistor 951 operate together as a composite NMOS device whose source voltage is set by signal BIAS2 on the gate of PMOS transistor 951, and whose equivalent “k” or g_m(transconductance) parameter relates directly to those of NMOS transistor 950 and PMOS transistor 951. NMOS transistors 952,953 form a 1:8 current mirror that draws the resulting drain current and combines it with that of PMOS transistor 949 in proper polarity. This arrangement allows the output devices to swing rail-to-rail (the signal voltage range including the power supplies), but also causes the large-signal non-linearities of the PMOS and NMOS output paths to cancel. Signal BIAS2 drives the gate of PMOS transistor 951 such that PMOS transistor 949 and NMOS transistor 950 operate with the same quiescent gate voltage.

[0093] This open loop linear output stage prevents additional distortion at signal frequencies, where the overall loop gain is reduced. The output stage operates in class-A push-pull until the load current exceeds its quiescent current in either direction, and then the output stage makes a smooth symmetrical transition to class-AB with higher efficiency. Filter circuit 200 can therefore be programmed for lower current operation at a cost of increased distortion.

[0094] At the input stage of op amp circuit 900, a tradeoff exists between distortion and noise. With a unity gain frequency ω_Uand a given input-pair current source 954, sizing input PMOS transistors 955,956 affects distortion and noise in opposite directions. (A high ratio of op amp slew rate to signal slew rate reduces distortion. Slew rate is a time rate of change of, for example, an output voltage under specified conditions.) Preferably, noise is favored in this tradeoff, because noise is the limiting factor in the dynamic range of signals that the filter can process.

[0095] FIGS. 10-13 show exemplary embodiments of several types of active-RC filters constructed with configurable filter 700 in accordance with the present invention (for clarity, shutdown circuitry 730 and bias ,generator 732 are not shown in FIGS. 10-13).

[0096] FIG. 10 shows an exemplary embodiment of a dual 4th order 100 k Hz Butterworth low pass filter 1000 constructed with configurable filter 700. A Butterworth filter provides a maximally-flat passband frequency response. Each half of filter 1000 provides a unity-passband-gain low pass response. The positive voltage V+ is preferably about 5 volts and the negative voltage V− is preferably about −5 volts. The circuit elements of filter 1000 are preferably about as follows:

[0097] resistor 1001A=10 k ohms

[0098] resistor 1005A=5.62 k ohms

[0099] resistor 1003A=10 k ohms

[0100] resistor 1001B=10 k ohms

[0101] resistor 1005B=13 k ohms

[0102] resistor 1003B=10 k ohms

[0103] resistor 1001C=10 k ohms

[0104] resistor 1005C=5.62 k ohms

[0105] resistor 1003C=10 k ohms

[0106] resistor 1001D=10 k ohms

[0107] resistor 1005D=13 k ohms

[0108] resistor 1003D=10 k ohms

[0109] capacitor 1007=0.1 μF

[0110] capacitor 1009=0.1 μF

[0111] FIG. 11 shows an exemplary embodiment of an 8th order 100 k Hz low pass elliptic filter 1100 constructed with configurable filter 700. Filter 1100 includes additional capacitors 1111A and 1111C in the input path, and sums these parallel resistor-capacitor paths at input terminals 704A-D to produce two notches in the low pass stopband (the band of frequencies filtered out). The positive voltage V+ is preferably about 5 volts and the negative voltage V− is preferably about −5 volts. The circuit elements of filter 1100 are preferably about as follows:

[0112] resistor 1101A=19.6 k ohms

[0113] capacitor 1111A=87 pF

[0114] resistor 1105A=13 k ohms

[0115] resistor 1103A=8.87 k ohms

[0116] resistor 1101B=8.06 k ohms

[0117] resistor 1111B=12.1 k ohms

[0118] resistor 1105B=8.87 k ohms

[0119] resistor 1103B=12.1 k ohms

[0120] resistor 1101C=69.8 k ohms

[0121] capacitor 1111C=47 pF

[0122] resistor 1105C=28 k ohms

[0123] resistor 1103C=8.87 k ohms

[0124] resistor 1001D=7.32 k ohms

[0125] resistor 1111D=17.8 k ohms

[0126] resistor 1105D=6.98 k ohms

[0127] resistor 1103D=17.8 k ohms

[0128] capacitor 1107=0.1 μF

[0129] capacitor 1109=0.1 μF

[0130] FIG. 12 shows an exemplary embodiment of an 8th order Chebyshev high pass filter 1200 with about 0.05 dB ripple and a cutoff frequency of about 30 k Hz constructed with configurable filter 700. Ripples are increases and decreases in gain within the passband (the range of frequencies allowed to pass through the filter). Each of the four cascaded filter circuits 200A-D has an external capacitor in the input path (capacitors 1201A-D). Resistors 1203A-D and 1205A-D set the respective center frequency and quality factor of filter circuits 200A-D. The positive voltage V+ is preferably about 5 volts and the negative voltage V− is preferably about −5 volts. The circuit elements of filter 1200 are preferably about as follows:

[0131] capacitor 1201A=150 pF

[0132] resistor 1205A=10.2 k ohms

[0133] resistor 1203A=35.7 k ohms

[0134] capacitor 1201B=150 pF

[0135] resistor 1205B=22.1 k ohms

[0136] resistor 1203B=66.5 k ohms

[0137] capacitor 1201C=150 pF

[0138] resistor 1205C=54.9 k ohms

[0139] resistor 1203C=107 k ohms

[0140] capacitor 1201D=150 pF

[0141] resistor 1205D=98.9 k ohms

[0142] resistor 1203D=127 k ohms

[0143] capacitor 1207=0.1 μF

[0144] capacitor 1209=0.1 μF

[0145] FIG. 13 shows an exemplary embodiment of quadruple 3rd order 100 k Hz Butterworth low pass filter 1300 constructed with configurable filter 700. Filter 1300 illustrates the flexibility of the virtual-ground inputs by permitting an extra, independent real pole to be added by replacing the input resistor (e.g., resistor 401 of FIG. 4) with an R-C-R “T” network. As shown in FIG. 13, the input resistor is split into two parts (resistors 1301A-D and 1321A-D). This parallel combination forms a 100 k Hz real pole (for the values shown below) with external capacitors 1311A-D. As shown, four such 3rd order Butterworth low pass filters can be constructed with configurable filter 700 for a total of twelve poles. The same technique can advantageously add additional real poles to other filter configurations as well, such as, for example, modifying dual 4th order low pass filter 1000 to produce a dual 5th order or dual 6th order filter. The positive voltage V+ of filter 1300 is preferably about 5 volts and the negative voltage V− is preferably about −5 volts. The circuit elements are preferably about as follows:

[0146] resistor 1301A=6.19 k ohms

[0147] capacitor 1311A=680 pF

[0148] resistor 1321A=3.83 k ohms

[0149] resistor 1305A=10 k ohms

[0150] resistor 1303A=10 k ohms

[0151] resistor 1301B=6.19 k ohms

[0152] capacitor 1311B=680 pF

[0153] resistor 1321B=3.83 k ohms

[0154] resistor 1305B=10 k ohms

[0155] resistor 1303B=10 k ohms

[0156] resistor 1301C=6.19 k ohms

[0157] capacitor 1311C=680 pF

[0158] resistor 1321C=3.83 k ohms

[0159] resistor 1305C=10 k ohms

[0160] resistor 1303C=10 k ohms

[0161] resistor 1301D=6.19 k ohms

[0162] capacitor 1311D=680 pF

[0163] resistor 1321D=3.83 k ohms

[0164] resistor 1305D=10 k ohms

[0165] resistor 1303D=10 k ohms

[0166] capacitor 1307=0.1μF

[0167] capacitor 1309=0.1 μF

[0168] Note that an exemplary embodiment of a quadruple 3rd order high pass filter can be constructed with filter 1300 simply by replacing resistors 1301A-D and 1321A-D with capacitors, and by replacing capacitors 1311A-D with resistors. The high pass outputs are then provided at terminals 706A-D of configurable filter 700.

[0169] Filter circuit 200 and configurable filter 700 can also be used to construct notch, or band-reject, filters. Notch filters ideally have zero signal gain at a notch frequency f_N. The input-output transfer function of a 2nd order notch filter is $H_N\left(s\right)=\frac{-H_N\left(s^2+{\omega }_N^2\right)}{s^2+\left({\omega }_O/Q\right)s+{\omega }_O^2}$

[0170] where

[0171] ω_N=2πf_N

[0172] and H_Nis set by circuit component values as described further below. The center frequency f_Oand quality factor Q are set as described above with respect to FIGS. 3, 4, and 6. At DC input voltages (i.e., f=0), the gain is $H_N\frac{f_N^2}{f_O^2}$

[0173] At very high input signal frequencies (f>>f_N), the gain is H_N. Thus, in addition to the notch, the gain of notch filter 1400 varies by a factor of: $\frac{\text{High Frequency Gain}}{DC\mathrm{Gain}}=\frac{f_O^2}{f_N^2}$

[0174] Notch filters can be constructed by adding a signal to a filtered replica of itself having equal gain and a 180° phase difference at the desired notch frequency f_N. The two signals then cancel each other out at f_N. The “notch depth” (i.e., the degree of signal cancellation) will be infinite to the extent that the two signals have matching gains. In other words, if the gains match and the signals are 180° out of phase at frequency f_N, the gain of the summed signals at f_Nis zero.

[0175] FIG. 14A shows notch filter 1400 constructed in accordance with the present invention. Notch filter 1400 provides a notch response V_Nat the output terminal of op amp 1410 when the inverted (i.e., 180° out-of-phase) band-pass output signal at output terminal 206 is summed with input signal V_INat op amp 1410 as shown, provided that the following resistor-value relationship exists: $\frac{R_{1411}}{R_{1417}}=\frac{R_{1405}}{R_{1401}}$

[0176] The depth of the notch depends on the accuracy of these resistor ratios. Note that op amp 1410 and resistor 1415 serve as a virtual-ground summing circuit.

[0177] If the value of capacitor 1402 is zero, the notch produced by filter 1400 equals the center frequency of the filter and the notch transfer function parameters are:

[0178] f_N=f_O $H_N=\frac{R_{1415}}{R_{1417}}$

[0179] Because f_N=f_O, the gain both at DC and at very high frequencies (f>>f_N) is H_N. The notch response for filter 1400 under these conditions is shown in FIG. 15. Such a response is impractical as a cascaded part of a high pass, low pass or band-pass filter (the most common uses for notches), because the notch frequency is limited to the center frequency.

[0180] If the value of capacitor 1402 is not zero, the notch frequency is greater than the center frequency and the notch response includes a low pass response as well as a notch, as shown in FIG. 16. Capacitor 1402 advantageously contributes phase lead, which increases the notch frequency above the center frequency of filter circuit 200A. For complete signal cancellation at f_N, the previous resistor-value relationship should be maintained. The notch transfer function parameters are: $f_N=\begin{array}{c}{f}_O\\ \sqrt{1-\frac{R_{1401}{C}_{1402}}{R_{1405}{C}_{\mathrm{INTERNAL}}}}\end{array}\mathrm{and}{H}_N=\frac{R_{1415}}{R_{1417}}\frac{f_O^2}{f_N^2}$

[0181] Notch filter 1400 is typically most useful as a stopband notch in a low pass filter or as an upper stopband notch in a band-pass filter. Resistors 1411 and 1417 simply replace the input resistor of either a low pass or resistor-input band-pass section of a cascaded filter circuit 200 (e.g., resistor 511 of filter circuit 200B of FIG. 5).

[0182] FIG. 14B shows high order cascaded notch filter 1450 constructed with filter circuit 200 in accordance with the present invention. Notch filter 1450 provides a two-notch 8-pole low pass response V_N. The frequency of the two notches are each greater than the center frequency. Notch filter 1450 includes filter circuits 200A-D, capacitors 1402 and 1452, and resistors 1401, 1403, 1405, 1411, 1413, 1415, 1417, 1451, 1453, 1455, 1461, 1463, 1465, and 1467, coupled as shown. Filter circuits 200A and 200C each create a notch, and filter circuits 200B and 200D each function as a summing circuit.

[0183] FIG. 17 shows notch filter 1700 constructed in accordance with the present invention. By summing the inverted low pass output signal at output terminal 202 with input signal V_INat op amp 1710 as shown, notch filter 1700 produces a notch at a frequency f_N<f_O. If the value of capacitor 1702 is zero, notch frequency fN is zero (i.e., the notch occurs when the input signal is DC).

[0184] If the value of capacitor 1702 is not zero, capacitor 1702 provides phase lead that shifts the notch frequency up from zero, resulting in a low frequency notch with a high pass response, as shown in the notch frequency response of FIG. 18. The notch transfer function parameters are: $f_N=f_O\sqrt{1-\frac{R_{\mathrm{INTERNAL}}{C}_{\mathrm{INTERNAL}}{R}_{1703}}{R_{1705}{C}_{1702}{R}_{1701}}}\mathrm{and}$ $H_N=\frac{R_{1715}}{R_{1717}}$

[0185] Complete signal cancellation at f_Noccurs when the following element and component values satisfy the relationship: $\frac{R_{1711}}{R_{1717}}=\frac{R_{1705}{C}_{1702}}{R_{\mathrm{INTERNAL}}{C}_{\mathrm{INTERNAL}}}$

[0186] Notch filter 1700 is typically most useful as a lower stopband notch in a band-pass filter. Resistors 1711 and 1717 simply replace the input resistor of a cascaded filter circuit 200 (e.g., resistor 511 of filter circuit 200B of FIG. 5).

[0187] FIG. 19 shows notch filter 1900 constructed in accordance with the present invention. Notch filter 1900 provides notches by summing the low pass and band-pass output signals, which are 90° out of phase, through resistor 1911 and capacitor 1912, respectively, at op amp 1910 as shown. The low pass output signal is provided at output terminal 202 and the band-pass output signal is provided at output terminal 206. Capacitor 1912 provides another 90° phase shift to the band-pass output signal. This results in a wideband 180° phase difference. Furthermore, the amplitude ratio between currents I_Rand I_Cis frequency-dependent. At the frequency where I_Rand I_Chave equal magnitude, Io becomes zero and a notch occurs. The notch transfer function parameters are: $f_N=\frac{1}{2\pi \sqrt{R_{1911}{C}_{1912}{R}_{\mathrm{INTERNAL}}{C}_{\mathrm{INTERNAL}}}}\mathrm{and}$ $H_N=\frac{R_{1915}}{R_{1901}}\frac{C_{1912}}{C_{\mathrm{INTERNAL}}}$

[0188] Notch filter 1900 provides the following gain relationships: $\begin{array}{c}\mathrm{DC}\mathrm{Gain}=\frac{R_{1915}}{R_{1901}}\frac{R_{1903}}{R_{1911}}\\ \frac{f_O^2}{f_N^2}=\frac{\mathrm{High}\mathrm{Frequency}\mathrm{Gain}}{\mathrm{DC}\mathrm{Gain}}=\frac{R_{1911}}{R_{1903}}\frac{C_{1912}}{C_{\mathrm{INTERNAL}}}\end{array}$

[0189] Advantageously, the completeness of signal cancellation at a notch frequency does not depend on element/component value matching relationships as in notch filters 1400 and 1700. Notch filter 1900 can therefore provide deep notches (i.e., substantially complete signal cancellation) independent of these element/component relationships. For example, variations in the values of resistor 1911 or capacitor 1912 vary the notch frequency, not the degree of signal cancellation. Moreover, as shown above, the notch frequency is independent of the center frequency. The notch frequency can thus be equal to, higher than, or lower than the center frequency without reconfiguring the circuit elements as is necessary to go to and from notch filters 1400 and 1700 when a notch frequency on the other side of the center frequency is desired.

[0190] Higher orders of notch filter 1900 can be constructed in accordance with the present invention by cascading filter circuits 200 as shown in FIG. 20. Notch filter 2000 includes filter circuits 200A and 200B, capacitors 1912 and 2012, and resistors 1901, 1903, 1905, 1911, 1915, 2003, and 2011. Node 2014 can be coupled to a summing circuit (e.g., op amp 1910 and resistor 1915) to provide a two-notch 4-pole gain response where one notch is created by filter circuit 200A and another is created by filter circuit 200B.

[0191] Alternatively, node 2014 can be coupled to another filter circuit 200 to obtain higher order responses. For example, if node 2014 were coupled to input terminal 204C of filter circuit 200C (not shown), notch filter 2000 could provide any of the following responses: if filter circuit 200C functioned as a summing circuit, then a two-notch 6-pole low pass or band-pass gain response is provided at output terminal 202C or 206C, respectively. If filter circuit 200C were configured as filter circuit 200B with node 2014C (not shown) coupled to a summing circuit, then notch filter 2000 would provide a three-notch 6-pole gain response where filter circuits 200A-C each create a notch. Notch filter 2000 can be accordingly cascaded further.

[0192] Thus it is seen that a 3-terminal operational filter circuit is provided that can be easily configured into various types of active filters. One skilled in the art will appreciate that the present invention can be practiced by other than the described embodiments, which are presented for purposes of illustration and not of limitation, and the present invention is limited only by the claims which follow.