On-Chip Word Line Voltage Generation for DRAM Embedded in Logic Process

1. Field of the Invention

The present invention relates to Dynamic Random Accessible Memory (DRAM). More particularly, this invention relates to DRAM fabricated using a conventional logic process. This invention further relates to the on-chip generation of precision voltages for the operation of DRAM embedded or fabricated using a conventional logic process.

2. Related Art

FIG. 1A is a schematic diagram of a conventional DRAM cell 100 that is fabricated using a conventional logic process. FIG. 1B is a cross sectional view of DRAM cell 100. As used herein, a conventional logic process is defined as a semiconductor fabrication process that uses only one layer of polysilicon and provides for either a single-well or twin-well structure. DRAM cell 100 consists of a p-channel MOS access transistor 1 having a gate terminal 9 connected to word line 3, a drain terminal 17 connected to bit line 5, and a source terminal 18 connected to the gate 11 of a p-channel MOS transistor 2. P-channel transistor 2 is configured to operate as a charge storage capacitor. The source and drain 19 of transistor 2 are commonly connected. The source, drain and channel of transistor 2 are connected to receive a fixed plate bias voltage V.sub.pp. The V.sub.pp voltage is a positive boosted voltage that is higher than the positive supply voltage V.sub.dd by more than a transistor threshold voltage V.sub.t.

As used herein, the electrode of the charge storage capacitor is defined as the node coupled to the access transistor, and the counter-electrode of the charge storage capacitor is defined as the node coupled to receive a fixed plate bias voltage. Thus, in DRAM cell 100, the gate 11 of transistor 2 forms the electrode of the charge storage capacitor, and the channel region of transistor 2 forms the counter-electrode of the charge storage capacitor.

To improve soft-error-rate sensitivity of DRAM cell 100, the cell is fabricated in an n-well region 14 which is located in a p-type substrate 8. To minimize the sub-threshold leakage of access transistor 1, n-well 14 is biased at the V.sub.pp voltage (at n-type contact region 21). However, such a well bias increases the junction leakage. As a result, the bias voltage of n-well 14 is selected such that the sub-threshold leakage is reduced without significantly increasing the junction leakage. When storing charge in the storage capacitor, bit line 5 is brought to the appropriate level (i.e., V.sub.dd or V.sub.SS) and word line 3 is activated to turn on access transistor 1. As a result, the electrode of the storage capacitor is charged. To maximize the stored charge, word line 3 is required to be driven to a negative boosted voltage V.sub.bb that is lower than the supply voltage V.sub.SS minus the absolute value of the threshold voltage (V.sub.tp) of access transistor 1.

In the data retention state, access transistor 1 is turned off by driving word line 3 to the V.sub.dd supply voltage. To maximize the charge storage of the capacitor, the counter electrode is biased at the positive boosted voltage V.sub.pp. The plate voltage V.sub.pp is limited by the oxide breakdown voltage of the transistor 2 forming the charge storage capacitor.

DRAM cell 100 and its variations are documented in U.S. Pat. No. 5,600,598, entitled "Memory Cell and Wordline Driver For Embedded DRAM in ASIC Process," by K. Skjaveland, R. Township, P. Gillingham (hereinafter referred to as "Skjaveland et al."), and "A 768k Embedded DRAM for 1.244 Gb.s ATM Switch in a 0.8 um Logic Process," P. Gillingham, B. Hold, I. Mes, C. O'Connell, P. Schofield, K. Skjaveland, R. Torrance, T. Wojcicki, H. Chow, Digest of ISSCC, 1996, pp. 262-263 (hereinafter referred to as "Gillingham et al.). Both Skjaveland et al. and Gillingham et al. describe memory cells that are contained in an n-well formed in a p-type substrate.

FIG. 2 is a schematic diagram of a word line control circuit 200 including a word line driver circuit 201 and a word line boost generator 202 described by Gillingham et al.. Word line control circuit 200 includes p-channel transistors 211-217, inverters 221-229, NAND gates 231-232 and NOR gate 241, which are connected as illustrated. Word line driver 201 includes p-channel pull up transistor 211, which enables an associated word line to be pulled up to the V.sub.dd supply voltage. P-channel pull down transistors 212-217 are provided so that the word line can be boosted down to a negative voltage (i.e., -1.5 V) substantially below the negative supply voltage V.sub.SS. However, the p-channel pull down transistors 212-217 have a drive capability much smaller (approximately half) than an NMOS transistor of similar size. As a result, the word line turn on of Gillingham et al. is relatively slow (>10 ns). Furthermore, in the data retention state, word line driver 201 only drives the word line to the V.sub.dd supply voltage. As a result, the sub-threshold leakage of the access transistor in the memory cells may not be adequately suppressed.

DRAM cells similar to DRAM cell 100 have also been formed using n-channel transistors fabricated in a p-type well region. To maximize stored charge in such n-channel DRAM cells during memory cell access, the associated word line is driven to a voltage higher than the supply voltage V.sub.dd plus the absolute value of the threshold voltage (V.sub.tn) of the access transistor. In the data retention state, the n-channel access transistor is turned off by driving the word line to V.sub.SS supply voltage (0 Volts) To maximize the charge storage of the capacitor in an n-channel DRAM cell, the counter electrode is biased at a plate voltage V.sub.bb that is lower than the V.sub.SS supply voltage.

A prior art scheme using n-channel DRAM cells includes the one described by Hashimoto et al. in "An Embedded DRAM Module using a Dual Sense Amplifier Architecture in a Logic Process", 1997 IEEE International Solid-State Circuits Conference, pp. 64-65 and 431. A p-type substrate is used, such that the memory cells are directly in contact with the substrate and are not isolated by any well structure. In the described design, substrate bias is not permitted. Moreover, application of a negative voltage to the word line is not applicable to ASICs that restrict substrate biasing to be zero. Consequently, the architecture achieves a negative gate-to-source voltage (V.sub.gs) by limiting bit line swing. The negative V.sub.gs voltage reduces sub-threshold leakage in the memory cells. Hashimoto et al. fails to describe the structure of the word line driver.

It would therefore be desirable to have a word line driver circuit that improves the leakage currents in DRAM cells fabricated using a conventional logic process. Moreover, it would be desirable to have improved methods for biasing DRAM cells fabricated using a conventional logic process.

Accordingly, the present invention provides a memory system that includes a dynamic random access memory (DRAM) cell, a word line, and a CMOS word line driver fabricated using a conventional logic process. The word line driver is controlled to selectively provide a positive boosted voltage and a negative boosted voltage to the word line, thereby controlling access to the DRAM cell.

A positive boosted voltage generator is provided to generate the positive boosted voltage, such that the positive boosted voltage is greater than the V.sub.dd supply voltage but less than the V.sub.dd supply voltage plus the absolute value of a transistor threshold voltage V.sub.t.

Similarly, a negative boosted voltage generator is provided to generate the negative boosted voltage, such that the negative boosted voltage is less than the V.sub.SS supply voltage, but greater than the V.sub.SS supply voltage minus the absolute value of a transistor threshold voltage V.sub.t.

A coupling circuit is provided between the word line driver and one of the positive or negative boosted voltage generators. For example, if the DRAM cell is constructed from PMOS transistors, then the coupling circuit couples the word line driver to the negative boosted word line generator. When the DRAM cell is being accessed, the coupling circuit couples the word line driver to the negative boosted voltage, thereby turning on the p-channel access transistor of the DRAM cell. However, when the DRAM cell is not being accessed, the coupling circuit couples the word line driver to the V.sub.SS supply voltage, thereby minimizing leakage currents associated with the negative boosted voltage.

In this embodiment, the coupling circuit can be configured to provide the V.sub.SS supply voltage to the word line driver when the word line is first activated. When the voltage on the word line falls below the V.sub.dd supply voltage, the coupling circuit provides the negative boosted voltage to the word line driver.

Conversely, if the DRAM cell is constructed from NMOS transistors, then the coupling circuit couples the word line driver to the positive boosted word line generator. When the DRAM cell is being accessed, the coupling circuit couples the word line driver to the positive boosted voltage, thereby turning on the n-channel access transistor of the DRAM cell. However, when the DRAM cell is not being accessed, the coupling circuit couples the word line driver to the V.sub.dd supply voltage, thereby minimizing leakage currents associated with the positive boosted voltage.

In this embodiment, the coupling circuit can be configured to provide the V.sub.dd supply voltage to the word line driver when the word line is first activated. When the voltage on the word line rises above the V.sub.SS supply voltage, the coupling circuit provides the positive boosted voltage to the word line driver.

The positive boosted voltage generator includes a charge pump control circuit that limits the positive boosted voltage to a voltage less than V.sub.dd plus V.sub.t. Similarly, the negative boosted voltage generator includes a charge pump control circuit that limits the negative boosted voltage to a voltage greater than V.sub.SS minus V.sub.t. The positive and negative boosted voltages are limited in this manner because, for normal logic applications using sub 0.25 micron processes, the gate oxide breakdown voltage is usually less than a threshold voltage V.sub.t above the positive supply voltage V.sub.dd.

The positive boosted voltage generator includes a charge pump control circuit that limits the positive boosted voltage to a voltage that is greater than the V.sub.dd supply voltage by less than one transistor threshold voltage. In one embodiment, this charge pump control circuit includes a first p-channel transistor having a source coupled to the V.sub.dd supply voltage and a drain coupled to a first reference current source. The gate of the first p-channel transistor is coupled to the gate of a second p-channel transistor. The first and second p-channel transistors have first and second channel widths, respectively, wherein the second channel width is greater than the first channel width. A second reference current source is coupled to the drain of the second p-channel transistor. The drain of the second p-channel transistor provides an inhibit control signal for the charge pump control circuit. A third p-channel transistor has a gate and a drain connected to a source of the second p-channel transistor, and a source coupled to receive the positive boosted voltage. The ratio of the first and second channel widths is selected such that the inhibit control signal is asserted when positive boosted voltage is less than one transistor threshold voltage greater than the V.sub.dd supply voltage. In one embodiment, the first reference current source has a negative temperature coefficient to compensate for temperature effects of the second p-channel transistor. The second reference current source can be provided with a positive temperature coefficient to compensate for temperature effects of the third p-channel transistor.

Similarly, the negative boosted voltage generator includes a charge pump control circuit that limits the negative boosted voltage to a voltage that is less than the V.sub.SS supply voltage by less than one transistor threshold voltage V.sub.t. In one embodiment, this charge pump control circuit includes a first n-channel transistor having a source coupled to the V.sub.SS supply voltage and a drain coupled to a first reference current source. The gate of the first n-channel transistor is coupled to the gate of a second n-channel transistor. The first and second n-channel transistors have first and second channel widths, respectively, wherein the second channel width is greater than the first channel width. A second reference current source is coupled to the drain of the second n-channel transistor. The drain of the second n-channel transistor provides an inhibit control signal for the charge pump control circuit. A p-channel transistor has a source coupled to the source of the second n-channel transistor, and a drain and gate coupled to receive the negative boosted voltage. The ratio of the first and second channel widths is selected such that the inhibit control signal is asserted when negative boosted voltage is greater than the V.sub.SS supply voltage minus the absolute value of a transistor threshold voltage V.sub.t. In one embodiment, the first reference current source has a negative temperature coefficient to compensate for temperature effects of the second n-channel transistor. The second reference current source can be provided with a positive temperature coefficient to compensate for temperature effects of the p-channel transistor.

The present invention will be more fully understood in view of the following description and drawings.

FIG. 1A is a schematic diagram of a conventional DRAM memory cell formed by p-channel MOS transistors fabricated using a conventional logic process.

FIG. 1B is a cross sectional diagram of the DRAM memory cell of FIG. 1A.

FIG. 2 is a schematic diagram of a conventional word line control circuit, including a word line driver and a word line voltage generator.

FIG. 3A is a schematic diagram of a DRAM memory cell that is supplied by voltage sources in accordance with one embodiment of the present invention.

FIG. 3B is a cross sectional view of the DRAM memory cell of FIG. 3A.

FIG. 4 is a schematic diagram of a word line driver in accordance with one embodiment of the present invention.

FIG. 5 is a block diagram illustrating a word line driver system that includes a first plurality of word line drivers, a second plurality of V.sub.SSB coupling circuits, a V.sub.CCB voltage generator and a V.sub.BBS voltage generator in accordance with one embodiment of the present invention.

FIG. 6 is a schematic diagram of a V.sub.SSB coupling circuit in accordance with one embodiment of the present invention.

FIG. 7 is a waveform diagram illustrating various signals generated during the operation of the V.sub.SSB coupling circuit of FIG. 6.

FIG. 8 is a block diagram of V.sub.CCB and V.sub.SSB boosted voltage generators in accordance with one embodiment of the present invention.

FIG. 9A is a simplified schematic diagram of a charge pump control circuit used in a conventional positive boosted voltage generator.

FIG. 9B is a simplified schematic diagram of a charge pump control circuit used in a conventional negative boosted voltage generator.

FIG. 10 is a schematic diagram of a V.sub.CCB charge pump control circuit in accordance with the one embodiment of the present invention.

FIG. 11 is a schematic diagram of a V.sub.BBS charge pump control circuit in accordance with the one embodiment of the present invention.

FIGS. 12-17 are schematic diagrams of reference current sources in accordance with various embodiments of the present invention.

FIG. 18 is a schematic diagram of a word line driver and a V.sub.BBC voltage coupling circuit in accordance with an embodiment of the present invention that uses NMOS transistors to form the DRAM cells.

The following describes the voltages and biasing of a DRAM memory fabricated using a conventional logic process which is a single or twin well process with a single polycrystalline silicon layer and one or more layers of metal. In the described examples, the positive supply voltage is designated as supply voltage V.sub.dd. In general, the positive supply voltage V.sub.dd can have a nominal value such as 3.3 Volts, 2.5 Volts, 1.8 Volts, etc., depending on the fabrication process. The ground supply voltage, having a nominal value of 0 Volts, is designated as supply voltage V.sub.SS.

As shown in FIG. 3A, a DRAM memory cell used in the described embodiments consists of a p-channel access transistor 301 and a p-channel storage transistor 302 that is configured as a storage capacitor. The gate of the access transistor 301 is connected to word line 303 and the drain of access transistor 301 is connected to bit line 305. The source of access transistor 301 is coupled to the source region of transistor 302. In the described embodiment, only the source region of transistor 302 is actually formed (i.e., there is no drain region of transistor 302). In another embodiment, both the source and drain regions are formed, and these regions are commonly connected to the source of access transistor 301. The channel of transistor 302 forms the electrode of the storage capacitor, and the gate of transistor 302 forms the counter-electrode of the storage capacitor. The channel of storage transistor 302 (i.e., the electrode of the storage capacitor) is coupled to the source of access transistor 301. The gate of transistor 302 (i.e., the counter-electrode of the storage capacitor) is connected to receive a negative boosted bias voltage V.sub.bb1. The bias voltage V.sub.bb1 is limited by the break-down voltage (V.sub.bd) of the gate oxide of capacitor 302 and the highest voltage (V.sub.1) stored on the electrode. In general, bias voltage V.sub.bb1 is set to a voltage that is greater than V.sub.1 minus V.sub.bd. In the preferred embodiment, V.sub.1 is equal to the positive supply voltage V.sub.dd, and bias voltage V.sub.bb1 is set to -0.3 Volts.

In general, the bias voltage V.sub.bb1 is selected to have a magnitude less than one diode voltage drop. That is, the bias voltage V.sub.bb1 is selected to have a magnitude less than about 0.7 Volts. The negative bias voltage V.sub.bb1 linearizes the operation of storage capacitor 302 by increasing the capacitance of capacitor 302 when the electrode is charged to the V.sub.dd supply voltage. Without the negative plate bias V.sub.bb1, the capacitance of capacitor 302 tends to decrease rapidly as the voltage across the capacitor becomes smaller than the threshold voltage of the MOS structure.

As illustrated in FIG. 3B, DRAM memory cell 300 is contained in an n-doped well 304 of a p-type monocrystalline silicon substrate 306. Multiple memory cells can share the same n-well 304. N-well 304 is biased to a boosted positive voltage (V.sub.pp1) that is greater than the V.sub.dd supply voltage by a voltage that is approximately equal to the absolute value of the threshold voltage (V.sub.tp) of p-channel access transistor 301. In addition, the boosted positive voltage V.sub.pp1 is selected to be lower than the oxide break down voltage of p-channel access transistor 301. N-well 304 is biased by a connection to n-type contact region 315. In the present embodiment, the V.sub.pp1 voltage is controlled to be approximately 0.3 Volts greater than the V.sub.dd supply voltage (i.e., V.sub.tp =0.3 Volts). Applying the V.sub.pp1 voltage to n-well 304 decreases the sub-threshold leakage of access transistor 301, and minimizes the possibility of forward biasing the junction between the electrode of capacitor 302 and n-well 304 due to supply noise. However, applying the V.sub.pp1 voltage to n-well 304 also increases the junction leakage at the electrode of storage capacitor 302, especially at higher voltages.

When data is written to memory cell 300, bit line 305 is coupled to the V.sub.dd supply voltage to write a logic zero data value, or to the V.sub.SS supply voltage to write a logic one data value. In addition, word line 303 is coupled to receive a word line voltage V.sub.SSB, which has a potential of about -0.3 Volts. Generation of the V.sub.SSB voltage is described in more detail below.

When memory cell 300 is in the data retention state, bit line 305 is pre-charged to a voltage of about one half the V.sub.dd supply voltage. Sub-threshold leakage of memory cell 300 tends to be higher when bit line 305 or the electrode of capacitor 302 is at a potential close to the V.sub.dd supply voltage. This sub-threshold leakage is more severe for sub-micron transistors because of their lower threshold voltages (e.g., V.sub.tp =-0.5 Volts). To reduce the sub-threshold leakage during the data retention state, word line 303 is coupled to an internally generated positive boosted voltage (V.sub.CCB) which has a potential about 0.3 Volts greater than the V.sub.dd supply voltage. This is different from the conventional memory cells described above, in which the word line is coupled to the V.sub.dd supply voltage during the data retention state. Generation of the positive boosted voltage V.sub.CCB is described in more detail below.

FIG. 4 is a schematic diagram of a word line driver 400 used to drive word line 303 in accordance with one embodiment of the present invention. Thus, the output voltages supplied by word line driver 400 are provided to word line 303 (FIG. 3A). Word line driver 400 consists of P-channel transistors 401-402 and N-channel transistors 403-405. To deactivate word line 303, transistor 401 is turned on, thereby pulling word line 303 up to the positive boosted word line voltage V.sub.CCB. The V.sub.CCB word line voltage is high enough to turn off access transistor 301. To activate word line 303, pull-down transistor 403 is turned on, thereby pulling down word line 303 to the V.sub.SSB voltage. The generation of the V.sub.SSB word line voltage is described in more detail below.

The gate of word line pull-up transistor 401 and the gate of word line pull-down transistor 403 are commonly connected to a pass gate formed by p-channel transistor 402. Transistor 402, when turned on, couples transistors 401 and 402 to receive an output signal X.sub.i provided by a row address decoder 410. The gate of transistor 402 is coupled to receive another output signal X.sub.j # from row address decoder 410. When the memory cells connected to word line 303 are selected for access, row address decoder 410 first drives the X.sub.i signal high, and then drives the X.sub.j # signal low. The low state of the X.sub.j # signal turns on pass transistor 402, which provides the logic high X.sub.i signal to the gates of the pull up and pull down transistors 401 and 403. Under these conditions, pull down transistor 403 is turned on, thereby coupling word line 303 to receive the V.sub.SSB word line voltage.

As described in more detail below, row address decoder 410 controls a first subset of word lines that includes word line 303 and a plurality of other word lines. If word line 303 is not selected for access (but another word line in the first subset of word lines is selected for access), then row address decoder 410 provides logic low values for both the X.sub.i and X.sub.j # signals. Under these conditions, the gates of pull up and pull down transistors 401 and 403 are maintained at logic low states by n-channel transistor 404. Note that the gate of transistor 404 is connected to word line 303, which is maintained at a logic high value when word line 303 is not being accessed. As a result, transistor 404 is turned on when word line 303 is not being accessed, thereby coupling the gates of transistors 401 and 403 to the V.sub.SS supply voltage. The V.sub.SS supply voltage turns on pull up transistor 401 and turns off pull down transistor 403, thereby maintaining a logic high voltage (i.e., V.sub.CCB) on word line 303.

During the data retention state (i.e., when none of the word lines in the first subset of word lines is being accessed) , row address decoder 410 drives the X.sub.j # signal high, thereby turning on n-channel transistor 405. Turned on transistor 405 couples the gates of pull up and pull down transistors 401 and 403 to the V.sub.SS supply voltage. As a result, pull up transistor 401 is turned on and pull down transistor 403 is turned off. At this time, transistor 401 couples word line 303 to receive the V.sub.CCB voltage, thereby turning off access transistor 301 of memory cell 300.

Pull down transistor 403 is selected to be an n-channel transistor to speed up the turn on of word line 303. However, in the present embodiment, the bulk of all n-channel transistors formed are connected to receive the V.sub.SS supply voltage. (See, FIG. 3B, which illustrates p-type substrate 306 coupled to receive the V.sub.SS supply voltage). As a result, the minimum value of the V.sub.SSB control voltage is limited to one diode voltage drop below the V.sub.SS supply voltage (i.e., one diode voltage drop below ground). Moreover, each row of memory cells has an associated word line driver. There are usually numerous rows of memory cells (e.g., more than 100) in an embedded memory. As a result of the large number of word line drivers, the reverse junction leakage between the substrate and the sources of the n-channel pull down transistors (such as pull down transistor 403) can be quite substantial. The reverse junction leakage increases exponentially as the V.sub.SSB control voltage becomes more negative. To limit the reverse junction leakage, the word line drivers are divided into groups of 32, with each group being coupled to a common V.sub.SSB coupling circuit 500.

FIG. 5 is a block diagram illustrating a word line driver system 500 that includes a first plurality of word line drivers 400, a second plurality of V.sub.SSB coupling circuits 600, a V.sub.CCB voltage generator 700 and a V.sub.BBS voltage generator 800. Each V.sub.SSB coupling circuit 600 is coupled to a corresponding group of 32 word line drivers 400. As described in more detail below, when one of the word lines in a group is to be turned on, the corresponding V.sub.SSB coupling circuit 600 is controlled to couple the V.sub.BBS voltage generator 800 to the corresponding group of 32 word line drivers. As a result, the V.sub.SSB coupling circuit routes the negative boosted voltage V.sub.BBS generated by the V.sub.BBS voltage generator 800 as the V.sub.SSB voltage. As described in more detail below, V.sub.BBS voltage generator 800 generates a V.sub.BBS voltage having a value less than one threshold voltage (V.sub.tp) below the V.sub.SS supply voltage. When none of the word lines in a group is to be turned on, the corresponding V.sub.SSB coupling circuit 500 is controlled to couple the V.sub.SS voltage supply to the corresponding group of 32 word line drivers. That is, the V.sub.SSB coupling circuit 500 routes the V.sub.SS supply voltage as the V.sub.SSB voltage.

Because only a subset of the word line drivers 400 is coupled to receive the V.sub.BBS voltage at any given time, the reverse junction leakage is substantially reduced. Moreover, by limiting the V.sub.BBS voltage to a voltage less than one threshold voltage below the V.sub.SS supply voltage, the reverse junction leakage is further reduced.

FIG. 6 is a schematic diagram of V.sub.SSB coupling circuit 600 in accordance with one embodiment of the present invention. V.sub.SSB coupling circuit 600 includes p-channel transistors 601-603, n-channel transistor 604 and inverters 611-614. P-channel transistor 601 is connected between the V.sub.SSB and V.sub.BBS voltage supply lines. The gate of transistor 601 is coupled to node N2. Transistor 602 is connected between node N2 and the V.sub.BBS voltage supply line. P-channel transistor 603 is connected as a capacitor, with its source and drain commonly connected to node N1, and its gate connected to node N2. N-channel transistor 604 is connected between the V.sub.SSB voltage supply line and the V.sub.SS voltage supply terminal. The gate of transistor 604 is connected to node N1. Inverters 611-614 are connected in series, with inverter 611 receiving the X.sub.j # signal from row address decoder 410, and inverter 614 providing the delayed X.sub.j # signal to node N1.

FIG. 7 is a waveform diagram illustrating various signals generated during the operation of V.sub.SSB coupling circuit 600.

Prior to activating word line 303, the X.sub.i signal is low and the X.sub.j # signal is high. Under these conditions, the chain of inverters 611-614 provides a logic high signal to node N1, thereby turning on n-channel transistor 604. As a result, the V.sub.SSB supply line is maintained at the V.sub.SS supply voltage (0 Volts). Also, prior to activating word line 303, the sub-threshold leakage of transistor 602 pulls node N2 to a voltage less than one threshold voltage drop (V.sub.t) above V.sub.BBS, thereby preventing transistor 601 from turning on.

As described above in connection with FIG. 4, the X.sub.i signal is driven high and then the X.sub.j # signal is driven low to activate word line 303. Under these conditions, pull down transistor 403 (FIG. 4) of word line driver 400 turns on, thereby coupling word line 303 to the V.sub.SSB supply line. Immediately after transistor 403 is turned on, the low state of the X.sub.j # is propagating through the chain of inverters 611-614 and has not reached node N1. During this time, n-channel transistor 604 remains on, coupling the V.sub.SSB supply line to receive the V.sub.SS supply voltage. Also during this time, the high state of node N1 pulls the source and drain of capacitor-coupled transistor 603 to a high state. Transistor 602 is connected as an MOS diode with its gate and drain connected to the V.sub.BBS supply line. Transistor 602 therefore limits the voltage at node N2 to no more than one threshold voltage (V.sub.t) above the V.sub.BBS voltage, or to a potential approximately equal to the V.sub.SS supply voltage. Consequently, capacitor 603 is initially charged to a voltage approximately equal to the V.sub.dd supply voltage (i.e., the voltage across transistor 603 is approximately equal to V.sub.dd).

When the low state of the X.sub.j # signal reaches node N1, transistor 604 is turned off, thereby de-coupling the V.sub.SSB voltage supply line from the V.sub.SS voltage supply terminal. The low voltage at node N1 also causes capacitor 603 to pull node N2 down to a voltage equal to -V.sub.dd. The -V.sub.dd voltage at node N2 turns on p-channel transistor 601, thereby coupling the V.sub.SSB voltage supply line to the V.sub.BBS voltage supply line. Note that only 32 word line drivers are coupled to the V.sub.BBS voltage supply line (and therefore the V.sub.BBS voltage generator 800) at this time. Because a relatively small number of word line drivers are connected to the V.sub.BBS supply line, the resulting junction leakage is relatively small.

The on-chip V.sub.BBS voltage generator 800 is designed to maintain V.sub.BBS at approximately -0.3 Volts below the V.sub.SS supply voltage despite the junction leakage. Note that during the activation of word line 303, this word line 303 is initially coupled to receive the V.sub.SS supply voltage. When the voltage of word line 303 drops below the V.sub.dd supply voltage, then word line 303 is coupled to receive the negative boosted voltage V.sub.BBS. This limits the source-to-drain voltage of word line pull down transistor 403 to be less than V.sub.CCB minus V.sub.BBS, thereby preventing transistor 403 from being exposed to high voltage stress.

To de-activate word line 303, the X.sub.j # signal is driven high by row address decoder 410. In response, pull up transistor 401 in word line driver 400 is turned on, thereby pulling up word line 303 to the V.sub.CCB voltage. In V.sub.SSB coupling circuit 600, the high state of the X.sub.j # signal propagates through the delay chain formed by inverters 611-614, thereby providing a high voltage at node N1 which turns on transistor 604. The high voltage at node N1 also couples node N2 to a voltage of about V.sub.SS, thereby turning off transistor 601. Under these conditions, the V.sub.SSB voltage supply line is coupled to the V.sub.SS voltage supply terminal.

The V.sub.CCB and V.sub.SSB voltages are generated by on-chip charge pump circuits in accordance with one embodiment of the present invention. FIG. 8 is a block diagram showing the general construction of the V.sub.CCB and V.sub.SSB boosted voltage generators 700 and 800 in accordance with one embodiment of the present invention. Each of the V.sub.CCB and V.sub.SSB boosted voltage generators consists of a ring oscillator 801, a charge pump 802 and a pump controller 803, which controls the operation of the oscillator 801 and thus charge pump 802. Ring oscillator 801 and charge pump 802 are conventional elements that are well documented in references such as U.S. Pat. Nos. 5,703,827 and 5,267,201.

FIG. 9A is a simplified schematic diagram of a charge pump control circuit 901 used in a conventional positive boosted voltage generator. Charge pump control circuit 901 includes a p-channel transistor 911 having a gate coupled to receive the V.sub.dd supply voltage, a source and bulk coupled to receive the positive boosted voltage V.sub.boost+, and a drain coupled to a reference current source 912. The drain of transistor 911 is also connected to the Inhibit control line. Current source 912 can be replaced with a resistor.

When the V.sub.boost+ voltage is higher than the V.sub.dd supply voltage by one threshold voltage (V.sub.tp), transistor 911 is turned on. The source current from transistor 911 is compared to the reference current I.sub.REF provided by current source 912. As the potential difference between the V.sub.boost+ and V.sub.dd voltages increases, the source current from transistor 911 increases. When the source current is larger than the reference current I.sub.REF, the Inhibit control line is coupled to receive the V.sub.boost+ voltage. The high state of the Inhibit signal disables the ring oscillator 801, thereby shutting down the charge pump 802 and stopping V.sub.boost+ from going higher. Depending on the magnitude of the reference current I.sub.REF, the boosted voltage V.sub.boost+ can be regulated at a voltage equal to the V.sub.dd supply voltage plus one threshold voltage (V.sub.tp) or higher. Note that the bulk of transistor 911 is coupled to receive the V.sub.boost+ voltage so that the source-to-bulk junction of this transistor is not forward biased. However, this connection is possible only when the bulk of transistor 911 is an N-well which can be isolated from the substrate, or when transistor 911 is formed in an n-type substrate that is biased to a voltage equal to or more positive than V.sub.boost+.

FIG. 9B is a simplified schematic diagram of a charge pump control circuit 902 used in a conventional negative boosted voltage generator. Charge pump control circuit 902 includes an n-channel transistor 921 having a gate coupled to receive the V.sub.SS supply voltage, a source and bulk coupled to receive the negative boosted voltage V.sub.boost-, and a drain coupled to a reference current source 922. The drain of transistor 921 is also connected to the Inhibit# control line. Current source 922 can be replaced with a resistor.

When the V.sub.boost- voltage is lower than the V.sub.SS supply voltage by one threshold voltage (V.sub.tn), transistor 921 is turned on. The drain current from transistor 921 is compared to the reference current I.sub.REF provided by current source 922. As the potential difference between V.sub.boost- and V.sub.SS increases, the drain current from transistor 921 increases. When the drain current is larger than the reference current I.sub.REF, the Inhibit# control line is coupled to receive the V.sub.boost- voltage. The low state of the Inhibit# signal disables the ring oscillator 801, thereby shutting down the charge pump 802 and stopping the V.sub.boost- voltage from going more negative. Depending on the magnitude of the reference current I.sub.REF, the V.sub.boost- voltage can be regulated at a voltage equal to V.sub.SS minus one threshold voltage (V.sub.tn) or more. Note that the bulk of transistor 921 is coupled to receive the V.sub.boost- voltage so that the source-to-bulk junction of this transistor is not forward biased. This connection is possible only when the bulk of transistor 921 is a p-well which can be isolated from the substrate, or when transistor 921 is formed in a p-type substrate that is biased a voltage equal to or more negative than V.sub.boost-.

Charge pump control circuits 901 and 902 cannot co-exist in a conventional logic process because such a process has the limitation that only one type of transistor can be isolated in a well. That is, both n-wells and p-wells are not available in a conventional logic process as defined herein. Moreover, because the p-type substrate of memory cell 300 is biased at the V.sub.SS voltage (FIG. 3B), the p-type substrate of memory cell 300 cannot be biased at a voltage equal to or more negative than the negative boosted word line voltage V.sub.BBS. Furthermore, because charge pump control circuit 901 results in a V.sub.boost+ voltage greater than or equal to V.sub.dd plus V.sub.tp, this charge pump control circuit 901 cannot generate a V.sub.boost+ voltage greater than the V.sub.dd supply voltage, but less than the V.sub.dd supply voltage plus the threshold voltage V.sub.tp as required by the present invention.

Similarly, because charge pump control circuit 902 results in a V.sub.boost- voltage less than or equal to the V.sub.SS supply voltage minus the threshold voltage V.sub.tn, this charge pump control circuit 902 cannot generate a V.sub.boost- voltage less than the V.sub.SS supply voltage, but greater than the V.sub.SS supply voltage minus the absolute value of the threshold voltage V.sub.tn as required by the present invention.

FIG. 10 is a schematic diagram of a V.sub.CCB charge pump control circuit 1000 in accordance with the one embodiment of the present invention. V.sub.CCB charge pump control circuit 1000 is used to replace charge pump control circuit 803 (FIG. 8), thereby creating a V.sub.CCB reference voltage generation circuit that is capable of generating the desired V.sub.CCB voltage. V.sub.CCB charge pump control circuit 1000 includes p-channel transistors 1001-1003 and reference current sources 1004-1005. The source of p-channel transistor 1001 is coupled to receive the V.sub.dd supply voltage, and the gate and drain of p-channel transistor 1001 are commonly connected to reference current source 1004. P-channel transistor 1001 is thereby connected as a diode between the V.sub.dd voltage supply and reference current source 1004. Reference current source 1004 generates a reference current, I.sub.REFP, which establishes a reference voltage, V.sub.REFP, on the gate of p-channel transistor 1002.

P-channel transistor 1001 has a channel width of W.sub.p. P-channel transistors 1001 and 1002 have the same channel lengths. However, p-channel transistor 1002 has a channel width of m times W.sub.p, where m is a multiplying constant. The drain of transistor 1002 is connected to another reference current source 1005, which generates a reference current, I.sub.REFP1. The source of transistor 1002 is connected to node V.sub.p. Node V.sub.p is also connected to the drain and gate of p-channel transistor 1003. The source of transistor 1003 is connected to receive the positive boosted voltage V.sub.CCB from charge pump 802. If the reference currents I.sub.REFP and I.sub.REFP1 are equal, and transistor 1002 has the same channel width as transistor 1001 (i.e., m=1), then node V.sub.p will be held at a voltage equal to the V.sub.dd supply voltage. Under these conditions, the positive boosted voltage V.sub.CCB will be higher than the V.sub.dd supply voltage by a voltage greater than the absolute value of the threshold voltage V.sub.tp of p-channel transistor 1003.

In the present embodiment, reference current I.sub.REFP is set approximately equal to reference current I.sub.REFP1, and the multiplying constant m is set equal to four. Because the channel length of transistor 1002 is four times longer than the channel length of transistor 1001, the source-to-gate voltage of transistor 1002 is less than the source-to-gate voltage of transistor 1001. As a result, the voltage on node V.sub.p is less than the V.sub.dd supply voltage. For example, if reference currents I.sub.REFP and I.sub.REFP1 are both set equal to about 50 .mu.A, then the voltage on node V.sub.p will be about 0.2 Volts less than the V.sub.dd supply voltage. The channel width of transistor 1003 is selected to be relatively large (e.g., on the order of 50 .mu.m) such that the source-to-gate voltage of transistor 1003 is approximately equal to the threshold voltage of transistor 1003 (e.g., 0.5 Volts). As a result, the V.sub.CCB voltage is maintained at a voltage about 0.3 Volts greater than the V.sub.dd supply voltage. The V.sub.CCB voltage is therefore less than one threshold voltage greater than the V.sub.dd supply voltage.

In another embodiment, p-channel transistor 1003 can be eliminated, such that the V.sub.CCB voltage is provided directly to node V.sub.p. However, in this embodiment, the channel width of transistor 1002 must be selected to smaller than the channel width W.sub.p of transistor 1001. That is, the multiplier constant m must be selected to be less than one, such that the source-to-gate voltage of transistor 1002 is greater than the source-to-gate voltage of transistor 1001 by about 0.3 Volts (or another voltage that is less than the p-channel threshold voltage).

FIG. 11 is a schematic diagram of a V.sub.BBS charge pump control circuit 1100 in accordance with the one embodiment of the present invention. V.sub.BBS charge pump control circuit 1100 is used to replace charge pump control circuit 803 (FIG. 8), thereby creating a V.sub.BBS reference voltage generation circuit that is capable of generating the desired V.sub.BBS voltage. V.sub.BBS charge pump control circuit 1100 includes n-channel transistors 1101-1102, p-channel transistor 1103 and reference current sources 1104-1105. The source of n-channel transistor 1101 is connected to receive the V.sub.SS supply voltage. The drain and gate of transistor 1101 are commonly connected to reference current source 1104. Thus, transistor 1101 is connected as a diode. Reference current source 1104 is connected between the V.sub.dd voltage supply and the commonly connected drain and gate drain of n-channel transistor 1101. Reference current source 1104 provides a reference current I.sub.REFN1 to n-channel transistor 1101. The reference current I.sub.REFN1 establishes a reference voltage, V.sub.REFN, on the gate of n-channel transistor 1102.

N-channel transistor 1101 has a channel width of W.sub.n. N-channel transistors 1101 and 1102 have the same channel lengths. However, n-channel transistor 1102 has a channel width of n times W.sub.n, where n is a multiplying constant. The drain of transistor 1102 is connected to another reference current source 1105, which generates a reference current, I.sub.REFN. The source of transistor 1102 is connected to node V.sub.N. Node V.sub.N is also connected to the source of p-channel transistor 1103. The drain and gate of transistor 1103 are commonly connected to receive the negative boosted voltage V.sub.BBS. If the reference currents I.sub.REFN and I.sub.REFN1 are equal, and transistor 1102 has the same channel width as transistor 1101 (i.e., n=1), then node V.sub.N will be held at a voltage equal to the V.sub.SS supply voltage. Under these conditions, the negative boosted voltage V.sub.BBS will be regulated at a voltage approximately one threshold voltage (V.sub.tp) below the V.sub.SS supply voltage.

In the present embodiment, reference current I.sub.REFN is set approximately equal to reference current I.sub.REFN1, and the multiplying constant n is set equal to four. Because the channel width of transistor 1102 is four times longer than the channel width of transistor 1101, the source-to-gate voltage of transistor 1102 is less than the source-to-gate voltage of transistor 1101. As a result, the voltage potential on node V.sub.N is higher than the V.sub.SS supply voltage. For example, if reference currents I.sub.REFN and I.sub.REFN1 are both set equal to about 50 .mu.A, then the voltage on node V.sub.N will be about 0.2 Volts greater than the V.sub.SS supply voltage. The channel width of transistor 1103 is selected to be relatively large (e.g., on the order of 50 .mu.m) such that the source-to-gate voltage of transistor 1103 is approximately equal to the threshold voltage of transistor 1103 (e.g., 0.5 Volts). As a result, the V.sub.BBS voltage is maintained at a voltage about 0.3 Volts less than the V.sub.SS supply voltage. The V.sub.BBS voltage is therefore less than one threshold voltage than the V.sub.SS supply voltage.

In another embodiment, p-channel transistor 1103 can be eliminated, such that the V.sub.BBS voltage is provided directly to node V.sub.N. However, in this embodiment, the channel width of transistor 1102 must be selected to smaller than the channel width W.sub.n of transistor 1101. That is, the multiplier constant n must be selected to be less than one, such that the source-to-gate voltage of transistor 1102 is greater than the source-to-gate voltage of transistor 1101 by about 0.3 Volts (or another voltage that is less than the p-channel threshold voltage).

It is desirable to keep the V.sub.CCB and V.sub.BBS voltages relatively constant for variations in temperature. In general, the transistor threshold voltage V.sub.t tends to decrease as the temperature increases. To compensate for this temperature effect, reference current sources 1004 and 1104 are constructed such that reference currents I.sub.REFP and I.sub.REFN1 have negative temperature coefficients (i.e., reference currents I.sub.REFP and I.sub.REFN1 decrease as the temperature increases).

FIG. 12 is a schematic diagram of reference current source 1004 in accordance with one embodiment of the present invention. Reference current source 1004 includes p-channel transistors 1201-1202, resistor 1203 and n-channel transistors 1204-1206. Resistor 1203 is connected between the V.sub.dd voltage supply and the gate of transistor 1201, thereby setting the bias for transistor 1201. The current I.sub.R through resistor 1203 is equal to the threshold voltage V.sub.tp of transistor 1201 divided by the resistance of resistor 1203. The current I.sub.R is therefore directly related to the threshold voltage V.sub.tp. The current I.sub.R flows through p-channel transistor 1202 and n-channel transistor 1205.

The gate and source of transistor 1202 are coupled to the drain and gate, respectively, of transistor 1201. The voltage on the gate of transistor 1202 is translated to the drain of transistor 1202. N-channel transistors 1204-1206 each have a source terminal coupled to the V.sub.SS voltage supply and a gate terminal coupled to the drain of transistor 1202, thereby forming a current mirror circuit. The current I.sub.R is thereby translated to transistor 1206. As a result, the current through n-channel transistor 1206 (i.e., I.sub.REFP) is directly related to the threshold voltage V.sub.tp of p-channel transistor 1201.

Reference current source 1004 provides temperature compensation as follows.

As the temperature increases, the threshold voltages V.sub.tp of transistors 1002 and 1003 (FIG. 10) decrease, thereby causing the V.sub.CCB voltage to decrease. However, as the temperature increases, the threshold voltage V.sub.tp of transistor 1201 (FIG. 12) decreases. In response, the current I.sub.R decreases, thereby reducing the I.sub.REFP current. As a result, the gate-to-source voltage of p-channel transistor 1001 (FIG. 10) decreases, thereby increasing the V.sub.REFP voltage. The increased V.sub.REFP voltage, in turn, causes the voltage V.sub.p to increase, thereby increasing the V.sub.CCB voltage. The temperature effect of the threshold voltage V.sub.tp of transistors 1002 and 1003 is thereby partially compensated by the negative temperature coefficient of the I.sub.REFP current. In this manner, reference current source 1004 provides temperature compensation to V.sub.CCB pump control circuit 1000.

FIG. 13 is a schematic diagram of reference current source 1104 in accordance with one embodiment of the present invention. Because reference current source 1104 is similar to reference current source 1004 (FIG. 12), similar elements in FIGS. 12 and 13 are labeled with similar reference numbers. Thus, reference current source 1104 includes p-channel transistors 1201-1202, resistor 1203 and n-channel transistors 1204-1205. In addition, reference current source 1104 includes a p-channel transistor 1301 having a gate coupled to the gate of transistor 1201, and a source coupled to receive the V.sub.dd supply voltage.

Reference current source 1104 provides temperature compensation as follows.

As the temperature increases, the threshold voltages V.sub.t of transistors 1102 and 1103 (FIG. 11) decrease, thereby causing the V.sub.BBS voltage to increase. However, as the temperature increases, the threshold voltage V.sub.tp of p-channel transistor 1201 decreases. As a result, the current I.sub.R decreases. Because transistors 1201 and 1301 are coupled to form a current mirror circuit, the decrease in the current I.sub.R results in a decrease in the current I.sub.REFN1. A decrease in the current I.sub.REFN1, in turn, causes a decrease in the voltage V.sub.REFN (FIG. 11). The decrease in V.sub.REFN results in a decrease of the voltage V.sub.N, which in turn, causes a decrease in the V.sub.BBS voltage. In this manner, reference current source 1104 provides temperature compensation to V.sub.BBS pump control circuit 1100.

If the I.sub.REFP1 current is temperature independent, then reference current source 1004 (FIG. 12) mainly compensates for the temperature effect of transistor 1002, thereby leaving the temperature effect of transistor 1003 largely uncompensated. Similarly, if the I.sub.REFN current is temperature independent, then reference current source 1104 (FIG. 13) mainly compensates for the temperature effect of transistor 1102, leaving the temperature effect of transistor 1103 largely uncompensated. To compensate for the uncompensated temperature effects of transistors 1003 and 1103, reference current sources 1005 and 1105 are constructed such that reference currents I.sub.REFP1 and I.sub.REFN have positive temperature coefficients (i.e., reference currents I.sub.REFP1 and I.sub.REFN increase as the temperature increases).

FIG. 14 is a schematic diagram of reference current source 1005 in accordance with one embodiment of the present invention. Reference current source 1005 includes p-channel transistors 1401-1403, n-channel transistors 1411-1414, PNP bipolar transistors 1421-1422 and resistor 1431. Transistors 1401, 1411 and 1421 are connected in series between the V.sub.dd and V.sub.SS voltage supplies. Transistors 1402, 1412 and 1422 and resistor 1431 are connected in series between the V.sub.dd and V.sub.SS voltage supplies. Transistor 1403 is connected in series with parallel-connected transistors 1413-1414 between the V.sub.dd and V.sub.SS voltage supplies. P-channel transistors 1401-1403 are configured to form a current mirror circuit, such that the same current flows through all three of these transistors 1401-1403. The emitter of transistor 1422 is selected to be m times larger than the emitter of transistor 1421, where m is a multiplying constant. In the described embodiment, the multiplying constant m is equal to 4. The multiplying constant m and the resistor value of resistor 1431 is selected such that the resultant current I.sub.REFP1 is approximately equal to I.sub.REFP. The voltages at the sources of transistors 1411 and 1412 are maintained at the same voltage by transistors 1401-1402 and 1411-1412. As a result, the voltage across transistor 1421 is equal to the voltage across resistor 1431 and transistor 1422.

The operation of reference current source 1005 is well documented in references such as "Analysis and Design of Analog Integrated Circuits", by P. R. Gray and R. G. Meyer, pp. 330-333, which is hereby incorporated by reference. The current I.sub.R through resistor 1431 is equal to V.sub.T /R ln(m). V.sub.T =kT/q, where k is equal to Boltzmann's constant, T is equal to absolute temperature, and q is equal to electron charge. The current through resistor 1431 is therefore directly related to temperature. The current I.sub.R through resistor 1431 is translated to create the I.sub.REFP1 current through transistors 1403 and 1413-1414. As a result, the I.sub.REFP1 current is directly related to temperature. Thus, as temperature increases, the I.sub.REFP1 current increases. The increased I.sub.REFP1 current increases the gate-to-source voltages of transistors 1002 and 1003 in FIG. 10, thereby offsetting the decrease in the threshold voltage V.sub.tp of transistor 1003 which occurs with increases in temperature. As described above, the decrease of the threshold voltage V.sub.tp of transistor 1003 tends to decrease the V.sub.CCB voltage. However, the increased I.sub.REFP1 current tends to increase the V.sub.CCB voltage. The net result is that the V.sub.CCB voltage is maintained relatively constant throughout the operating temperature range.

FIG. 15 is a schematic diagram of reference current source 1105 in accordance with one embodiment of the present invention. Reference current source 1105 includes p-channel transistors 1401-1402 and 1501, n-channel transistors 1411-1412, PNP bipolar transistors 1421-1422 and resistor 1431. Transistors 1401-1402, 1411-1412, 1421-1422 and resistor 1432 are connected in the manner described above for FIG. 14. In addition, the gate of transistor 1501 is commonly connected to the gates of transistors 1401-1402. As described above, the current I.sub.R through resistor 1431 is directly related to temperature. Thus, as the temperature increases, the I.sub.R current through resistor 1431 increases. This increased current is translated to transistor 1501, thereby resulting in an increased I.sub.REFN current. The increased I.sub.REFN current increases the gate-to-source voltages of transistors 1102 and 1103 in FIG. 11, thereby offsetting the decrease in threshold voltage V.sub.tp of transistor 1103 in FIG. 11. As described above, the decrease in the threshold voltage V.sub.tp of transistor 1103 tends to increase the V.sub.BBS voltage. However, the increased I.sub.REFN current tends to decrease the V.sub.BBS voltage. The result is that the V.sub.BBS voltage is maintained relatively constant in the operating temperature range of the reference current circuit 1104.

FIG. 16 is a schematic diagram illustrating reference current circuit 1600 in accordance with another embodiment of the present invention. Reference current circuit 1600 combines reference current circuits 1004 and 1104 in a single circuit, thereby reducing the required layout area of the resulting circuit. Similar elements in FIGS. 12, 13 and 16 are labeled with similar reference numbers. Reference current circuit 1600 operates in the same manner as reference current circuits 1104 and 1104.

FIG. 17 is a schematic diagram illustrating reference current circuit 1700 in accordance with another embodiment of the present invention. Reference current circuit 1700 combines reference current circuits 1005 and 1105 in a single circuit, thereby reducing the required layout area of the resulting circuit. Similar elements in FIGS. 14, 15 and 17 are labeled with similar reference numbers. Reference current circuit 1700 operates in the manner as reference current circuits 1005 and 1105.

The preferred embodiment described above uses PMOS transistors for the memory cells. The p-channel transistors are fabricated in N-well on P-substrate. In another embodiment, the memory cells can be fabricated using NMOS transistors. In such an embodiment, the word line is activated high and deactivated low.

FIG. 18 is a schematic diagram illustrating word line driver circuit 1600 and a V.sub.BBC coupling circuit 1800 that can be used to drive memory cells constructed from NMOS transistors. Word line driver circuit 1600 includes p-channel pull-up transistor 401 and n-channel pull-down transistor 403, which were described above in connection with word line driver 400 (FIG. 4). The remainder of word line driver 1600 is a reciprocal circuit of word line driver 400. The reciprocal circuit is obtained by replacing PMOS transistors NMOS transistors, replacing NMOS transistors with PMOS transistors, replacing connections to the V.sub.dd voltage supply with connections to the V.sub.SS voltage supply, and replacing connections to the V.sub.SS voltage supply with connections to the V.sub.dd voltage supply. Thus, in addition to pull up and pull down transistors 401 and 403, word line driver 1600 includes n-channel transistor 1601, p-channel transistors 1602-1603 and row address decoder 1610.

N-channel pull-down transistor 403 of word line driver 400 is coupled directly to V.sub.BBS voltage generator 800. In this embodiment, the V.sub.BBS voltage generator provides a V.sub.BBS voltage about -0.3 V below the V.sub.SS supply voltage. The p-channel pull-up transistor 401 of word line driver 400 is coupled to receive a V.sub.BBC voltage from V.sub.BBC coupling circuit 1800. Row address decoder 1610 provides control signals X.sub.i # and X.sub.j, which are the inverse of the control signals X.sub.i and X.sub.j # provided by row address decoder 410 (FIG. 4).

V.sub.BBC coupling circuit 1800 is the reciprocal the coupling circuit 600 of FIG. 6. Thus, V.sub.BBC coupling circuit 1800 includes n-channel transistors 1801-1803, p-channel transistor 1804 and inverters 1811-1814, as illustrated.

Prior to activating word line 303, the X.sub.i # signal is high and the X.sub.j signal is low. Under these conditions, transistor 1602 is turned on, thereby applying the Vdd supply voltage to the gates of transistors 401 and 403. As a result, pull-down transistor 403 turns on, thereby providing the VBBS voltage to word line 303. Also under these conditions, the chain of inverters 1811-1814 provides a logic low signal to node N1, thereby turning on p-channel transistor 1804. As a result, the V.sub.BBC supply line is maintained at the V.sub.dd supply voltage. Also, prior to activating word line 303, the sub-threshold leakage of transistor 1802 pulls node N2 to a voltage greater than one threshold voltage drop (V.sub.t) below V.sub.CCB, thereby preventing transistor 1801 from turning on.

The X.sub.i # signal is driven low and then the X.sub.j signal is driven high to activate word line 303. Under these conditions, pull up transistor 401 turns on, thereby coupling word line 303 to the V.sub.BBC voltage coupling circuit 1800. Immediately after transistor 401 is turned on, the high state of the X.sub.j signal is propagating through the chain of inverters 1811-1814 and has not reached node N1. During this time, p-channel transistor 1804 remains on, coupling the V.sub.BBC supply line to receive the V.sub.dd supply voltage. Also during this time, the low state of node N1 pulls the source and drain of capacitor-coupled transistor 1803 to a low state. Transistor 1802 is connected as an MOS diode with its gate and drain connected to the V.sub.CCB supply line. Transistor 1802 therefore limits the voltage at node N2 to no more than one threshold voltage (V.sub.t) below the V.sub.CCB voltage, or to a potential approximately equal to the V.sub.dd supply voltage. Consequently, capacitor 1803 is initially charged to a voltage approximately equal to the V.sub.dd supply voltage (i.e., the voltage across transistor 1803 is approximately equal to V.sub.dd)

When the high state of the X.sub.j signal reaches node N1, transistor 1804 is turned off, thereby de-coupling the V.sub.BBC voltage supply line from the V.sub.dd voltage supply terminal. The high voltage at node N1 also causes capacitor 1803 to pull node N2 up to a voltage equal to 2V.sub.dd. The 2V.sub.dd voltage at node N2 turns on n-channel transistor 1801, thereby coupling the V.sub.CCB voltage supply line to the V.sub.BBC voltage supply line.

Although the invention has been described in connection with several embodiments, it is understood that this invention is not limited to the embodiments disclosed, but is capable of various modifications which would be apparent to a person skilled in the art. Thus, the invention is limited only by the following claims.