Precision Temperature Sensor Integrated Circuit

1. Field of the Invention

This invention relates generally to integrated circuits which measure ambient temperature. Specifically, the present invention is a precision temperature sensor that produces an approximately linear capacitor charging current, which in turn is used to produce a clock frequency that is predictable with variances in ambient temperature. The invention incorporates a relaxation oscillator, two independent current generators, a reference oscillator and a frequency counter to compute ambient temperature. The invention is implemented on a single, monolithic integrated circuit.

2. Description of the Prior Art

The prior art is described by FIG. 1 which shows a typical integrated temperature sensor based on the well known .DELTA.V.sub.BE model that generates a voltage proportional to temperature. The prior art integrated circuit temperature sensors require a differential amplifier and an analog to digital (A-to-D) converter to convert a voltage from a diode, thermistor or other source into a digital temperature equivalent. The traditional, voltage oriented A-to-D converter may also require unnecessary hardware and software overhead not needed for temperature sensing applications. Furthermore, the prior art circuit requires a precision reference voltage which can be very costly depending on the requirements for accuracy.

Therefore, a need existed to provide a temperature sensor implemented on an integrated circuit which is capable of accurate measurement of temperature variations and which reduces complexity, board space and/or die area and pin count.

It is an object of the present invention to provide a temperature sensor that does not rely on a voltage oriented A-to-D converter, but instead incorporates a current to frequency A-to-D circuit.

It is another object of the present invention to provide a temperature sensor which does not require a precise voltage reference.

It is another object of the present invention to provide a temperature sensor with reduced cost to manufacture, die area and pin count.

In accordance with one embodiment of the present invention, a temperature sensor that produces an accurate measurement over wide variations of ambient temperature is disclosed. The precision relaxation oscillator is comprised of an oscillation generator, a first current generator for producing a first output current, a second current generator for producing a second output current, a reference oscillator and a frequency counter. The invention is implemented on a single, monolithic integrated circuit.

In accordance with another embodiment of the present invention the reference oscillator is comprised of a second oscillation generator.

In accordance with another embodiment of the present invention the reference oscillator is comprised of a crystal oscillator.

In accordance with another embodiment of the present invention one or more external resistors may be coupled to either the first or second current generators to produce the respective output currents.

The foregoing and other objects, features, and advantages of the invention will be apparent from the following, more particular, description of the preferred embodiments of the invention, as illustrated in the accompanying drawings.

FIG. 1 is a schematic diagram of the prior art showing a temperature sensor circuit.

FIG. 2 is a block diagram of the present invention.

FIG. 2A is a graph depicting the linear characteristics and the relationships between the current parameters of the present invention.

FIG. 3 is a block diagram of the Proportional to Absolute Temperature (PTAT) current generator found in the present invention.

FIG. 4 is a block diagram of the Complementary to Absolute Temperature (CTAT) current generator found in the present invention.

FIG. 5 is a block diagram of the moderate precision temperature sensor embodiment of the present invention.

FIG. 6 is a timing diagram of specific parameters of the present invention.

Referring to FIG. 2, a precision temperature sensor 1 that produces an accurate measurement over wide variations of ambient temperature is shown.

The precision temperature sensor 1 is comprised of an oscillation generator 100, a first current generator 300 which is typically a Proportional to Absolute Temperature (PTAT) current generator, a second current generator 200 which is typically a Complementary to Absolute Temperature (CTAT) current generator, a reference oscillator 400 and a frequency counter 500. However, those skilled in the art will recognize that the roles for the current generator may be reversed, i.e. the first current generator 200 may be the CTAT and the second current generator 300 may be the PTAT. In the preferred embodiment of the present invention, the precision temperature sensor 1 is implemented on a single, monolithic integrated circuit.

Referring to FIGS. 2 and 2A, the independent CTAT 200 and PTAT 300 current generators provide a CTAT current 290 and a PTAT current 390 which are approximately linear with respect to temperature. In the preferred embodiment, the CTAT current 290 and PTAT current 390 (FIGS. 4 and 5) are combined to form a capacitor charging current (I.sub.CCC) 190 (I.sub.CCC 190=CTAT current 290+PTAT current 390) which has a constant, positive slope with respect to temperature.

In the preferred embodiment the PTAT current 390 is the dominant component of I.sub.CCC 190 and has a constant, positive slope directly proportional to temperature, but which may vary slightly from device to device because of fabrication anomalies or process drift. The CTAT current 290 has a slope, which is proportional to temperature and is combined with the PTAT current 390 so that a predictable, uniform I.sub.CCC 190, which varies approximately linearly with temperature, is achieved from device to device. In the preferred embodiment CTAT current has a negative slope with respect to temperature. In other embodiments, the PTAT current 390 may be the sole component of I.sub.CCC 190 and may be digitally trimmed as described below. In still other embodiments, the CTAT current 290 may be the sole component of I.sub.CCC 190 and may also be digitally trimmed. I.sub.CCC 190 is subsequently used by the oscillation generator 100 (FIG. 2) to produce a predictable clock frequency that varies with temperature.

In the preferred embodiment as illustrated in FIG. 2, the oscillation generator 100 is comprised of a set-reset flip-flop 160, a comparator circuit 180 further comprised of two comparators 182 and 184, two capacitors 110 and 120, four transistor switches 130, 132, 134 and 136, two inverters 140 and 142 and a bandgap reference voltage circuit 150 for producing a reference voltage 152. Those skilled in the art will recognize that other combinations of these components or similar components are possible to achieve substantially the same results.

The transistor switches 130 and 134 provide charging paths for the capacitors 110 and 120, respectively. The transistor switches 132 and 136 provide discharging paths for the capacitors 110 and 120, respectively. In the preferred embodiment, the transistor switches 130, 132, 134 and 136 are MOSFET transistors, however, those skilled in the art will recognize that the invention is not limited to this technology.

The oscillation generator 100 operates by having one capacitor charge while the other capacitor discharges. The discharge path for the capacitor 110 is connected via transistor switch 132 to an input of the comparator 182. The discharge path for the capacitor 120 is connected via transistor switch 136 to an input of the comparator 184.

In the preferred embodiment, and for best performance, a stable reference voltage source such as a bandgap reference voltage circuit 150 is used. The bandgap reference voltage circuit 150 provides a single reference voltage 152, which is connected to second inputs of comparators 182 and 184, and is used to set the common mode voltage at each comparator 182 and 184. The P.sub.BIAS input 325 for the bandgap reference voltage circuit 150 is an output of the PTAT bias generator 310 which is described below. The bandgap reference voltage circuit 150 has the advantage of stabilizing I.sub.CCC 190 and minimizing the error due to variance in comparator input slew and propagation delay. Furthermore, in order to cancel the effects of reference voltage drift, the CTAT 200 current generator relies on the same reference voltage 152 as the comparators 182 and 184. There are various embodiments of the bandgap reference voltage circuit 150 which are well known to those skilled in the art. However, the novel way in which the bandgap reference voltage circuit 150 is implemented in the present invention is not disclosed by the prior art.

The output of comparator 182 is connected to the set input 162 of the flip-flop 160. The output of comparator 184 is connected to the reset input 164 of the flip-flop 160. Thus, as the capacitors 110 and 120 alternatively charge and discharge, the outputs of the comparators 182 and 184 alternatively set and reset the flip-flop 160 thus producing a clock output.

The Q output 166 of the flip-flop 160 provides a clock frequency that is directly related to temperature variation. In the preferred embodiment, the Q output 166 is also coupled to transistor switch 132 and via inverter 140 to transistor switch 130. Thus, the Q output 166 provides the signal that controls the transistor switches 130 and 132, which in turn open and close the charging and discharging paths for capacitor 110.

The complementary Q output 168 of flip-flop 160 provides a second clock frequency that is also directly related to temperature and the complement of Q output 166. The complementary Q output 168 is coupled to transistor switch 136 and via inverter 142 to transistor switch 134. Thus, the complementary Q output 168 provides the signal that controls the transistor switches 134 and 136 which in turn open and close the charging and discharging paths for capacitor 120.

The reference oscillator 400, has a temperature coefficient much smaller than that of the current generators 200 and 300, thereby providing a stable clock frequency over wide variations in temperature. The reference oscillator 400 and the Q output 166 of flip-flop 160 are coupled to the frequency counter 500. For moderate precision, the reference oscillator 400 may be a second oscillation generator 100 as described below.

For higher precision, a crystal oscillator is the preferred embodiment because it has high stability over temperature. The frequency counter 500 compares the temperature sensitive Q output 166 with the temperature insensitive reference oscillator 400 and provides an output 510 that is an accurate representation of temperature. Frequency counters of various embodiments are well know to those skilled in the art and therefore need not be described further.

Referring to FIG. 3, wherein like numerals reflect like elements, the PTAT current generator 300, known to those skilled in the art as a .DELTA.V.sub.BE circuit, is comprised of PTAT bias generator 310 and a PTAT current mirror 350 for producing a PTAT current 390.

The PTAT bias generator 310 is comprised of an amplifier circuit 320, a first bias circuit 330 for producing a first bias voltage across a selectable resistor with a small linear temperature coefficient 332 or 334 and a second bias circuit 340 for producing a second bias voltage. The present invention provides that an external resistor 334 may be selected over the internal resistor 332 by a select bit 336.

The first and second bias voltages provide the inputs to the amplifier 320. The output of the amplifier 320 is P.sub.BIAS 325 which is coupled to the first 330 and second 340 bias circuits, the PTAT current mirror 350 and the bandgap reference voltage generator 150 (FIG. 1).

The PTAT current mirror 350 is comprised of a plurality of transistors 352 from one to n. As shown in FIG. 3, the control electrodes of the transistors 352 are coupled to P.sub.BIAS 325. Trimming for the proper output current is performed digitally by programmable selecting one or more of the current mirror transistors 352 via the calibration select 354 to obtain the desired PTAT current 390. In the preferred embodiment, the calibration select switches 354 are PMOS transistors. However, those skilled in the art will recognize that other types of switches are possible which result in substantially the same result.

In the preferred embodiment, the current mirror 350 acts as a current divider which is well known to those skilled in the art. In other embodiments the current mirror 350 may be configured as a current multiplier. The PTAT current 390 is the sum of the selected outputs from the current mirror transistors 352. Thus, the PTAT current generator 300 permits determination of the slope and intercept of the approximately linear PTAT current 390 with respect to temperature as shown in FIG. 2A.

Referring to FIG. 4, wherein like numerals reflect like elements, the CTAT current generator 200 is comprised of a CTAT bias generator 210 and a current mirror 250 for producing the CTAT current 290.

The CTAT bias generator 210 is comprised of an amplifier circuit 220, at least one resistor with a small positive temperature coefficient 232 and 234 for regulating the input current to the amplifier and a transistor 240 for providing an input current to the amplifier 220. The amplifier 220 is a cascode configuration for supply and noise rejection. The reference voltage 152 is coupled to an input of the amplifier 220. The present invention provides that an external resistor 234 may be selected over the internal resistor 232 by a select bit 236.

The current mirror 250 is comprised of a plurality of transistors 252 from one to n. As shown in FIG. 4, the control electrodes of the transistors 252 are coupled to the output of the CTAT bias generator amplifier 220. Trimming the CTAT current 290 for achieving the proper PTAT:CTAT balance is performed digitally by programmable selecting one or more of the current mirror transistors 252 via the calibration select 254 to obtain the desired CTAT current 290. In the preferred embodiment, the calibration select switches 254 are PMOS transistors. However, those skilled in the art will recognize that other types of switches are possible which result in substantially the same result.

Thus, trimming is the process of controlling the output currents 290 and 390 of the current mirrors 250 and 350, respectively, with respect to temperature. When the trimmed output currents 290 and 390 are combined at the oscillation generator 100 (FIG. 2), the proper slope and intercept (current versus temperature) for I.sub.CCC 190 is achieved for the purpose of producing a clock frequency 166 (FIG. 2) that predictably varies with temperature. The approximately linear I.sub.CCC 190 produces an approximately linear clock frequency 166 with respect to temperature. Therefore, it follows that the trimmed output currents 290 and 390 are combined as I.sub.CCC 190 to control the proper slope and intercept of clock frequency 166 over temperature as illustrated in FIG. 2A.

In the preferred embodiment, the current mirror 250 acts as a current divider which is well known to those skilled in the art. In other embodiments the current mirror 250 may be configured as a current multiplier. The CTAT current 290 is the sum of the selected outputs from the current mirror transistors 252.

Referring to FIG. 5, wherein like numerals reflect like elements and numbers with primes (') represent additional circuitry of like characteristics, the moderate precision temperature sensor embodiment is shown. In this embodiment a common PTAT bias generator 310 and CTAT bias generator 210 each drive a unique pair of PTAT current mirrors 350 and 350' and CTAT current mirrors 250 and 250', respectively. The PTAT bias generator 310 also drives the reference voltage generator 150, as previously described. The PTAT/CTAT current mirrors 350 and 250 produce temperature sensitive output currents 390 and 290. These PTAT/CTAT currents 390 and 290 drive the oscillation generator 100 to produce a clock frequency that varies proportionally with temperatures as described above.

The PTAT/CTAT current mirrors 350' and 250' are very similar to the PTAT/CTAT current mirrors 350 and 250, except that PTAT/CTAT current mirrors 350' and 250' produce temperature insensitive output currents 390' and 290' by controlling the ratio of PTAT/CTAT current 390' and 290' so that the slope of the combined current is insensitive with respect to temperature. These PTAT/CTAT currents 390' and 290' drive the oscillation generator 100', which is structurally similar to oscillation generator 100. The output of oscillation generator 100' is a clock frequency which does not vary, or varies very little, with respect to temperature. The frequency counter 500 compares the two clock frequencies from the oscillation generators 100 and 100' to produce a digital temperature measurement, as described above.

Referring to FIG. 6, wherein like numerals reflect like elements, a timing diagram for the temperature sensor 1 is shown. V1 112 reflects the charging and discharging of capacitor 110 (FIG. 1). Note that the positive slope (charging) of V1 112 is equal to I.sub.CCC 190 divided by the capacitance of capacitor 110. The maximum amplitude of V1 112 is equal to the reference voltage 152. CMP1 reflects the output of the comparator 182 which is coupled to the set input 162 of the flip-flop 160.

V2 122 reflects the charging and discharging of capacitor 120. In this case the positive slope of V2 122 is equal to I.sub.CCC 190 divided by the capacitance of capacitor 120. CMP2 reflects the output of the comparator 184 which is coupled to the reset input 164 of the flip-flop 160. CLK is the Q output 166 of the flip-flop 160.

For a 50 percent duty cycle, the values of capacitors 110 and 120 are identical which result in similar slopes for V1 112 and V2 122. As the capacitor voltage exceeds the reference voltage 152, the respective comparator 182 and 184 pulses low which causes the flip-flop 160 to change state. RST (reset) is used to initialize the comparators 182 and 184 and the flip-flop 160 to a known state and to operate the sensor in a one shot mode.

The present invention produces a clock frequency which is directly proportional to temperature. This is accomplished by providing bias currents which when trimmed and summed yield a predictable capacitor charging current that is approximately linearly proportional to temperature. The circuit 1 performs digital trimming of the bias currents 290 and 390 via the programmable current mirrors 250 and 350 to eliminate process variations, using a stable voltage reference such as a bandgap reference voltage circuit 150. The circuit 1 produces a temperature sensitive clock 166 via a dual capacitor, dual comparator oscillation generator 100. The frequency counter 500 compares the output of the oscillation generator 100 to the output of a reference oscillator to compute a digital temperature 510. Also, analog design techniques, well known to those skilled in the art, such as component matching and cascode current sources enhance the stability of the circuit.

Although the invention has been particularly shown and described with reference to a preferred embodiment thereof, it will be understood by those skilled in the art that changes in form and detail may be made therein without departing from the spirit and scope of the invention.