Spread Spectrum Chip Rate Tracking System

The field of this invention relates to spread spectrum techniques which have been used for many years to enhance communication effectiveness in signal jamming environments or to hide communications from detection by hostile intercept receivers. More particularly, this invention relates to spread spectrum receivers in a direct and nondirect communication system such as in repeater or satellite communications environments.

As intercept receiver technology advances, the pseudo random characteristics of the modulating signal used to "spread" the communication signal must increase in complexity to counter the detection threat. The capability to greatly increase the code complexity without affecting the correlation of time was the subject of a patent application still pending entitled, "Spread Spectrum Communications System," filed as U.S. patent application Ser. No. 07/795,165 on Nov. 18, 1991, now U.S. Pat. No. 5,677,928, which has been classified. The correlation approach used in that invention created a programmable digital matched filter and associated circuitry to retard the phase of a locally-generated, direct-sequence spread spectrum signal in relation to an incoming communication signal. This and other design approaches rely on transmitting and receiving systems being in direct contact, with complete control of the frequency of both communicating nodes. This technique does not allow communications where frequency transponder or repeating node techniques are required since the frequency aging drift or other uncertainties of that node are not directly controlled by one of the communicating devices. Accordingly, a degradation of the incoming signal results because the incoming detection is based on a synchronized frequency system. The instant invention provides a communications technique that allows communication in direct and nondirect cases without degrading the correlation time or complexity of the spreading signal. In this spread spectrum system, the transmitted communication signal is created using a direct sequence spread spectrum technique that spreads the power transmitted over a large frequency range and therefore the identifying signal is at or below the RF noise floor. The resulting signal is undetectable to any receiver, other than one employing the specific spreading code sequence of the transmitter used to spread the signal. Active digital matched filter designs allow communication using random-length, Pseudo-Random Number (PN) Codes that can be changed anytime required and may be selected from thousands of candidate codes. The added complexity of electronically varying not only the bit content but the length of the spreading pseudo random code greatly increases the required complexity and therefore the ability of an intercept receiver to detect the presence of the communicating signal. The design disclosed in the U.S. Pat. No. 5,677,928 for example uses a reference oscillator frequency to derive the pseudo-random code rate, which then must be used by the RF oscillator as a phase-lock reference to communicate this frequency to the receiver for pseudo-random code clock lock and signal correlation. The addition of another oscillator in the communication path, such as would occur if the communication is through a terrestrial repeater or satellite transponder, injects an unknown frequency into the path. The prior design correlation would then attempt to compensate for this additional drift and quickly lose PN sequence clock lock to the intended transmitter by attempting to lock onto a frequency which differs from that transmitted. The use of the design of the receiver disclosed herein permits recovery of such spread signals when communicating between nodes which are not within radio line-of-sight by adding circuitry for the receiver to track the code clock (CHIP) rate itself without regard to variations in the RF frequency received.

The communication equipment proposed herein consists of a new receiver design. The specific transmitter unit is unchanged from the prior patent application but may also be substantially as described in the prior art. The transmitter is initially activated by the input of the pseudo-random generator code. This code may be any alpha-numeric series, but must be identical to that input to the receiver. When the code is input through the keypad, the transmitter uses internal algorithms to generate a unique shift-register feedback combination. This creates a pseudo-random (PN) code which repeats after a programmable number of bits, depending upon desired maximal signal correlation time. That is the longer the code the longer the correlation time. This PN code sequence is clocked at the desired clock (CHIP) rate inside the transmitter unit. The PN code sequence which is thus generated is then added in phase to the data which is a substantially lower rate than the CHIP rate. After mixing with the PN code and data, the RF frequency CW signal power is spread across a frequency band equivalent to twice the PN code clock rate. The signal power density per hertz is now well below an identical non-spread signal at the communications data rate and ideally below the systems noise level. At the receiver, the same pseudo-random code which was used to transmit the information is entered into the receiver unit for activation. This is then used to configure three programmable PN sequence generators which output three identical PN code sequences which vary only in a one-half phase shift of the CHIP rate from each other. The signals received by the antenna are converted to an intermediate frequency and mixed with the locally generated PN code sequences. When code alignment occurs, threshold detector circuitry senses the presence of the correlated data signals. The relative level of the signals are then sensed by the microprocessor which varies the PN clock oscillator to maintain lock with the incoming signal. The data is now available as useable information at the data port. The system is designed so that the transmitter and receiver signals meet specific correlation requirements and operational performance over transponder or repeated links, as well as direct communications. The receiver incorporates several approaches to meet this correlation time design goal and non-direct communications capability as described hereafter.

FIG. 1 is a block diagram of the transmitter;

FIG. 2 is a pictorial diagram of the Signal transmission characteristics;

FIG. 3 is a block diagram of the receiver;

FIG. 4 is a more detailed block diagram of the receiver;

FIG. 5 is a block diagram of the PN sequence generator; and

FIG. 6 is a pictorial diagram of the spread spectrum signal reception/correlation system.

The communication equipment proposed herein consists of a multiple correlation path receiver design. The particular transmitter unit used herein is unchanged from the prior patent application and may be substantially as described in the prior art provided the same PN code sequence length and content is used. With reference to FIG. 1, a transmitter 1 is initially activated by the input of the pseudo-random generator code (Operator code). This code may be any alpha-numeric series, but must be identical to the code used by the receiver 2, shown in FIG. 3, so as to generate the identical PN code. When the Operator code is input through the keypad 3, (the keypad may alternatively be a set of thumb wheel switches) the processor 4 of transmitter 1, applies the code to the PN code generator 5 of transmitter 1 which uses internal algorithms to create a unique random like numeric shift-register feedback combination and initial register seed. The resulting pseudo-random (PN) code sequence repeats after a programmable number of bits, depending upon desired maximal signal correlation time. The PN sequence in the receiver is clocked at the desired clock (CHIP) rate in the transmitter 1 by dividing the reference oscillator 6 by a specified, pre-selected amount K. The PN sequence which is thus generated is then added by a Phase Coherent Adder 7, in phase with the data to be communicated, which is at a relatively low rate compared to the PN sequence as is well known in the art. The composite PN code/data bits are then mixed in a mixer 8, with the desired RF frequency CW signal 9 for transmission. The system described can operate at any RF carrier frequency and at PN code rates up to approximately 100 megahertz as implemented although higher rates are theoretically possible.

Design enhancements of this invention as indicated herein allow this operational performance over transponder or repeater links, as well as direct communications due to the fact that the receiver and the transmitter need not be locked together. The invention incorporates several approaches to meet fast correlation time design goals and non-direct communications capability. In operation, the transmitter 1 outputs a base-band modulated RF signal at the desired operating frequency. This signal will vary from the nominal frequency because of internal oscillator age, temperature and frequency drift. Additional frequency variances are caused by external oscillator drift and the transponder or repeater, as well as Doppler effects.

With reference to FIG. 2, after mixing the PN code and data, the RF frequency CW signal power 12 is spread across a frequency band equivalent to twice the PN code clock rate. The signal power density 10 per hertz of the spread spectrum signal 1, is now well below the signal power density 12 per hertz of an identical non-spread signal 13 and ideally below the system noise level.

With reference to FIG. 3 the same pseudo-random code (Operator) which was used to generate the transmitter PN code is entered through the keypad 14 of the receiver 2 to generate the receiver 2 PN codes. The receiver 2 microprocessor 15 then generates a clock code which is applied to a clock generator 16 and, in parallel, sends the Operator code to configure the PN sequence generators 17. Clock pulses are then applied to the PN sequence generators 17 by the clock generator 16 and identical internal algorithms generate the same PN code sequence as used by the transmitter 1. However, in the instant receiver design, there are three PN sequence generators 17 which output three identical PN sequences which vary only in phase by one-half cycle from each other.

The signals received by the antenna 18 are down converted by the receiver 2 mixes the incoming spread spectrum RF signal against the locally generated local oscillator 19 signal tuned below the RF carrier by convenient frequency difference to yield an intermediate frequency (IF) as is well known in the art. All under the control of the processor 15, this mixing yields a spread spectrum signal well below the noise floor as centered at the IF with a frequency uncertainty equivalent to the sum of the above-mentioned frequency variances. The system is designed so that the transmitter 1 and receiver 2 signals meet specific correlation requirements. These are correct RF frequency, CHIP rate, PN code and PN code phase match. To be useful in a communications environment, these criteria must be evaluated very quickly. The current system is designed to achieve complete correlation of data output in less than one second, with PN Codes which repeat every 10,000 bits, adding to the signal covertness by allowing "burst" half-duplex communication. With reference to FIG. 4, the signal is then mixed with three locally generated PN code sequences from the three PN code generators 17. The three identical PN sequences are separated in phase by plus or minus one-half CHIP. When correlation has occurred another IF signal will appear at the input of each of the three threshold detector circuitry as a spike of correlated signal energy. The frequency of this signal will be within the sum of the frequency variations from the center IF. This signal first passes through crystal band pass filters 23, with the band centered around the IF and a bandwidth equivalent to the sum of the frequency variations. Then the filtered signal is passed to the threshold detectors 20. A Phase Locked Loop 24, (PLL) senses the filtered correlation signal and changes frequency to phase lock to it. The IF Phase Locked Loop 24 (PLL) signal is used to demodulate the data from the correlator received signal based on the true clock signal. The relative power level of the three IF correlator signals are monitored by the microprocessor 15 which controls the speed of the clock generator 16 which drives the PN sequence generators 17. The microprocessor 15 increases or decreases the speed of the code clock generator 16 under digital control to achieve the maximal correlation signal power in the main PN correlation mixer 25 by maintaining equal signal power in the leading one-half phase path+ and lagging one-half phase correlator path-. That is when the signal power level of the leading one-half phase and lagging one-half phase correlator paths are equal then the base signal power of the true clock PN correlator is at its maximum. The true clock signal is in turn demodulated by a demodulator 33 as shown in FIG. 4 through straight forward PSK techniques and the low rate data is made available as useable information at the data port 21.

In this invention the transmitter and receiver units have equivalent circuitry, creating the identical PN code sequences driving the spreading and correlating mixers. A transmitting unit requires one PN sequence generator circuit and the receiver requires three PN sequence generating circuits in order to lock onto the true data signal as described herein. As shown in FIG. 5, the PN sequence generator is a programmable length shift register 26 with programmable feedback taps 27 including multiplexors 27a and 27b and a variable Operator data tap 28 which permits the creation of create thousands of multiple length PN codes used to spread the low rate data. The receiver contains logic circuitry (not shown) which divides the code clock between the various sequence generators so only one oscillator is required. In order to achieve maximal signal correlation and "de-spread" the incoming signal, the phase transitions in the incoming signal must occur at precisely the same time as the PN code mixing sequence 20 in the receiver unit correlator mixer as is shown in FIG. 6. In a properly constructed PN code, this occurs only when the received code and the internally generated codes are phase aligned The correlation circuitry includes a CHIP retard/delay circuit, well known in the art, which allows very fast PN code alignment, of the three threshold detectors, using analog to digital converters. It is important for the establishment of initial lock for the threshold of the true clock to be evaluated by the microprocessor 15. The microprocessor 15 monitors the relative IF power signal levels 31, 32 and 33 respectively. In certain circumstances the threshold detected for the phase shifted signals may indeed be equal while the true clock threshold is actually below these thresholds. This is an indication that proper PN sequence phase lock has not been achieved. The microprocessor computes an average noise level of the environment from the threshold detector and determines if the incoming signal is properly phased by computing the difference in the level of the true clock signal energy and the average of the environment. In the preferred environment this difference is set to a value of at least 4 dB. If the power level difference is less than the threshold, the microprocessor 15 then causes a delay in the clock by retarding it one CHIP and then again computing the difference between the noise environment and the true clock signal energy. Once the true clock threshold reaches the predetermined level above the environment (e.g. 4 dB) which is deemed acceptable, the microprocessor varies the rate of the receiver PN sequence clock as required to keep the base or true clock threshold signal level at a maximum and the leading 31 and lagging 32 one-half phase CHIP rate threshold power levels equal. The CHIP rate and PN sequence for the receiver and the received true signal are then exactly matched and maximal processing gain is achieved. The assembly language program used to drive the microprocessor 15 which implements the digital control elements in the receiver is set forth in table I below along with initial variables and a memory map for the CHIP rate configuration information and PN code generator configuration selections. ##SPC1##

The spread spectrum communication system described herein greatly increases the utility of a direct spread spectrum communication system design. The advantage of a programmable variable length code sequences and ultra-fast signal correlation allowed in the design permits the spread spectrum communication system to be used for non-direct communication and the length of the PN code to be changed even while receiving data. As the design can be used at any RF frequency and CHIP rate, the system can be tailored for use over any existing or future communications link any where in the world.