Using Video Signals From Auxiliary Analog TV Receivers for Detecting NTSC Interference in Digital TV Receivers

The present invention relates to digital television systems, and more particularly, to circuits employed in the digital television receiver for determining whether or not there is co-channel interference from NTSC analog television signals.

A Digital Television Standard published 16 Sep., 1995 by the Advanced Television Subcommittee (ATSC) specifies vestigial sideband (VSB) signals for transmitting digital television (DTV) signals in 6-MHz-bandwidth television channels such as those currently used in over-the-air broadcasting of National Television Subcommittee (NTSC) analog television signals within the United States. The VSB DTV signal is designed so its spectrum is likely to interleave with the spectrum of a co-channel interfering NTSC analog TV signal. This is done by positioning the pilot carrier and the principal amplitude-modulation sideband frequencies of the DTV signal at odd multiples of one-quarter the horizontal scan line rate of the NTSC analog TV signal that fall between the even multiples of one-quarter the horizontal scan line rate of the NTSC analog TV signal, at which even multiples most of the energy of the luminance and chrominance components of a co-channel interfering NTSC analog TV signal will fall. The video carrier of an NTSC analog TV signal is offset 1.25 MHz from the lower limit frequency of the television channel. The carrier of the DTV signal is offset from such video carrier by 59.75 times the horizontal scan line rate of the NTSC analog TV signal, to place the carrier of the DTV signal about 309,877.73 kHz from the lower limit frequency of the television channel. Accordingly, the carrier of the DTV signal is about 2,690122.27 Hz from the middle frequency of the television channel. The exact symbol rate in the Digital Television Standard is (684/286) times the 4.5 MHz sound carrier offset from video carrier in an NTSC analog TV signal. The number of symbols per horizontal scan line in an NTSC analog TV signal is 684, and 286 is the factor by which horizontal scan line rate in an NTSC analog TV signal is multiplied to obtain the 4.5 MHz sound carrier offset from video carrier in an NTSC analog TV signal. The symbol rate is 10.762238 megasymbols per second, which can be contained in a VSB signal extending 5.381119 MHz from DTV signal carrier. That is, the VSB signal can be limited to a band extending 5.690997 MHz from the lower limit frequency of the television channel.

The ATSC standard for digital HDTV signal terrestrial broadcasting in the United States of America is capable of transmitting either of two high-definition television (HDTV) formats with 16:9 aspect ratio. One HDTV format uses 1920 samples per scan line and 1080 active horizontal scan lines per 30 Hz frame with 2:1 field interlace. The other HDTV format uses 1280 luminance samples per scan line and 720 progressively scanned scan lines of television image per 60 Hz frame. The ATSC standard also accommodates the transmission of DTV formats other than HDTV formats, such as the parallel transmission of four television signals having normal definition in comparison to an NTSC analog television signal.

DTV transmitted by vestigial-sideband (VSB) amplitude modulation (AM) during terrestrial broadcasting in the United States of America comprises a succession of consecutive-in-time data fields each containing 313 consecutive-in-time data segments. There are 832 symbols per data segment. So, with the symbol rate being 10.76 MHz, each data segment is of 77.3 microseconds duration. Each segment of data begins with a line synchronization code group of four symbols having successive values of +S, -S, -S and +S. The value +S is one level below the maximum positive data excursion, and the value -S is one level above the maximum negative data excursion. The initial line of each data field includes a field synchronization code group that codes a training signal for channel-equalization and multipath suppression procedures. The training signal is a 511-sample pseudo-noise sequence (or "PN-sequence") followed by three 63-sample PN sequences. The middle ones of the 63-sample PN sequences in the field synchronization codes are transmitted in accordance with a first logic convention in the first line of each odd-numbered data field and in accordance with a second logic convention in the first line of each even-numbered data field, the first and second logic conventions being one's complementary respective to each other.

The data within data lines are trellis coded using twelve interleaved trellis codes, each a 2/3 rate trellis code with one uncoded bit. The interleaved trellis codes are subjected to Reed-Solomon forward error-correction coding, which provides for correction of burst errors arising from noise sources such as a nearby unshielded automobile ignition system. The Reed-Solomon coding results are transmitted as 8-level (3 bits/symbol) one-dimensional-constellation symbol coding for over-the-air transmission, which transmissions are made without symbol precoding separate from the trellis coding procedure. The Reed-Solomon coding results are transmitted as 16-level (4 bits/symbol) one-dimensional-constellation symbol coding for cablecast, which transmissions are made without precoding. The VSB signals have their natural carrier wave, which would vary in amplitude depending on the percentage of modulation, suppressed.

The natural carrier wave is replaced by a pilot carrier wave of fixed amplitude, which amplitude corresponds to a prescribed percentage of modulation. This pilot carrier wave of fixed amplitude is generated by introducing a direct component shift into the modulating voltage applied to the balanced modulator generating the amplitude-modulation sidebands that are supplied to the filter supplying the VSB signal as its response. If the eight levels of 4-bit symbol coding have normalized values of -7, -5, -3, -1, +1, +3, +5 and +7 in the carrier modulating signal, the pilot carrier has a normalized vale of 1.25. The normalized value of +S is +5, and the normalized value of-S is -5.

The video carrier of an NTSC signal is 1.25 MHz from edge of the 6-MHz-wide broadcast channel, while the carrier for a digital TV signal for terrestrial through-the-air broadcast is 310 kHz from edge of the 6-MHz-wide broadcast channel. A co-channel NTSC signal does not exhibit symmetrical amplitude-modulation sidebands with respect to the carrier of the vestigial-sideband amplitude-modulation (VSB AM) carrying digital information. Accordingly, the NTSC video carrier at 940 kHz remove from digital TV signal carrier and its sidebands at further remove from digital TV signal carrier are not well canceled in the digital TV signal. Nor, of course, is the NTSC audio carrier at 5.44 MHz remove from digital TV signal carrier.

In the earlier development of the DTV art it was contemplated that the DTV broadcaster might be called upon to decide whether or not to use a symbol precoder at the transmitter, which symbol precoder would follow the symbol generation circuitry and provide for matched filtering of symbols, when used together with a comb filter in each DTV receiver used before the data-slicer in the symbol decoder circuitry as a symbol post-coder. This decision would have depended upon whether interference from a co-channel NTSC broadcasting station were expected or not. Symbol precoding would not have been used for data line synchronization code groups or during data lines in which data field synchronization data were transmitted. Co-channel interference is reduced at greater distances from the NTSC broadcasting station(s) and is more likely to occur when certain ionospheric conditions obtain, the summertime months during years of high solar activity being notorious for likelihood of co-channel interference. Such interference will not obtain if there are no co-channel NTSC broadcasting stations, of course. If there were likelihood of NTSC interference within his area of broadcast coverage, it was presumed that the HDTV broadcaster would use the symbol precoder to facilitate the HDTV signal being more easily separated from NTSC interference; and, accordingly, a comb filter would be employed as symbol post-coder in the DTV receiver to complete matched filtering. If there were no possibility of NTSC interference or there were insubstantial likelihood thereof, in order that flat spectrum noise would be less likely to cause erroneous decisions as to symbol values in the trellis decoder, it was presumed that the DTV broadcaster would discontinue using the symbol precoder; and, accordingly, the symbol post-coder would then be disabled in each DTV receiver. A problem with using preceding at the transmitter is that, while preceding may be preferable insofar as certain DTV receivers are concerned, precoding may not be desirable for other DTV receivers receiving transmissions from the transmitter. DTV receivers less remote from the transmitter or more remote from a co-channel analog TV transmitter may not suffer significant amounts of co-channel interference, for example.

The Digital Television Standard the ATSC published 16 Sep., 1995 does not allow for the use of precoding of all data at the DTV transmitter to compensate for post-coding incidental to subsequent use of comb filtering in a DTV signal receiver to reject NTSC co-channel interference. Instead, only the initial symbol in the trellis decoding is precoded. This procedure by itself does not facilitate a DTV signal receiver using comb filtering to reject NTSC co-channel interference before data slicing procedures are undertaken. A DTV signal receiver that does not reject NTSC co-channel interference before data slicing procedures are undertaken will not have good reception under strong NTSC co-channel interference conditions as may be caused by the DTV receiver being remote from the DTV transmitter or having an analog TV transmitter very closeby.

In U.S. patent application Ser. No. 08/746,520 filed by the inventor on 12 Nov., 1996 issued 5 May, 1998 as U.S. Pat. No. 5,748,226 and entitled "DIGITAL TELEVISION RECEIVER WITH ADAPTIVE FILTER CIRCUITRY FOR SUPPRESSING NTSC CO-CHANNEL INTERFERENCE", the inventor advocates preceding data-slicing in a DTV receiver with comb filtering to suppress NTSC co-channel interference when that interference is sufficiently large as to affect data-slicing adversely. The inventor teaches how to compensate in the symbol decoding procedure for the effects of such comb filtering upon symbol coding when it is selectively done. It is, then, still useful to be able to determine when NTSC co-channel interference is larger than a prescribed value denominated as being acceptably small, so that this determination can be used for controlling the selective use of comb filtering to suppress NTSC co-channel interference.

Comb filtering procedures are not entirely satisfactory for suppressing the NTSC audio carrier, particularly under conditions of frequency modulation in which carrier frequency deviation is large, since correlation (or anti-correlation) of samples of the FM carrier at times separated by any substantial fixed delay is apt not to be particularly good. U.S. Pat. No. 5,748,226 advocates that the filtering used to establish the overall bandwidth of intermediate-frequency (IF) amplification be such as to reject the FM audio carrier of any co-channel interfering NTSC analog TV signal. Surface-acoustic-wave filtering in an ultra-high-frequency (UHF) intermediate-frequency band is particularly suitable for rejecting the FM audio carrier of any co-channel interfering NTSC analog TV signal. Comb filtering of the baseband symbol codes is more satisfactory for suppressing the NTSC video carrier and nearby sidebands, which tend to exhibit good correlation between samples separated by certain specific delay intervals and to exhibit good anti-correlation between samples separated by other certain specific delay intervals. Suitable comb filtering procedures are also satisfactory for suppressing chroma sidebands near the NTSC chrominance subcarrier, which tend to exhibit good correlation between samples separated by certain specific delay intervals and to exhibit good anti-correlation between samples separated by other certain specific delay intervals. Suppressing the NTSC audio carrier during IF amplification allows comb filtering of the symbol codes with comb filters that reject NTSC video carrier and its chrominance subcarrier, but do not reject NTSC audio carrier very much. One such comb filter, which additively combines symbol codes having six-symbol-epochs differential delay, is favored since both the NTSC video carrier and its chrominance subcarrier tend to exhibit high degrees of self-anti-correlation over such a short differential delay.

U.S. Pat. No. 5,260,793 issued 9 Nov., 1993 to R. W. Citta et alii and entitled "RECEIVER POST CODER SELECTION CIRCUIT" selectively employs a post-coder comb filter for suppressing NTSC interference accompanying a real or in-phase baseband component (I channel) of the complex output signal of a demodulator used in a digital high-definition television (HDTV) receiver. The presence of NTSC interference in the I-channel component of the demodulator response is detected for developing control signals automatically to enable or disable the comb filter being used for suppressing NTSC co-channel interference. During each data field sync interval, the input signal to and the output signal from an NTSC suppression filter of comb filter type in the HDTV receiver are each compared with a respective signal that is known a priori and is drawn from memory within the HDTV receiver. If the minimum result of comparison with the input signal has less energy than the minimum result of comparison with the output signal from the NTSC suppression filter, this is indicative that the primary cause of variance from expected reception is random noise rather than NTSC co-channel interference. Insofar as the particular digital television receiver is concerned, reception would be better were precoding and post-coding not employed in the system, and it is presumed that the broadcaster has not employed precoding. If the minimum result of comparison with the input signal has more energy than the minimum result of comparison with the output signal from the NTSC suppression filter, this is indicative that the primary cause of variance from expected reception is NTSC co-channel interference rather than random noise. Insofar as the particular digital television receiver is concerned, reception would be better were precoding and post-coding employed in the system, and it is presumed that the broadcaster has employed precoding.

U.S. Pat. No. 5,546,132 issued 13 Aug., 1996 to K. S. Kim et alii and entitled "NTSC INTERFERENCE DETECTOR" describes attempts to detect co-channel NTSC interference from data within the data segments other than those transmitting data field sync information. A comb filter that subtractively combines differentially delayed symbol coding in the I-channel is used at times it is necessary to suppress co-channel NTSC interference; and the use of a comb filter that additively combines differentially delayed symbol coding in the I-channel is advocated for extracting co-channel NTSC interference, so that the energy of the co-channel NTSC interference within a data segment can be determined. The problem with this approach is that the comb filter Kim et alii propose to use for extracting co-channel NTSC interference also has some response to symbol coding associated with the digital signal transmission. This is particularly true when the DTV transmitter does not use preceding, so symbol code energy is concentrated at portions of the frequency spectrum that interleave with portions of the frequency spectrum in which NTSC signal energy is concentrated. The energy associated with symbol coding can be higher than the energy of co-channel NTSC interference that will cause significant error in data slicing. Clearly, a better approach is required for determining, from data within the data segments other than those transmitting data field sync information, whether there are significant amounts of co-channel NTSC interference.

The Kim et alii apparatus can be improved by replacing the absolute value circuit they use after the comb filter used for extracting co-channel NTSC interference with a synchronous detector operable at the NTSC video carrier frequency of 59.75 times horizontal scan line rate, it is here pointed out. However, this introduces quite a bit of complexity into the Kim et alii apparatus. The advantages of synchronous detection of NTSC video carrier with regard to being able to reject DTV modulation are better achieved in another way.

So long as analog television transmissions are made, causing NTSC co-channel interference to be a problem, DTV receivers will for commercial reasons probably be provided with auxiliary receiver apparatus for receiving analog TV broadcasting. The inventor points out that such auxiliary receiver apparatus can be designed to provide an excellent means for detecting the presence of NTSC co-channel interference, rather than attempting to detect the presence of NTSC co-channel interference in the baseband symbol coding recovered by the DTV portions of the receiver. For example, the degree of NTSC co-channel interference can be evaluated by detecting the amplitude of the intercarrier sound intermediate-frequency generated from the mixing together of the video and audio carriers of the NTSC co-channel interfering signal as described in a concurrently filed U.S. patent application Ser. No. 08/821,945 of the inventor entitled "USING INTERCARRIER SIGNALS FOR DETECTING NTSC INTERFERENCE IN DIGITAL TV RECEIVERS". The degree of NTSC co-channel interference can also be evaluated proceeding from the NTSC video signal recovered by synchronous video detection in auxiliary receiver apparatus for receiving analog TV broadcasting.

Amplitude-modulation (AM) detection of the NTSC video carrier in such auxiliary receiver apparatus is arranged to be synchronous in nature in implementing the invention, with an in-phase synchronous detector for detecting the NTSC video carrier amplitude modulation and with a quadrature-phase synchronous detector for developing automatic frequency and phase control (AFPC) signal for a controlled oscillator supplying synchrodyning signals to the synchronous detectors. Since all the energy of the NTSC co-channel interfering signal is in the real portion of the AM complex-frequency spectrum, but the energy of the DTV signal appears in like amount in the real and imaginary portions of the AM complex-frequency spectrum, synchronously detecting the real portion of the AM complex-frequency spectrum to separate it from the imaginary portion allows the NTSC co-channel interfering signal to gain 2:1 in energy over the DTV artifacts as compared to what obtains in the results of simple envelope detection. To the extent that the double sideband structure of the lower-frequency AM of the NTSC video carrier is maintained in the IF signal supplied to the synchronous video detector, there is further gain of lower video frequencies over DTV artifacts in the baseband response of the in-phase synchronous video detector to the real portion of the AM complex-frequency spectrum. The fixed-amplitude artifact of the DTV signal pilot carrier is translated to 940 KHz, above the frequencies of most energy in the luminance signal portion of an NTSC co-channel interfering signal. A considerable amount of the modulation energy of the DTV signal is in artifacts at considerable remove from 0 Hz in that luminance signal because of symbol level randomization in the DTV signal to avoid direct components of modulation other than the fixed pilot carrier. The peak energy in that luminance signal is contained in the synchronizing pulses, at frequencies below 940 KHz.

Accordingly, in certain embodiments of the invention a lowpass filter with a cut-off frequency below 940 KHz is connected in cascade after the in-phase synchronous video detector, and the response of this lowpass filter is used during DTV signal reception to provide an indication of the strength of an NTSC co-channel interfering signal. A threshold detector responds to this indication to generate a decision as to whether a co-channel interfering analog TV signal has sufficient energy to cause problems when symbol decoding. Responsive to a decision that co-channel interfering analog TV signal does have sufficient energy to cause problems when symbol decoding, symbol coding is comb filtered to suppress co-channel interfering analog TV signal before performing symbol decoding.

In each of alternative embodiments of the invention, the in-phase synchronous video detector response is filtered with a filter providing frequency-selective response suppressing frequencies at which DTV artifacts are expected to exhibit most energy, which response is used during DTV signal reception to provide an indication of the strength of an NTSC co-channel interfering signal. A threshold detector responds to this indication to generate a decision as to whether a co-channel interfering analog TV signal has sufficient energy to cause problems when symbol decoding. Responsive to a decision that co-channel interfering analog TV signal does have sufficient energy to cause problems when symbol decoding, symbol coding is comb filtered to suppress co-channel interfering analog TV signal before performing symbol decoding.

In still other embodiments of the invention, the Hilbert transform of the quadrature-phase synchronous video detector response is linearly combined with the in-phase synchronous video detector response for separating the NTSC co-channel interfering signal from DTV artifacts. The results of the linear combining is used during DTV signal reception to provide an indication of the strength of an NTSC co-channel interfering signal. A threshold detector responds to this indication to generate a decision as to whether a co-channel interfering analog TV signal has sufficient energy to cause problems when symbol decoding. Responsive to a decision that co-channel interfering analog TV signal does have sufficient energy to cause problems when symbol decoding, symbol coding is comb filtered to suppress co-channel interfering analog TV signal before performing symbol decoding.

FIG. 1 is a schematic diagram of radio receiver portions of a DTV receiver which embody the invention in a first form.

FIG. 2 is a schematic diagram of radio receiver portions of a DTV receiver which embody the invention in a second form.

FIG. 3 is a schematic diagram of radio receiver portions of a DTV receiver which embody the invention in a third form.

FIG. 4 is a schematic diagram of radio receiver portions of a DTV receiver which embody the invention in a fourth form.

FIG. 5 is a schematic diagram showing details of symbol decoding apparatus for use in the circuitry of FIG. 1, FIG. 2, FIG. 3 or FIG. 4.

FIG. 6 is a schematic diagram showing details of alternative symbol decoding apparatus for use in the circuitry of FIG. 1, FIG. 2, FIG. 3 or FIG. 4.

FIG. 7 is a schematic diagram showing details of analog filtering for suppressing DTV artifacts in NTSC video signals synchronously detected in the circuitry of FIG. 1, FIG. 2, FIG. 3 or FIG. 4 for evaluating the strength of co-channel interference those NTSC video signals cause.

FIG. 8 is a schematic diagram showing details of alternative analog filtering for suppressing DTV artifacts in NTSC video signals synchronously detected in the circuitry of FIG. 1, FIG. 2, FIG. 3 or FIG. 4 for evaluating the strength of co-channel interference those NTSC video signals cause.

FIG. 9 is a schematic diagram showing details of digital filtering for suppressing DTV artifacts in NTSC video signals synchronously detected in the circuitry of FIG. 1, FIG. 2, FIG. 3 or FIG. 4 for evaluating the strength of co-channel interference those NTSC video signals cause.

FIG. 10 is a schematic diagram showing details of alternative digital filtering for suppressing DTV artifacts in NTSC video signals synchronously detected in the circuitry of FIG. 1, FIG. 2, FIG. 3 or FIG. 4 for evaluating the strength of co-channel interference those NTSC video signals cause.

FIG. 11 is a schematic diagram of radio receiver portions of a DTV receiver which embody the invention in a fifth form.

FIG. 12 is a schematic diagram of radio receiver portions of a DTV receiver which embody the invention in a sixth form.

FIG. 1 shows the invention used with the radio receiver portions of a digital TV receiver with an auxiliary analog TV receiver, which combined TV receiver is of a first plural-conversion type. Species of this first plural-conversion type of receiver are described in detail by the inventor in U.S. patent application Ser. No. 08/825,711 filed 19 Mar., 1977 and entitled "RADIO RECEIVER DETECTING DIGITAL AND ANALOG TELEVISION RADIO-FREQUENCY SIGNALS WITH SINGLE FIRST DETECTOR". Over-the-air type television broadcasting signals as received by an antenna 1 are amplified by an adjustably tuned radio-frequency amplifier 2 and supplied to a first detector 3. The RF amplifier 2 and the first detector 3 have adjustable tuning and together function as a tuner for selecting said digital television signal from one of channels at different locations in a frequency band. The first detector 3 includes a first local oscillator supplying first local oscillations tunable over a frequency range above the UHF TV broadcast band and a first mixer for mixing the first local oscillations with a TV signal selected by the adjustably tuned RF amplifier 2 for upconverting the selected TV signal to generate a UHF intermediate-frequency signal in a 6-MHz-wide UHF intermediate-frequency band located at frequencies above the assigned channels in the UHF TV broadcast band. The first detector 3 supplies the high-IF-band signal to a UHF-band intermediate-frequency amplifier 4 for ATSC DTV signals, to a UHF-band intermediate-frequency amplifier 5 for NTSC video signals, and to a UHF-band intermediate-frequency amplifier 6 for NTSC audio signals. The responses of the UHF-band IF amplifiers 4, 5 and 6 are supplied to respective second detectors 7, 8 and 9 to be downconverted to respective VHF-band intermediate-frequency signals in a VHF band below the very high frequencies assigned as TV broadcast channels. The second detectors 7, 8 and 9 share a common second local oscillator for generating second local oscillations and have respective second mixers for mixing those second local oscillations with the responses of the UHF-band IF amplifiers 4, 5 and 6, respectively. The VHF-band IF signals from the second detector detectors 7, 8 and 9 are respectively supplied to a VHF-band intermediate-frequency amplifier 10 for ATSC DTV signals, to a VHF-band intermediate frequency amplifier 11 for NTSC video signals, and to a VHF-band intermediate frequency amplifier 12 for NTSC audio signals.

The UHF-band IF amplifiers 4, 5 and 6 include surface-acoustic-wave (SAW) filters for UHF-IF-band ATSC DTV signal, for UHF-IF-band NTSC video signal and for UHF-IF-band NTSC audio signal, respectively. SAW filters with steep rejection skirts, but with pass bands having linear group delay and flat amplitude response, are more easily implemented at UHF than at VHF. This is the reason for preferring to determine overall IF response for ATSC DTV signal, for NTSC video signal and for NTSC audio signal in the UHF IF band rather than in the VHF IF band. The SAW filter determining overall IF response for ATSC DTV signal is preferably one that rejects NTSC audio signal but otherwise has substantially flat amplitude response over the remainder of the 6-MHz-wide TV broadcast channel as translated to the UHF IF band, and this SAW filter has substantially linear phase response throughout its passband. The SAW filter determining overall IF response for NTSC video signal rejects NTSC audio signal and is preferably one that exhibits a roll-off for the lower frequencies of the 6-MHz-wide TV broadcast channel as translated to the UHF IF band; this SAW filter preferably has substantially linear phase response throughout its passband. The UHF-band IF amplifiers 4, 5 and 6 can include wideband constant-gain amplifiers for driving their component SAW filters from source impedances that minimize multiple reflections and for overcoming the insertion losses of their component SAW filters.

The VHF-band IF amplifiers 10, 11 and 12 include respective controlled-gain amplifiers that provide up to 60 dB or more amplification. The VHF-band IF amplifier 10 is provided with reverse automatic gain control derived in response to its output signal level, reverse AGC being preferred for the linearity of gain it affords. The VHF-band IF amplifiers 11 and 12 each include stages with forward automatic gain control derived in response to the output signal level of the IF amplifier 11, forward AGC being preferred for the better noise figure it affords. The RF amplifier 2 is provided with delayed reverse automatic gain control in response to the output signal level of the IF amplifier 10 when a DTV signal is indicated as being received and in response to the output signal level of the IF amplifier 11 when a DTV signal is not indicated as being received

Output signal from the VHF-band IF amplifier 10 for ATSC DTV signals is applied to an ATSC symbol code detector 13, which detects baseband symbol codes therefrom. The symbol code detector 13 is one which uses an in-phase synchronous detector for detecting the vestigial-sideband amplitude-modulation of the data carrier and uses a quadrature-phase synchronous detector for developing automatic frequency and phase control (AFPC) signal for a controlled oscillator supplying synchrodyning signals to the synchronous detectors. The in-phase synchronous detector operates in the analog regime and its output signal is digitized with 10-bit or so resolution by an analog-to-digital converter 14. Alternatively, the symbol code detector 13 and succeeding ADC 14 can be replaced by a third detector for converting the VHF-band response of the IF amplifier 10 to a final intermediate-frequency band just above baseband, an analog-to-digital converter for digitizing the third detector response, and digital synchrodyning circuitry for synchrodyning the digitized third detector response to baseband. Such alternative circuitry is described by C. B. Patel et alii in U.S. Pat. No. 5,479,449, issued 26 Dec., 1995 and entitled "DIGITAL VSB DETECTOR WITH BANDPASS PHASE TRACKER, AS FOR INCLUSION IN AN HDTV RECEIVER", and in U.S. Pat. No. 5,548,617, issued 20 Aug., 1995 and entitled "DIGITAL VSB DETECTOR WITH BANDPASS PHASE TRACKER USING RADER FILTERS, AS FOR USE IN AN HDTV RECEIVER", by way of examples. When a DTV signal is being received, a direct signal resulting from the synchronous detection of the pilot signal accompanies the symbol codes as reproduced at baseband and is detected by a pilot carrier detector 15 to generate a DTV ENABLE signal. The pilot carrier detector 15 can, as shown in FIG. 1, be of a type to respond to digital input signal or alternatively can be of a type to respond to analog input signal as supplied directly from the symbol code detector 13.

The DTV ENABLE signal conditions the display portions of the DTV receiver to display DTV images with 16:9 aspect ratio and conditions the RF amplifier 2 to receive delayed AGC signal developed in response to the level of the output signal from the VHF-band IF amplifier 10. By way of example, reverse AGC signal for the VHF-band IF amplifier 10 and delayed reverse AGC signal for the RF amplifier 2 can be provided by an AGC detector (not shown in FIG. 1) which AGC detector is responsive to the direct signal resulting from the synchronous detection of the pilot signal. This is more particularly described by C. B. Patel et alii in U.S. patent application Ser. No. 08/573,454 filed 15 Dec., 1995 issued 3 Jun., 1997 as U.S. Pat. No. 5,636,252 and entitled "AUTOMATIC GAIN CONTROL OF RADIO RECEIVER FOR RECEIVING DIGITAL HIGH-DEFINITION TELEVISION SIGNALS".

FIG. 1 shows the digitized baseband symbol codes being supplied from the ADC 14 to a symbol decoder 20 of the type more particularly described in the above-referred-to U.S. patent application Ser. No. 08/746,520. The symbol decoder 20 comprises a data slicer 21 for data-slicing the symbol decoder 20 input signal to produce a first symbol decoder response, an NTSC-artifact-rejection comb filter 22 supplying a response to the symbol decoder 20 input signal which response suppresses any NTSC co-channel interfering signal, a data slicer 23 for data-slicing the comb filter 22 response for generating an erroneous symbol decoder response, a matching comb filter 24 for correcting that erroneous symbol decoder response to produce a second symbol decoder response, and a multiplexer 25 for selecting one of the first and second symbol decoder responses as the ultimate symbol decoder response supplied by the symbol decoder 20 to a trellis decoder 16 typical to a DTV receiver. In the absence of an indication of substantial NTSC co-channel interfering signal being received, the multiplexer 25 selects the first symbol decoder response from the data slicer 21 to provide the symbol decoder 20 output signal to the trellis decoder 16. In the presence of the indication of substantial NTSC co-channel interfering signal being received, except during symbol decoder initialization intervals, the multiplexer 25 selects the second symbol decoder response from the matching comb filter 24 to provide the symbol decoder 20 output signal to the trellis decoder 16.

The indication of substantial NTSC co-channel interfering signal being received is developed by the auxiliary receiver for analog TV signals and, more particularly, from the baseband composite video response of an NTSC video detector 17 to output signal from the VHF-band IF amplifier 11 for NTSC video signal.

FIG. 1 shows the baseband composite video response from the video detector 17 being supplied to the remaining portions of the analog TV receiver part of the DTV receiver. In regard to baseband composite video signal, these remaining portions typically include sync separation circuitry, color signal reproduction circuitry, and circuitry for adapting the 4:3 aspect ratio NTSC image for presentation on a 16:9 viewscreen used for displaying DTV images.

Furthermore, FIG. 1 shows, in the lower left thereof, the baseband composite video response from the video detector 17 being supplied to a DTV artifacts suppression filter 18 operative in the analog regime, the response of which filter is digitized by an analog-to-digital converter 19. The digital response of the ADC 19 is shown being supplied to a DTV artifacts suppression filter 30 operative in the digital regime. The digital baseband composite video signal with DTV artifacts suppressed therefrom is supplied to a digital squarer 31, which could be constructed from a digital multiplier receiving that signal both as multiplier and as multiplicand, but which is more practical to realize in read-only memory. The squarer 31 output signal is an indication of the energy of NTSC co-channel interfering signal during DTV reception.

A digital threshold detector 32 determines when this indication is strong enough to exceed a threshold value below which NTSC co-channel interfering signal is considered not to be substantial enough to be likely to introduce uncorrectable error into the operation of the data slicer 21. The threshold detector 32 response is supplied to multiplexer control circuitry 33. The multiplexer control circuitry 33 controls the selection by the multiplexer 25 between first and second symbol decoder responses that determines the ultimate symbol decoder response supplied as the symbol decoder 20 output signal. The multiplexer control circuitry 33 conditions the multiplexer 25 to select the first symbol decoder response as the symbol decoder 20 output signal during symbol decoder initialization intervals. At other times the multiplexer control circuitry 33 conditions the multiplexer 25 to select the first symbol decoder response as the symbol decoder 20 output signal as long as the threshold detector 32 response indicates that NTSC co-channel interfering signal is considered not to be substantial enough to be likely to introduce uncorrectable error into the operation of the data slicer 21, otherwise conditioning the multiplexer 25 to select the second symbol decoder response as the symbol decoder 20 output signal.

Output signal from the VHF-band IF amplifier 12 for NTSC audio signals is applied to an intercarrier sound detector 34, which supplies 4.5 MHz intercarrier sound intermediate-frequency signals to an intercarrier sound intermediate-frequency amplifier 35 which amplifies and in most designs symmetrically limits the amplified response for application to an FM detector 36. The FM detector 36 reproduces baseband composite audio signal supplied to the remaining portions of the analog TV receiver part of the DTV receiver. In regard to baseband composite audio signal, these remaining portions typically include stereophonic decoder circuitry. If the NTSC audio signals are selected with narrowband filtering in the IF amplifiers 6 and 12 that pass only the FM audio carrier as translated to intermediate frequencies, the intercarrier sound detector 34 can be provided by a multiplier that multiplies the IF amplifier 12 response by video carrier selected to the multiplier by a narrowband filter responsive to the response of the IF amplifier 10 or 11. If the NTSC audio signals are selected with filtering in the IF amplifiers 6 and 12 that passes both the NTSC video and audio carriers as translated to intermediate frequencies, for implementing "quasi-parallel" sound, the intercarrier sound detector 34 can be a simple rectifier or can be a square-law device.

FIG. 2 shows the invention used with the radio receiver portions of a digital TV receiver with an auxiliary analog TV receiver, which combined TV receiver is of a second plural-conversion type. Species of this second plural-conversion type of DTV receiver are described in detail by the inventor in U.S. patent application Ser. No. 08/820,193 filed 19 Mar., 1997 and entitled "DIGITAL-AND-ANALOG-TV-SIGNAL RECEIVERS, EACH WITH SINGLE FIRST DETECTOR AND SHARED HIGH-BAND I-F AMPLIFICATION". The separate UHF IF amplifiers 4 and 5 for ATSC DTV signal and for NTSC video signal are dispensed with in favor of a shared UHF IF amplifier 37, and the separate second detectors 7 and 8 for ATSC DTV signal and for NTSC video signal are dispensed with in favor of a shared second detector 38. A SAW filter in the shared UHF IF amplifier 37, which determines overall IF response for ATSC DTV signal and for NTSC video signal, is one that rejects NTSC audio signal but otherwise has substantially flat amplitude response over the remainder of the 6-MHz-wide TV broadcast channel as translated to the UHF IF band; and this SAW filter has substantially linear phase response throughout its passband. The roll-off of lower video carrier sideband for NTSC video signal as translated to intermediate frequencies can no longer be introduced in the UHF intermediate frequencies. The VHF IF amplifier 11 for NTSC video signal is replaced by a VHF IF amplifier 39 that introduces the roll-off of lower video carrier sideband as translated to intermediate frequencies.

FIG. 3 shows the invention used with the radio receiver portions of a digital TV receiver with an auxiliary analog TV receiver, which combined TV receiver is similar to that of FIG. 2 but does not replace the VHF IF amplifier 11 by a VHF IF amplifier 39 that introduces the roll-off of lower video carrier sideband as translated to intermediate frequencies. Instead, a match filter 40 for NTSC composite video signal as synchrodyned to baseband is introduced after the video detector 17, and the response of the match filter 40 is supplied to the rest of the analog TV receiver circuitry. The match filter 40 may be an analog filter receiving the composite video signal directly from the video detector 17, as shown in FIG. 1. Alternatively, the composite video signal from the video detector 17 may be digitized for application to a digital match filter for NTSC video signal.

FIG. 4 shows the invention used with the radio receiver portions of a digital TV receiver with an auxiliary analog TV receiver, which combined TV receiver is of a third plural-conversion type. This third plural-conversion type of combined TV receiver modifies the FIG. 3 combined TV receiver by replacing the separate VHF-band IF amplifiers 10 and 11 for ATSC DTV signal and for NTSC video signal with a common VHF-band IF amplifier 41 supplying its response to the symbol code detector 10 and to the video detector 17 as their respective input signals.

FIG. 5 shows in more detail the multiplexer control circuitry 33 and the symbol decoder 20 as it appears in certain embodiments of the invention. In those certain embodiments the NTSC-artifact-rejection comb filter 22 comprises a clocked delay circuit 221 for differentially delaying the input signal supplied to the symbol decoder 20 by twelve symbol epochs and a digital subtractor 222 for differentially combining the input signal and the input signal delayed by twelve symbol epochs. The comb filter comprising elements 221 and 222 suppresses NTSC artifacts near NTSC video carrier frequency offset from DTV carrier, near NTSC chroma subcarrier frequency offset from DTV carrier, and near NTSC audio carrier frequency offset from DTV carrier. The matching comb filter 24 for the NTSC-artifact-rejection comb filter 22 comprises a clocked delay circuit 241 for differentially delaying the data-slicer 23 response by twelve symbol epochs and a digital adder 242 for additively combining the data-slicer 23 response and the data-slicer 23 response delayed by twelve symbol epochs.

The multiplexer control circuitry 33 shown in FIG. 5 operates to condition the multiplexer 25 to select the first symbol decoder response as the symbol decoder 20 output signal during symbol decoder initialization intervals that occur the first symbol periods following the data segment sync intervals. A data segment sync detector 43 response changes from logic ZERO to logic ONE responsive to the occurrence of data segment sync codes in the digitized baseband symbol codes supplied to the detector 43 from the ADC 14. The data segment sync detector 43 is designed so its output signal goes to its logic ONE condition only during the first symbol period following each data segment sync code group. A pulse stretcher 331 responds to each logic ONE output from the data segment sync detector 43 with an output signal that is logic ONE during the twelve symbol periods immediately following each data segment sync code group and that is otherwise logic ZERO. A first input connection of an OR gate 332 is connected to receive the output signal of the pulse stretcher 331. The OR gate 332 responds to the pulse stretcher 331 logic ONE condition with an output ONE that conditions the multiplexer 25 to select the first symbol decoder response from the data-slicer 21 as the symbol decoder 20 output signal during the twelve symbol periods immediately following each data segment sync code group.

The multiplexer control circuitry 33 shown in FIG. 5 includes a ten-stage binary counter 333 capable of counting up to 1024 and reset to zero count responsive to the data segment sync detector 43 going to its logic ONE condition at the beginning of each data segment. The counter 333 is used for counting the number of symbol periods during the current data segment that the threshold detector 32 indicates the data-slicer 21 is apt to err in symbol decoding, owing to accompanying NTSC co-channel interference being substantial. The response of the threshold detector 32 and a clock signal supplied at symbol rate are applied to a two-input AND gate 334 to generate count input signal for the counter 333. The count output of the counter 333 is supplied to a digital threshold detector 335. The threshold detector 335 generates a logic ONE when and only when the number of symbol periods during the current data segment that the threshold detector 32 indicates the data-slicer 21 is apt to err in symbol decoding reaches a number indicative that the error correcting capabilities of the trellis decoder 16 and a subsequent Reed-Solomon decoder are likely to be exceeded. The threshold detector 335 otherwise generates a logic ZERO. A logic inverter 336 one's complements the threshold detector 335 output signal, and the logic inverter 336 output signal is supplied to a bit latch 337 as conditional input signal.

The multiplexer control circuitry 33 shown in FIG. 5 decides at the beginning of each data segment, before the first symbol period finishes, as to whether or not symbols in the remaining symbol periods of that data segment require comb filtering to suppress objectionably high levels of NTSC co-channel interfering signal. The bit latch 337 stores the results of that decision throughout the remaining symbol periods of that data segment, a logic ZERO being stored if the decision is that the level NTSC co-channel interfering signal is substantial enough to require the comb filtering, and a logic ONE being stored if the decision is that the level NTSC co-channel interfering signal is not so substantial. The decision is stored in the bit latch 337 in response to the data segment sync detector 43 response being logic ONE during the first symbol period. A second input connection of the OR gate 332 is connected to receive the decision stored in the bit latch 337. If the decision is that the level NTSC co-channel interfering signal is not substantial enough to require the comb filtering, the logic ONE output signal from the bit latch 337 is repeated in its output signal to condition the multiplexer 25 to select the first symbol decoder response from the data-slicer 21 as the symbol decoder 20 output signal during the remaining symbol periods of the current data segment. If the decision is that the level NTSC co-channel interfering signal is substantial enough to require the comb filtering, the logic ZERO output signal from the bit latch 337 is repeated in its output signal to condition the multiplexer 25 to select the matching comb filter 24 response as the symbol decoder 20 output signal during the remaining symbol periods of the current data segment.

Considerable attention must be given to arranging delays in the circuitry for generating control signal for the multiplexer 25. The resetting of the counter should be delayed sufficiently that the bit latch 337 is able to update correctly. Symbol code data supplied to the symbol decoder 20 has to lag sufficiently that the decision as to whether or not the current data segment requires comb filtering to suppress artifacts of co-channel interfering NTSC is stored in the bit latch 337 before the current data segment passes through the symbol decoder 20. The requisite lag can be achieved by following the ADC 14 with a clocked delay register (not explicitly shown in the drawing figures), but such delay register has to store 10-parallel-bit words.

Alternatively, the decision as to whether or not the current data segment requires comb filtering to suppress artifacts of co-channel interfering NTSC can be based, not on the statistics of the current data segment, but rather on the statistics of the preceding data segment. Delay adjustments can then be made on the logic inverter 336 output signal using, an intermediate clocked bit latch (not explicitly shown in the drawing figures).

Another approach to reducing the delay a clocked delay register has to provide following the ADC 14 is to shorten the interval for deciding on whether comb filtering is needed to suppress artifacts of co-channel interfering NTSC and to clock the bit latch 337 at the close of each shortened interval. The interval for decision can be shortened to one half or one quarter of a data segment period, by way of example.

Further, the statistics for the trellis codes in a current data segment can be separately evaluated and used to control the multiplexer 25 independently for each of the trellis codes.

FIG. 6 shows in more detail the multiplexer control circuitry 33 and the symbol decoder 20 as it appears in certain other embodiments of the invention. In those certain other embodiments the NTSC-artifact-rejection comb filter 22 comprises a clocked delay circuit 223 for differentially delaying the input signal supplied to the symbol decoder 20 by six symbol epochs and a digital adder 224 for additively combining the input signal and the input signal delayed by six symbol epochs. The comb filter comprising the elements 223 and 224 suppresses NTSC artifacts near NTSC video carrier frequency offset from DTV carrier and near NTSC chroma subcarrier frequency offset from DTV carrier, but does not suppress NTSC artifacts near NTSC audio carrier frequency offset from DTV carrier. The matching comb filter 24 for the NTSC-artifact-rejection comb filter 22 comprises a clocked delay circuit 243 for differentially delaying the data-slicer 23 response by six symbol epochs and a digital subtractor 244 for differentially combining the data-slicer 23 response and the data-slicer 23 response delayed by six symbol epochs. The multiplexer control circuitry 33 is similar to that shown in FIG. 5.

FIG. 7 shows a form that the DTV artifacts suppression filter 18 operative in the analog regime can take which comprises a cascade connection of a filter 181 providing a rejection notch at 4.44 MHz, a filter 182 providing a rejection notch at 1.75 MHz, a filter 183 providing a rejection notch at 940 KHz, a filter 184 providing a rejection notch at 772 kHz, and a filter 185 providing a rejection notch at 603 kHz. That is, rejection notches are provided at those frequencies at which DTV artifacts are likely to have the most energy. The filter 183 providing a rejection notch at 940 KHz rejects the artifact generated by the DTV pilot carrier, and the other filters reject frequencies likely to occur because of data being randomized to avoid components close to data carrier.

FIG. 8 shows another form that the DTV artifacts suppression filter 18 operative in the analog regime can take, a lowpass filter 186 with a cut-off frequency of about 0.5 MHz.

FIG. 9 shows one form the DTV artifacts suppression filter 30 operative in the digital regime can take. The FIG. 9 DTV artifacts suppression filter cascades two comb filter sections. The first comb filter section in the FIG. 9 filter uses a digital adder 301 for additively combining the ADC 19 response with that response as delayed six symbol epochs by a clocked delay circuit 302. The second comb filter section in the FIG. 9 filter uses a digital adder 303 for additively combining the adder 301 sum response with that sum response as delayed four symbol epochs by a clocked delay circuit 304. The adder 303 sum response is applied to the squarer 31 as its input signal.

FIG. 10 shows another form the DTV artifacts suppression filter 30 operative in the digital regime can take. The FIG. 10 DTV artifacts suppression filter cascades three comb filter sections. The first comb filter section in the FIG. 10 filter uses a digital adder 301 for additively combining the ADC 19 response with that response as delayed six symbol epochs by a clocked delay circuit 302. The second comb filter section in the FIGURE 10 filter uses a digital adder 303 for additively combining the adder 301 sum response with that sum response as delayed four symbol epochs by a clocked delay circuit 304. The third comb filter section in the FIG. 10 filter uses a digital adder 305 for additively combining the adder 303 sum response with that sum response as delayed two symbol epochs by a clocked delay circuit 306. The adder 305 sum response is applied to the squarer 31 as its input signal.

FIG. 11 shows a variant of the FIG. I radio receiver portions of a DTV receiver which embodies the invention in somewhat different form. Rather than suppressing the DTV artifacts in the composite video signal supplied from the video detector 17 per FIG. 1, by filtering the video detector 17 response after it is generated, the DTV artifacts are removed by filtering IF signal. This is better done in the UHF IF band. FIG. 11 shows the UHF IF amplifier 5 of FIG. 1 being replaced by a UHF IF amplifier 44 driving a surface acoustic wave filter 45, the response characteristic of which is notched by traps for the DTV artifacts at UHF intermediate frequencies. The SAW filter 45 also includes an in-channel sound trap and can provide the roll-off required for matched filtering of NTSC video signal. If the SAW filter 45 does not provide this roll-off, it can be introduced by modifying the VHF IF amplifier 11, or it can be introduced at baseband by suitable frequency-selective filtering of the video detector 17 response.

FIG. 12 shows a variant of the FIG. 4 radio receiver portions of a DTV receiver which embodies the invention in somewhat different form. Both an in-phase synchronous detector and a quadrature-phase synchronous detector are used in the circuitry 46 for synchrodyning NTSC video carrier modulation to baseband; synchrodyning is presumed to be carried out in the digital regime after converting to a final intermediate-frequency band just above baseband, so the final intermediate-frequency can be digitized. Alternatively, synchrodyning NTSC video carrier modulation to baseband can be done in the analog regime, and the responses of an in-phase synchronous detector and a quadrature-phase synchronous detector used for this purpose can be digitized using respective analog-to-digital converters. The response of the quadrature-phase synchronous detector is the Hilbert transform of the artifacts of the DTV signal as they appear in the response of the in-phase synchronous detector. This Hilbert transform phase shifted by 90.degree. at all frequencies (except possibly the lowest at which there should be little response) by inverse Hilbert transform circuitry 47. Addition and subtraction are considered as being alternative forms of linearly combining. The inverse Hilbert transform response of the circuitry 47 is linearly combined with the response of the in-phase synchronous detector in a linear combiner 48 of suitable type to generate a composite video signal with high frequencies boosted to correct levels for application to the rest of the analog TV receiver circuitry. The inverse Hilbert transform response of the circuitry 47 is linearly combined with the response of the in-phase synchronous detector in a linear combiner 49 of suitable type to generate a luminance signal cutting off somewhat above 750 kHz, which luminance signal is free of DTV artifacts. This luminance signal can, if desired, be further filtered using a lowpass filter 50 with a cut-off frequency of 1 MHz or so, as shown in FIG. 12. The luminance signal is then squared by the squaring circuitry 31 for generating an indication of the energy of NTSC co-channel interfering signal during DTV reception.

One skilled in the art will by acquaintance with the foregoing disclosure be enabled to design further embodiments of the invention, and this should be borne in mind when interpreting the scope of the claims which follow. In the claims which follow, the word "said" is used whenever reference is made to an antecedent, and the word "the" is used for grammatical purposes other than to refer back to an antecedent.