Semiconductor Emulation of Vacuum Tubes

This invention relates to vacuum tube amplifier emulation, and to non-linear semiconductor amplifiers.

Tube amplifier emulation does model the tube amplifier in such detailed terms as plate resistance and effects of the screen grid. The plate resistance of a triode is particularly unique since all semiconductors have extensive regions of current limits.

Accurate modeling of the plate resistance is needed when tubes drive non-resistive or non-linear loads. A popular analysis of vacuum tube distortion components lumps the variable gain with the variable output impedance and represent it as a resistive load. This analysis does not hold with reactive or non-linear loads and there are no closed form solutions which would lead to a model.

The object of this invention is a semiconductor emulation for emulating a vacuum tube with one or more of the following characteristics: non-linear output, non-linear transfer, non-linear input, reverse transfer, and second harmonic generation.

Another object is an emulator with a second output to emulate a phase splitter.

Another object is a semiconductor emulator of a cathode follower.

Still another object is a compressor emulating the effects of a power supply droop and screen grid behavior.

Other objects, advantages and novel features of the present invention will become apparent from the following detailed description of the invention when considered in conjunction with the accompanying drawings.

FIGS. 1 and 2 are schematics of non-linear resistor-diode networks.

FIG. 3 is a schematic of a non-linear semiconductor network.

FIG. 4 is a symbol for a non-linear network.

FIG. 5 is a schematic of a triode emulation circuit.

FIG. 6 is a graph of the output characteristics of the circuit of FIG. 5.

FIG. 7 is a schematic of another embodiment of a triode emulation circuit.

FIG. 8 is a schematic of a cathode follower emulator.

FIG. 9 is a graph of the output characteristics of the circuit of FIG. 8.

FIG. 10 is a schematic of a phase splitter emulator for bipolar amplifiers.

FIG. 11 is a schematic of a bipolar amplifier.

FIG. 12 is a graph of the output characteristics of half of the circuit of FIG. 11.

FIG. 13 is a block diagram of a computer embodiment.

FIG. 14 is a schematic of a lamp compressor.

The resistor-diode network of FIG. 1 is described in U.S. patent application Ser. No. 07/466,865, now U.S. Pat. No. 5,133,014 issued Jul. 21, 1992. It is a plurality of parallel resistors 1-5 series diodes 11-14. For input voltages across terminals A and B of less than one diode drop, only resistor 1 conducts. For input voltages between one and two diode drops, resistors 1 and 2 conduct. Higher voltages make more resistors conduct, thereby lowering the dynamic resistance of the network.

The resistor-diode network of FIG. 2 has a plurality of parallel resistor and diode pairs in series. As the current flowing from terminal A to B increases, the voltage across the resistors increases. When the resistor voltage approaches the diode drop, the diode conducts and dynamically removes the resistor from the series string. When all of the diodes conduct, the resistance of the network is the resistance of resistor 25.

There is a rough equivalency between these networks: Equal resistors in FIG. 1 produces a current approximately proportional to the square of the voltage across the terminals. Similarly, if the resistors of FIG. 2 are in the ratios of 1, 1/3, 1/6, 1/10, 1/15 . . . and the last resistor, the nth, is 2/n, then it too produces a current approximately proportional to the square of the voltage across the terminals A and B.

It should be noted that the networks approximate the desired function over a region. The diodes tend to sectionalize the function and eventually all of the diodes are on and the network becomes linear.

FIG. 3 also produces a squared current using semiconductor behavior found in logarithm amplifiers. The voltage across the terminals A and B is converted to a current by resistor 31. This current produces a voltage on the base of transistor 34 proportional to twice the logarithm of the current. The transistor 34 converts that voltage to a current in an exponential manner proportional to the square of the current through resistor 31 or the voltage across the terminals A, B. This is made possible by biasing diode 35 with current source 36.

This non-linear circuit uses an active semiconductor, namely a transistor, to replace many passive semiconductors, diodes.

For brevity in the drawings, a new symbol shown in FIG. 4 will indicate a non-linear network as shown in FIGS. 1, 2, or 3.

The circuit of FIG. 5 approximates a triode. The terminals G, K, and P represent the grid, cathode, and plate respectively. The grid-to-cathode voltage is transferred by amplifiers 41 and 42 to resistor 43. The voltages at outputs 44 and 55 of amplifier 41 and 42 respectively are then proportional to the grid-to-cathode voltage. The components 45-49 form a non-linear inverting amplifier with input resistor 45 and feed-back created by resistor 47 and network 46. Transistor 48 is the amplifier and transistor 49 isolates the feedback from the output circuit. The gain from the grid G to the collector of 48 approximates the .mu. of the triode. Network 52 approximates the plate resistance, capacitor 53 is the plate-to-grid "Miller" capacitance while resistor 54 is a stabilizing resistor. The voltage at 55 scales the non-linear characteristics of diode 56 and resistor 57 to the grid-to-cathode diode characteristics, approximately. By proportioning the gains of amplifiers 41 and 42 to the resistances 45 and 57, any current flowing in diode 56 will produce an equal current into the cathode terminal K.

The optional diode 38 and transistor 39 sense the plate current and feedback a small portion to the grid G, about 0.1 percent, to simulate the negative grid current. This feedback is only important for extremely high input impedance amplifiers, such as capacitor microphone preamplifiers.

The resulting output characteristics are shown in FIG. 6. Although the curves are not precisely like a tube characteristics, they are very close to the same around load line 59. This is further evidenced by similar distortions characteristics and intermodulation spectra.

The output characteristics of FIG. 6 show the upward sweeping shape of a vacuum triode and do not show the current limiting characteristics of pentodes and semiconductors. Even gate 2-to-drain connected dual gate FETs show current limiting.

The networks 46 and 52 preferably produce currents proportional to the square of the voltages across them so the circuit produces second harmonics. Squaring networks as described above are not perfect and produce a desired amount of third harmonics also.

FIG. 7 illustrates another embodiment of a triode emulation circuit which makes the approximation that the plate current does not need to go through the cathode. Amplifier 61 transfers the grid-to-cathode voltage across resistor 62. Resistors 63 and 64 with amplifier 65 form a unity gain inverting amplifier. When the grid-to-cathode voltage is negative, resistor 66 and network 67 form the feedback for amplifier 61. When this voltage is positive, resistor 66 and network 68 forms the feedback under the control of amplifier 65. Network 69 forms the plate resistance. Capacitor 70 is the grid-to-plate "Miller" capacitance while resistor 71 stabilizes the circuit. Resistors 72 and 73 scale the conduction voltage of diode 74 to the grid-to-cathode diode 75. Although this does not pass the grid current to the cathode, it is a good approximation in high impedance grid circuits. Diode 74 can also be replaced or implemented by the base-emitter junction of a PNP transistor with its collector connected to the cathode terminal K. Alternatively, the network 72-74 may be deleted and diode 75 replaced by a Schottkey diode 75.

The circuit of FIG. 7 is biased differently than the circuit of FIG. 5 and produces plate voltages below the cathode voltage. But, the capacitive coupling to the next stage or the rebiasing of the next stage adjusts for the difference. The operation along the load line is similar since the intermodulation spectra is virtually identical to -70 dB. This was done with all of the networks having second order characteristics of squaring to produce second harmonics.

The networks 46, 67 and 68 provide harmonics without clipping. The networks 52 and 69 also provide harmonics without clipping and control the harmonic generation when going into cutoff. Diodes 56, 74 and 75 control the bottom of the waveform shape and consequently the harmonic generation.

Diodes 56, 74 and 75 may be replaced by a diode network.

An analysis of the networks 52 and 69 operating with a load resistor produces a cubic component which is trigonometrically converted to fundamental and third harmonics. The fundamental component is phased to reduce the output. This is one contributor to the compressive nature of vacuum tubes. The gain goes down as the signal is increased.

The circuit of FIG. 8 is a cathode follower emulator with transistor 82 providing gain and optional diode 83 assuring reverse current does not flow. Resistor 81 is proportional to the zero grid voltage plate resistance while resistor 84 is proportional to the inverse of the transconductance.

FIG. 9 is the approximate output characteristic of cathode followers and the circuit of FIG. 8. Line 88 represents the normal operation of the cathode follower. However, when the current becomes excessive, the grid voltage goes positive and conducts heavily. This is emulated by transistor 82 saturating and the base conducting heavily as line 89 shows. These characteristics belie the name of follower, since the circuit produces the same voltage out of it is able, otherwise produces no current or loads the previous stage.

FIG. 10 is an approximation to the phase splitter for a bipolar amplifier which requires both inputs in-phase. Since the two triodes in a differential amplifier phase splitter compensate each other, the stage produces very little distortion until clipping. The output resistance of the phase splitter is about twice the triode plate resistance normally, but becomes nearly infinite when clipping.

When the output of amplifier 91 goes high, network 92 pulls up voltage at 93. When the voltage at 93 approaches the plate voltage P, network 94 becomes more resistive and disconnects when voltage at 93 is greater than the voltage at P. At the same time, network 95 disconnects and the current from source 96 flows through network 97 to plate N. Symmetric behavior occurs when the amplifier 91 output goes negative: network 92 disconnects, P has current from current source 98, network 95 pulls down voltage at 99, and network 97 disconnects from plate N.

The networks 94, 95 and 97 use an extra diode in series with the input to keep reverse currents from flowing. Network 92 does not emulate the low gain of the differential amplifier when the triode opposite the input is cut off.

The circuit of FIG. 10 provides the soft cutoff for the grid circuit of the output state, of the circuit of FIG. 11. FIG. 11 shows a bipolar push-pull output which emulates push-pull pentodes with a poorly regulated supply. Since the negative half operates symmetrically to the positive half, only the positive half will be detailed. Outputs P and N of FIG. 10 are connected to inputs P and N of FIG. 11. Resistor 102 is the plate resistor. Capacitor 103 is the coupling capacitor. Diodes 104 and 105 emulate positive grid conduction. Zener diode 105 adjusts for the nominal zero bias of this stage. Resistor 106 is the grid resistor which drives amplifier 108 with feedback resistor 107. Network 109 is nominally a squaring, second order emulation of the pentode transfer characteristic. This gain varying characteristic provides smooth crossover and the variable gain for emulating tube compression. Amplifier 111, shown as a transistor, shifts the level of the signal to the output supply voltage +40 with the help of resistor 112. MOSFET 113 with source resistor 114 transfers the voltage on resistor 112 to a current through resistor 114. Bias resistor 110 is adjusted to overcome the threshold voltage of MOSFET 113. The remaining bias is established by the voltage on the base of transistor 111. Zener diode 115 allows the load to fly back some before it is clamped.

The components 103-106 form a bias-shifter as do the grid circuit and the grid conduction diodes of the circuits of FIGS. 5 and 7.

Inverting amplifier 120 and non-linear networks 121 and 122 feedback the output to emulate the plate resistance of a pentode. Notice that the feedback loop goes through both non-linear networks. Consequently, the plate resistance and transfer characteristic are functions of both the output and the input. This is seen in the different slopes of curves 141-143 of FIG. 12.

The saturation region 144 is emulated by resistor 114. Again, the entire characteristic is not perfect, but around the load line 140, it is a good approximation.

The poor regulation of the power supply coupled with screen grid operation creates the compression found in tube amplifiers. When the power supply sags under the load of large signals, the screen voltage goes down in a manner dictated by the power supply filter. The drop in screen voltage lowers the output current and lowers the gain of the tube.

The screen grid voltage shift can be lumped into a control grid shift according to Thomas Martin in his book Electronic Circuits, Prentice-Hall, pages 84-87 providing the signal is scaled appropriately.

Although the power supply could drive this circuit, it is simpler to estimate the output current with filter 130. The resulting signal is rectified by 131 and then filtered by 132 which has the same time constants and overshoot as the emulated power supply. The output of 132 is fed to the negative half by resistor 135 while being inverted by 133 and fed to the positive half by resistor 134. An increasing output then reduces the bias on networks 109 and 136, reduces the output currents, increases the resistance of these networks and lowers the gain. The compression control signal from the output of filter 132 is canceled in the output.

This is similar to the behavior of tube compressor/expanders such as the expander, FIG. 16.10, Radiotron Designers Handbook. RCA Corporation, 1954.

The difference lies in the characteristics of the filter 132. Compressors react quickly to rising signals and slower to falling signals. But the power supply reacts similarly in both directions at a medium rate. The filters of compressors are non-linear single pole while the power supply is multipole and may have overshoot.

The input to filter 130 may be taken from the output in a feedback fashion or may be taken from an input for a feed forward operation.

The amplifier is an example of a controlled means, others include voltage controlled amplifiers and photo-resistive devices.

FIG. 14 is an example of a lamp controlled compressor. The lamp 151 has a positive resistance change with temperature. A large signal on the input heats the lamp filament increasing its resistance and consequently the attenuation of the output as appears across resistor 152. The thermal characteristics of the lamp or other device with a high resistance temperature coefficient is a filter producing heat and the heat produces a resistance change which changes the ratio of input to output. By picking the right lamp or resistance temperature coefficient means properly, a good approximation to power supply response can be achieved. By picking the proper resistance, the proper compression of 1 or 2 dB can be easily achieved.

In addition to the triode configuration illustrated in FIGS. 5 and 7, a pentode emulation is achieved in FIG. 11. A control grid input would be on resistor 106, the screen grid input would be on resistor 134, the plate produces current through 114.

Real-time computer emulation of tube circuits is possible since the mathematical quantities of operational amplifiers, resistors, capacitors, and non-linear networks are readily established. See my U.S. Pat. No. 4,995,084 which is incorporated herein by reference. FIG. 13 shows the input acquired by analog-to-digital converter 151, processed by computer 152, and output by digital-to-analog converter 153.

Although the present invention has been described and illustrated in detail, it is to be clearly understood that the same is by way of illustration and example only, and is not to be taken by way of limitation. The spirit and scope of the present invention are to be limited only by the terms of the appended claims.