Discrete Step Microwave Attenuator

1. Field of the Invention

The invention pertains to the field of microwave attenuators and more particularly to matched microwave variable attenuation in monolithic microwave integrated circuits.

2. Description of the Prior Art

Microwave attenuators of the prior art provide attenuation in either discrete step attenuation or continuously variable. The continuously variable attenuators have one of two classic configurations; the resistive Pi circuit and the resistive T circuit. These attenuators utilize either PIN diodes or field effect transistors (FETs) for the series and shunt resistors. PIN diodes and FETs exhibit resistive changes with properly applied DC voltages and thus are useful as variable resistors. The resistive values of these Pi and T circuits for all levels of attenuation are chosen to provide an impedance that matches the impedance of a transmission line, or another microwave device, to suppress reflections in the system. To accomplish this, the ratio of the shunt and series resistors must change with attenuation changes, establishing a functional relationship of the ratio versus attenuation which is extremely non-linear. This presents a very difficult tracking problem, requiring that the dc characteristics of the PIN diodes or FETs utilized in the attenuators be matched over the entire attenuation range. As a result, the PIN diodes and FETs are generally controlled with separate power supplies. Though control circuitry can be provided to supply the DC voltages to the voltage controlled resistances in their proper functional relationship from a single power supply, such circuitry requires much more real estate than the attenuator it controls and is therefore rejected for most applications.

The problem of providing the proper ratios to maintain a constant characteristic impedance for the Pi and T circuits is exacerbated by the non-uniformity of the PIN diode and the FET characteristics that result with present day manufacturing processes. For example, the equivalent resistance value of a FET is a function of the pinch-off voltage, that voltage which must be exceeded by the gate voltage for current to flow in the FET. Present day manufacturing processes, however, yield FETs with pinch-off voltages that vary substantially. Since the resistance of the FET is a function of the pinch-off voltage, FETs exhibit resistance values having varying functional relationships of the gate voltage. Thus, for each attenuator a process is encountered for selecting three FETs with equal resistance versus gate voltage characteristics, greatly increasing cost of the attenuators.

Further, resistive Pi and T circuits cannot simultaneously realize low (off) state insertion loss and a large dynamic attenuation range with the variable resistors presently available. For both circuits a low insertion loss requires a low resistance value for the series elements and a high resistance value for the shunt elements. As attenuation increases from the minimum value the series resistance increases, while the shunt resistance decreases. Since the shunt resistances start at opposite ends of the functionality curve it is extremely difficult to provide the ratio of series resistance to shunt resistance required for the attenuation values desired and simultaneously maintain a constant characteristic impedance for the circuits.

Additionally, at high frequencies, the internal capacitances of the PIN diodes and FETs establish complex characteristics for the Pi and T circuits. To provide real characteristic impedances it is necessary to resonate these capacitances by shunting inductors across the elements of the Pi and T circuits. These resonant circuits severely limit the operating bandwidth of the attenuator.

An attenuator which provides improved performance over the Pi and T circuits is disclosed in U.S. Pat. No. 4,970,478 issued to the assignee of the present invention. This patent discloses a variable microwave attenuator which includes a plurality of ladder circuits (cells) each having a series inductance and a shunt circuit comprising a capacitor and a variable resistor in parallel. The cells are cascaded in a manner to establish an artificial transmission line with distributed loss provided by the variable resistor shunt elements. These variable resistor shunt elements may be realized by utilizing FETs which exhibit resistive changes of voltage applied to their gates. The series inductance, shunt capacitance, and shunt variable resistors are chosen to establish an impedance for the artificial line that is substantially independent of the shunt resistance value and to provide a low reflection coefficient with its concomitant low voltage standing wave ratio (VSWR). When cascaded, the internal ladder sections combine to form lossless symmetrical Pi cells with a variable resistor positioned between each cell. Symmetry of the artificial line may be completed with the addition of a shunt capacitor at one end of the artificial line to establish lossless symmetrical Pi end sections for the transmission that are identical to the internal lossless symmetrical Pi sections formed by cascading the ladder networks.

Though the artificial line of U.S. Pat. No. 4,970,478 provides variable attenuation by the adjustment of but one resistance value per cell and may provide a characteristic impedance which is independent of the shunt resistance value, such performance is difficult to achieve. Further, due to the resistance variation of PIN and FETs, previously described, a variable attenuator which provides attenuation with reasonable precision requires extensive calibration, adding appreciable cost to the device.

Another prior art attenuator which minimizes the problem of maintaining the characteristic impedance with attenuation variations is disclosed in U.S. Pat. No. 5,109,204 entitled "High Power RF Precision Attenuator" issued to Lyndon M. Keefer on Apr. 28, 1992 and assigned to the assignee of the present invention. This patent discloses a variable microwave attenuator utilizing a first pair of isolated quadrature hybrid ports as the input and output ports of the attenuator and providing the second pair of isolated ports with switchably coupled resistive terminations, the resistance at each termination being equal and selected in accordance with the attenuation desired. Since these resistors are not matched to the hybrid's characteristic impedance the portion of the signals incident to the terminated ports, that are not dissipated in the resistive termination, are reflected and coupled in the hybrid in a manner to be out-of-phase at the input port, thus cancelling thereat, and to be in-phase at the output port, thus adding thereat to provide the attenuated output signal. A throughput without attenuation is provide by shunting diodes across the terminating resistors. When these diodes are in the conducting state, short circuits are established which reflect the entire signal to the output port. This arrangement is wasteful of energy, requiring dc power whether or not the device is attenuating microwave signals. Further, conducting diodes do not provide complete short circuits. Thus some attenuation is realized in the throughput state.

Another variable attenuator of the prior art provides attenuation by switchably coupling resistors of equal value across the two output ports of a quadrature hybrid circuit, the output ports are then coupled to the input ports of a second quadrature hybrid. One input port of the first hybrid and one output of the second hybrid are terminated with a resistor having a resistance equal to the characteristic impedance of the hybrid to absorb power coupled to these ports. The remaining input port of the first hybrid and the remaining output port of the second hybrid serve as the input and output ports of the attenuator, respectively. Attenuation control is accomplished with switching diodes in series with the attenuator resistors. Each of these diodes are independently controlled through respective driver circuits directly connected to the associated diode. The attenuator therefore requires an appreciable number of circuit elements for the driver assembly and appreciable switching power to effectuate an attenuation change.

In accordance with the principles of the present invention, a programmable variable microwave attenuator includes first and second printed branchline couplers (quadrature hybrid couplers) constructed on a common substrate. An output set of isolated ports of the first branchline coupler is coupled to an input set of isolated ports of the second branchline coupler. One of the isolated input ports of the first coupler is terminated with a matched impedance while the second input port serves as the RF input to the device. In like manner one output port of the second coupler is matched terminated while the other serves as the output port of the attenuator. Shunt resistors, which determine the attenuation of the device, are coupled at the junction of the two branch lines through switching diodes. Switching is accomplished by a single driver circuit coupled to all the switching diodes via the branchline coupler. This driver establishes three diode states; all diodes off, two (2) diodes forward biased, and alterante pair of diodes forward biased. Each of these diode states provides a respective attenuation value. The all diodes off state permits a signal to pass through the device unattenuated while the other two states are associated with respective attenuations. RF phase shift and insertion loss are minimized by positioning appropriate capacitors in series with each resistor-diode combination to tune the inductance associated with these combinations. These tuning capacitors are by passed with low susceptance shunt inductors to permit the flow of dc current from the driver to the diodes.

The aspects and advantages of the invention will be understood more fully from the following description of the preferred embodiment thereof, which is by way of example only, with reference to the accompanying drawings.

FIG. 1 is a schematic diagram of a preferred embodiment of the invention.

FIG. 2 is a schematic diagram of a resistor shunted transmission line.

FIGS. 3A-3C are a microstrip configuration of the embodiment of the invention represented by the schematic diagram of FIG. 1.

A digital programmable attenuator, generally designated 10, using three stages of the invention, each providing digitally selectable attenuations, is shown schematically in FIG. 1. The three stages are substantially identical, differing only in the value of the resistors positioned between the branch line couplers (3 dB quadrature hybrids) which determine the attenuation values of the attenuation stage. Therefore, the invention will be explained with reference to the input stage 11.

In accordance with the invention, the output ports H1a and H1b of a first 3 dB quadrature hybrid H1 are respectively coupled to input ports H2a and H2b of a second 3 dB quadrature hybrid H2 via transmission lines 12 and 13. Shunted across the transmission line 12 are resistors R1a and R2a which are selectable through PIN diodes D1a and D2a, respectively. Capacitors C1a and C2a are also positioned in series with the resistor-diode combinations R1a-D1a and R2a-D2a, respectively, to tune out the inductive reactance inherent in resistors and diodes at radio frequencies. The series resistor-diode-capacitor circuit minimizes the phase shift and excess insertion loss, due to unwanted reflections, but does not provide a dc path to ground for the diode bias. This dc path is established by low susceptance inductors L1a and L1b coupled in parallel with the capacitors C1a and C2a, respectively. The inductance value of the inductors L1a and L2a is chosen to provide a resonance with the capacitors C1a and C2a that is well outside of the frequency band of interest while providing an effective rf choke and dc short circuit. Similarly, shunted across transmission line 13 are resistors R1b and R2b, diodes D1b and D2b, inductors L1b and L2b, and capacitors C1b and C2b. The elements shunted across transmission line 13 are arranged in the same configuration and are of the same value as their counter parts shunted across the transmission line 12.

The branch line couplers H1 and H2 are constructed as microstrip circuits in manner to provide a continuous dc path from a control terminal 14 to the four diodes D1a, D1b, D2a, and D2b. Capacitors 15 and 16 are coupled between attenuator stages to block the dc control signals of one stage from affecting the shunt resistor selection of another stage of the attenuator.

The diodes D1a and D1b have the same polarity as do the diodes D2a and D2b. The polarities of diodes D1a and D2a are opposite, as are the polarities of diodes D1b and D2b. Thus, for example, diodes D1a and D1b conduct, shunting resistors R1a and R1b across transmission lines 12 and 13, respectively, and diodes D2a and D2b are cut-off when a voltage -V is applied to the control terminal 14. Conversely, diodes D1a and D1b are cut-off and diodes D2a and D2b conduct, shunting resistors R2a and R2b across transmission lines 12 and 13, respectively, when a voltage +V is applied to control terminal 14. When zero (0) voltage is at control terminal 14 all diodes are cut-off, thus eliminating all shunt resistors across the transmission lines 12 and 13. Consequently, three resistive values may be shunted across each transmission line; O, (R1a,R1b), and (R2a,R2b). Inductance 17 and capacitor 18 are choke and by-pass elements, respectively, which isolate the dc control terminal from the rf signals applied to the attenuator circuit.

A transmission line of characteristic impedence R.sub.o shunted by a resistor R is shown schematically in FIG. 2. A signal having a voltage V.sub.inc at the input port 17a,17b provides an input power P.sub.inc equal to (V.sub.inc).sup.2 /R.sub.o to the transmission line 18. This power is split into three components by the shunt resistor R: P.sub.d equal to 4RR.sub.o P.sub.inc /(2R+R.sub.o).sup.2 representative of the power dissipated in the resistor R; P.sub.ref equal to 4R.sub.o 2P.sub.inc /(2R+R.sub.o).sup.2 representative of the power reflected due to the coupling of the resistor R across the transmission line 18; and P.sub.t equal to 4R.sup.2 P.sub.inc /(2R+R.sub.o).sup.2 representative of the power transmitted to the output port 19a,19b. Thus the resistance shunted transmission line provides an insertion loss IL=P.sub.t /P.sub.inc =4R.sup.2 /(2R+R.sub.o).sup.2. This insertion loss, however, is due to reflection caused by the insertion of the resistor and dissipation in the resistor. In addition to the insertion loss, a phase shift, dependent upon the length of the transmission 18, exists between the input port 17a,17b and the output port 19a,19b. This phase shift is unchanged when the purely resistive shunt impedance is switchably removed from across the transmission line 18. The shunted transmission line shown in FIG. 2 is not a true attenuator. A true attenuator has only dissipative loss.

A phase shift, dependent upon the length l of the transmission line 18 and the reactance of the shunting impedance, exists between the input port 17a,17b and the output port 19a, 19b. This phase shift maybe eliminated when the shunt impedance is purely resistive by providing a length l that is an integer multiple of the signal wavelength.

Positioning two resistance shunted transmission lines between two 3dB quadrature hybrids as shown in stage 11 and other stages of FIG. 1 establishes a true attenuator between the defined input and output ports of each stage and the overall attenuator. Assume a signal of voltage V.sub.inc is incident to input port 21 and coupled through dc blocking capacitor 22 to port H1c of branch line coupler H1. This signal is split by the hybrid to couple a signal to the port H1a which is equal V.sub.inc /.sqroot.2 and a signal to the port H1b which is equal to jV.sub.inc /.sqroot.2. The signals at the ports H1a and H1b are respectively coupled to transmission lines 12 and 13 wherefrom signals equal to .GAMMA.V.sub.inc /.sqroot.2 and j.GAMMA.V.sub.inc /.sqroot.2 are respectively reflected back to ports H1a and H1b. Those skilled in the art should recognize that .GAMMA. is equal -R.sub.0 /(2R+R.sub.0). The reflected signal at port H1a is split to provide a signal at port H1c which is equal to .GAMMA.V.sub.inc /2 and a signal which is equal jV.sub.inc /2 at a fourth port H1d of quadrature hybrid H1. Similarly, the reflected signal at the port H1b is split to provide a signal at port H1c which is equal to -.GAMMA.V.sub.inc /2 and a signal at port H1d which is equal to j.GAMMA.V.sub.inc /2. Consequently, the reflected signals at the input port H1c cancel so that no reflected signal exists thereat and the signals at port H1d add to establish a signal that is equal to j.GAMMA.V.sub.inc. This reflected signal is coupled through a dc blocking capacitor 23 to a matched termination 24 wherein the reflected signal is absorbed. For simplicity of presentation the signal phase shift due to the propagation path length has been neglected. In accordance with the invention the path lengths to the shunt circuits on the transmission lines 12 and 13 are equal. Thus, equal phase terms will be added to all the signals and the basic results are not altered.

As stated above the capacitors C1a, C2a, C1b, and C2b, are selected to tune out the inductance inherent in the resistor-diode combinations respectively in series with the capacitors. Assume that the bias voltage at the control terminal 14 is -V so that the equal resistors R1a and R1b are selected as the shunt resistors on the lines 12 and 13, respectively. In view of the above, the voltage across the shunt resistor R1a, which those skilled in the art will recognize to be (1+.GAMMA.)V.sub.inc /.sqroot.2, is coupled to port H2a of branch line hybrid H1 and is split to couple signals having voltages equal to (1+.GAMMA.)V.sub.inc /2 and j(1+.GAMMA.)V.sub.inc /2 to ports H2c and H2d, respectively. Similarly, the voltage j(1+.GAMMA.)V.sub.inc /.sqroot.2 across the resistor R1b is coupled to port H2b and split to couple signals having voltages equal to -(1+.GAMMA.)V.sub.inc /2 and j(1+.GAMMA.)V.sub.inc /2 to ports H2c and H2d, respectively. From the above it is evident that the signals at port H2c cancel and that the signal at port h2d has a voltage that is equal to j(1+.GAMMA.)V.sub.inc. This signal is coupled from the port H2d to a transmission line 25 of characteristic impedance R.sub.0 at a power level that is equal to Pinc (2R1a).sup.2 /(2R1a+R.sub.0).sup.2, thereby establishing an attenuator which provides an attenuation equal to the insertion loss of one of the transmission lines, with shunt resistances (R1a,R1b), coupled between the branch line couplers H1 and H2.

A microstrip configuration for the schematic diagram of FIG. 1 is shown in FIGS. 3A-3C, wherein reference numerals previously assigned indicate corresponding elements in the two figures. For purposes of clarity only a selected number of elements have been referenced. All other elements may be readily identified with the aid of the reference numerals provided. It is apparent from FIGS. 3A-3C that a dc voltage at the control terminal 14 is coupled through the microstrip circuity to bias the diodes as previously explained.

While the invention has been described in its preferred embodiments, it is to be understood that the words which have been used are words of description rather than of limitation and that changes within the purview of the appended claims may be made without departure from the true scope and spirit of the invention in its broader aspects.