AC Motor Drive System With a Two Phase Power Supply

This invention relates in general to induction motor drive systems and, more particularly, to an induction motor drive system for operating a single-phase, two winding induction motor from a two-phase power supply circuit and including a direction reversing capability.

A conventional split-phase capacitor start or capacitor run single phase induction motor, also known in the art and referred to hereinafter as a permanent split capacitor (PSC) motor, has two stator windings, a "main" winding and a "start" winding. FIG. 1 illustrates an exemplary PSC motor 100 that includes a main winding 102 and a start winding 104 that are commonly connected at one end. Main winding 102 and start winding 104 are mounted in the stator (not shown) of motor 100 and spatially separated from each other by an angle related to the rated speed of motor 100, e.g., 90.degree. for a two pole, 3600 RPM motor, as is well known in the art. Such PSC motors are designed to be operated with a run capacitor, such as a run capacitor 106, connected in series with start winding 104. It is a typical practice in the industry for the motor manufacturer to not supply the run capacitor with the motor, but instead to only specify parameters of the capacitor, e.g., capacitance and power rating, sufficient to enable a user to procure and install the capacitor.

In the operation of PSC motor 100, main winding 102 and the series combination of start winding 104 and run capacitor 106 are connected in parallel with each other and directly across a single phase power source 110. Since start winding 104 is energized through capacitor 106, the phase angle of the current flowing through start winding 104 is shifted with respect to the current flowing through main winding 102, such that the phase angle between the respective currents flowing in windings 102 and 104 is approximately 90.degree. while the motor is running. The phase angle between the currents in windings 102 and 104 and the spatial separation of those windings result in the creation of a rotating magnetic field which is inductively coupled to the rotor (not shown) of motor 100, to exert a rotational force on the rotor.

The rotor of motor 100 attempts to rotate in synchronism with the rotating magnetic field but lags the rotating magnetic field by a "slip" factor, resulting in a torque on the rotor which is in part proportional to the amount of slip.

The starting torque exerted on the rotor of motor 100 during a starting period when motor 100 is started and accelerated to rated speed is also proportional to the sine of the phase angle between the currents flowing in windings 102 and 104. Therefore, in order to maximize the starting torque, it is necessary to achieve a phase angle of 90.degree. during starting. However, the starting torque for a single phase PSC motor, such as motor 100, is generally poor because the specified parameters of the run capacitor are only optimized for running conditions, not starting conditions. Thus, the capacitance of run capacitor 106 is specified by the manufacturer based on the impedances of windings 102 and 104 that will be experienced during running of motor 100, rather than during starting. However, as known in the art, the apparent values of motor winding impedances vary during the starting period of a PSC motor and are therefore different during starting than during running. As a result of the capacitance of capacitor 106 being optimized for running and not for starting, its magnitude is too small for starting. This results in the phase angle between the currents flowing in windings 102 and 104 being less than 90.degree. during the starting period and the starting torque being less than a maximum possible starting torque.

One solution known in the art to compensate for the insufficient magnitude of capacitor 106 during starting is to connect a starting capacitor 112 across capacitor 106 to increase the total capacitance in series with start winding 104 and thereby increase the current flowing in the start winding, the phase angle and the starting torque of motor 100. Starting capacitor 112 is disconnected, e.g., by means of a centrifugal switch, positive temperature coefficient thermistor (PTC device), or relay, once the motor has reached running speed. Disadvantageously, although operation of starting capacitor 112 generally improves the starting torque of motor 100, its use still does not maximize torque throughout the starting period of motor 100. Ideally, the magnitude of the capacitance in series with start winding 104 would have to be continuously varied during the start period to maintain a desired phase angle while the respective impedances of windings 102 and 104 vary.

Conventional single phase PSC motors are commonly used in heating, ventilating and air-conditioning (HVAC) systems to drive system loads such as fans, pumps and compressors. HVAC systems are subject to widely varying demand cycles due to a variety of factors such as, for example, daily and seasonal fluctuations of ambient temperature, human activity in the controlled environment, and intermittent operation of other equipment in the controlled environment. Accordingly, in order to assure a satisfactory temperature of the controlled environment, the HVAC system must have the heating and/or cooling capacity to accommodate "worst case" conditions. As a result, under less than worst case conditions the HVAC system has a significant over-capacity and is necessarily operated at reduced loading. Since the maximum operating efficiency of a motor, such as a PSC motor, is normally obtained only when the motor is operating at full load, the reduced HVAC system load results in inefficient operation of the motor. Further, to the extent that motors are required to cycle on and off to meet HVAC load requirements less than the capacity of the HVAC system, further significant operating inefficiencies are experienced. Such further inefficiencies include the operating cost of frequently starting motors. A reduction in the useful life of such motors also results from the well known thermal and mechanical stresses existing in this field.

A solution for overcoming the above inefficiencies resulting from the excessive capacity of an HVAC system is to vary the system capacity to meet the demand on the system. One method for varying HVAC system capacity is by varying the speed of the motors driving the HVAC system loads in accordance with the demand. With respect to HVAC system loads driven by single phase motors, such as PSC motors, in order to effect a desired motor speed control, it would be necessary to vary the frequency of the single phase power supplied to the motor. However, with respect to PSC motors, the run capacitor, e.g., capacitor 106 of motor 100 (FIG. 1), is optimized for a particular set of running conditions, including operation at a nominal frequency, e.g., 60 Hz. As a result, operation of a PSC motor at other than the nominal frequency results in production of less than optimal torque and inefficient operation. While some applications may exist in which very limited speed control of a PSC motor is achieved by a small variation of the single phase source frequency, such variation from the nominal frequency results in less efficient operation since the motor is nevertheless designed for optimum performance at the nominal source frequency.

A conventional implementation of varying motor speed to modulate HVAC system capacity typically requires a two- or three-phase motor supplied with two- or three-phase power, respectively. The use of such polyphase motors and power supplies enables variation of motor speed by varying the frequency of the voltage applied to the motor while maintaining a constant volts/frequency (volts/hertz) ratio. Maintenance of a constant volts/hertz ratio corresponds to maintenance of an approximately constant air gap flux and efficient motor operation while delivering rated torque. The use of polyphase motors also offers several other advantages over that of a single phase motor such as, for example, lower locked rotor currents, higher starting torque, lower full load currents and improved reliability due to elimination of the start and/or run capacitor which are required in single phase motors. Disadvantageously, such polyphase motors are more expensive than single phase motors having the same horsepower rating.

Such applications employing polyphase motors generally require provision of variable frequency polyphase power from either a single phase or polyphase line source by means of a power supply circuit, including a polyphase inverter, coupled between the motor and the line source. One drawback to this arrangement occurs in the event that the power supply circuit fails and it is not possible to connect the polyphase motor directly to the line source, such as, for example, when a three-phase motor is driven by an inverter which receives power from a single phase line source. Failure of the power supply circuit therefore results in failure and unavailability of the system utilizing the polyphase motor.

Previous attempts to address the problem of backup power for polyphase motors fed from a single phase power source have required inverter redundancy or additional circuit means for temporarily directly connecting the polyphase motor to the single phase power source. However, the additional circuit means required to "simulate" polyphase power may not provide truly polyphase power and therefore may not drive the polyphase motor at optimum efficiency.

Further, in a few applications it has been desirable to provide a single phase motor which can easily reverse its direction. Direction reversal could be very useful in such applications as dishwashing machines, clothes washing machines, fans, and blowers, for example.

Conventionally, direction reversal is most easily achieved in DC motors by simply reversing the current flow in either the stator coil or armature coil. However, in single phase AC motors, direction reversal is not so simple. In the conventional single phase motor 100 shown in FIG. 1, run capacitor 106 and start capacitor 112 cause a negative phase differential between main winding 102 and start winding 104. In other words, the phase angle of main winding 102 "lags" that of start winding 104. As is well known, the phase relationship between the main and start windings determines the direction of rotation of the motor. Motor direction may be reversed by changing the phase relationship between the main and start windings from a lagging to a leading relationship, or vice versa.

In conventional motor 100, a leading phase relationship may be achieved by connecting the run capacitor 106, and during starting start capacitor 112, in series with the main winding 102, rather than in series with start winding 104. This will achieve the desired direction reversal. However, this approach suffers from many difficulties. For example, the main winding and start winding often have different characteristics, such as current carrying capability and impedance in many conventional motors. In such motors, moving the run and start capacitors to the main winding will cause inappropriate voltages to be applied to and currents to flow through the main and start windings, thereby reducing efficiency and performance. Furthermore, due to the typically high currents flowing through main winding 102 and start winding 104 during motor operation, an automatic switching system (i.e., a relay) would necessarily be undesirably large and expensive. Changing the connection of these capacitors from start winding 104 to main winding 106 or altering the internal connections of the motor is often impracticable in most applications and will result in impaired performance for many types of motors.

Finally, conventional motor 100 may suffer from the drawback of being acoustically noisy or of producing too much vibration for a desired application. Such noise and vibrations are primarily caused by the rotation of the rotor and by the fluctuating magnetic fields created by the windings of the motor.

It is an object of the present invention to provide an AC motor drive system which overcomes the aforementioned problems and disadvantages of conventional drive systems. To achieve the objects and in accordance with the purpose of the invention, as embodied and broadly described herein, the invention is directed to an AC motor drive system for driving an AC motor having a first winding and a second winding, the first winding having first and second ends and a first winding impedance and the second winding having first and second ends and a second winding impedance that is greater than the first impedance. The drive system comprises two-phase power supply means for converting an inputted power supply voltage into a two-phase AC output voltage. The two-phase power supply means has a common terminal, a first phase output terminal, and a second phase output terminal. The common terminal of the two-phase power supply means is adapted for connection to the respective first ends of the first and second windings. The first and second phase terminals of the two-phase power supply means are adapted for connection to the second ends of the first and second windings, respectively. The two-phase power supply means includes power switching means for selectively converting the inputted power supply voltage to the two-phase output voltage, and voltage control means for generating switching control signals for controlling the operation of the power switching means. The voltage control means includes oscillator means for generating an oscillating signal having a frequency representative of a desired operating speed of the motor. The voltage control means further includes waveform signal generating means, responsive to the oscillating signal, for generating a first sinusoidal waveform signal and a second sinusoidal waveform signal. The first and second sinusoidal waveform signals each have a frequency determined by the oscillating signal frequency and have a predetermined phase angle difference therebetween. The waveform signal generating means may include direction control means, responsive to a direction control signal, for selectively changing the predetermined phase angle difference between the first sinusoidal waveform signal and the second sinusoidal waveform signal. The voltage control means also includes means, coupled to the waveform signal generating means to receive the first and second sinusoidal waveform signals, for providing sinusoidally weighted pulse width modulated (PWM) switching signals as the switching control signals to control the operation of the power switching means. The two-phase power supply means converts the inputted power supply voltage to a two-phase voltage provided on the first, second, and common terminals to selectively drive the motor in either direction.

In accordance with another embodiment of the invention similar in some respects to the first embodiment, the voltage control means of the AC motor drive system includes oscillator means for generating an oscillating signal having a frequency representative of a desired operating speed of the motor. The voltage control means further includes waveform signal generating means, responsive to the oscillating signal, for generating a first, a second, and a third sinusoidal waveform signal. The first, second, and third sinusoidal waveform signals each have a frequency determined by the oscillating signal frequency, and the first and second sinusoidal waveform signals have a predetermined phase angle difference therebetween. The waveform signal generating means include direction control means, responsive to a direction control signal, for selectively changing the predetermined phase angle difference between the first sinusoidal waveform signal and the third sinusoidal waveform signal. The voltage control means also includes means, coupled to the waveform signal generating means to receive the first, second, and third sinusoidal waveform signals, for providing sinusoidally weighted pulse width modulated (PWM) switching signals as the switching control signals to control the operation of the power switching means. The two-phase power supply means converts the inputted power supply voltage to a two-phase voltage provided on the first, second, and common terminals to selectively drive the motor in either direction.

In accordance with yet another embodiment of the invention, there is provided an AC motor drive system for connection to a single-phase AC power supply having a line conductor and a neutral conductor and for driving an AC motor having a first winding and a second winding. The first winding has first and second ends and a first winding impedance, and the second winding has first and second ends and a second winding impedance that is greater than the first impedance. The drive system comprises two-phase power supply means, including a line terminal for coupling to the line conductor of the single-phase power supply, for converting the single-phase power supply into a two-phase output voltage. The two-phase power supply means has a first phase output terminal and a second phase output terminal. The drive system comprises a neutral terminal for connection to the neutral conductor of the single-phase power source and for connection to the respective first ends of the first and second windings. The first and second phase terminals are for connection to the second ends of the first and second windings, respectively. The two-phase power supply means includes power switching means for selectively converting the inputted power supply voltage to the two-phase output voltage, and voltage control means for generating switching control signals for controlling the operation of the power switching means. The voltage control means includes waveform signal generating means for generating a first sinusoidal waveform signal and a second sinusoidal waveform signal to have a predetermined phase angle difference therebetween and a common frequency that corresponds to a desired operating speed of the motor. The waveform signal generating means includes direction control means, responsive to a direction control signal, for selectively changing the predetermined phase angle difference between the first sinusoidal waveform signal and the second sinusoidal waveform signal. The two-phase power supply means further includes first means, coupled to the waveform signal generating means, for providing a first sinusoidally weighted pulse width modulated (PWM) switching signal, and second means, coupled to the waveform signal generating means, for providing a second sinusoidally weighted pulse width modulated (PWM) switching signal. The first and second PWM signals operate as switching control signals to control the operation of the power switching means. The two-phase power supply means converts power received from the single-phase power supply to a two-phase voltage to selectively drive the motor in either direction.

Additional objects and advantages of the invention will be set forth in part in the description which follows, and in part will be obvious from the description, or may be learned by practice of the invention. The objects and advantages of the invention will be realized and attained by means of the elements and combinations particularly pointed out in the appended claims.

The accompanying drawings, which are incorporated in and constitute a part of this specification, illustrate several embodiments of the invention and together with the description, serve to explain the principles of the invention.

FIG. 1 is a schematic diagram showing a conventional PSC motor coupled to a single-phase power supply;

FIG. 2 is a schematic diagram illustrating a motor coupled to a two-phase power supply in accordance with an embodiment of the present invention;

FIG. 3 illustrates a motor drive system constructed in accordance with an embodiment of the present invention;

FIG. 4(A) illustrates a power switching device included in an inverter circuit shown in FIG. 3;

FIG. 4(B) illustrates an alternate power switching device included in the inverter circuit shown in FIG. 3.

FIGS. 5(A) and 5(B) are phasor diagrams illustrating voltages produced by the inverter circuit of FIG. 3;

FIG. 5(C) is a phasor diagram illustrating voltages produced by the circuit shown in FIGS. 9(A) and 9(B).

FIG. 6(A) is a block diagram of a first embodiment of a circuit for controlling the operation of the inverter circuit of FIG. 3;

FIG. 6(B) is a block diagram of a second embodiment of a circuit for controlling the operation of the inverter circuit of FIG. 3;

FIG. 7 is a graphical illustration of voltage waveforms produced during operation of the circuits shown in FIGS. 3 and 6(A) and 6(B);

FIGS. 8 (A) and 8(B) are graphical illustrations of voltage waveforms produced during operation of the inverter circuit shown in FIG. 3;

FIG. 9(A) is a block diagram of a third embodiment of a circuit for controlling the operation of the inverter circuit of FIG. 3;

FIG. 9(B) is a block diagram of a fourth embodiment of a circuit for controlling the operation of the inverter circuit of FIG. 3.;

FIG. 10 is a graphical illustration of voltage waveforms produced during operation of the circuit shown in FIG. 9;

FIG. 11 illustrates a motor drive system constructed in accordance with another embodiment of the present invention; and

FIG. 12 is a block diagram of a control circuit for use in the system shown in FIG. 11.

Reference will now be made in detail to the present preferred embodiments of the invention, examples of which are illustrated in the accompanying drawings. Wherever possible, the same reference numbers will be used throughout the drawings to refer to the same or like parts.

In accordance with illustrated embodiments of the present invention, an induction motor drive system is provided in which a two winding, single-phase induction motor, such as a PSC motor, is coupled to a two-phase power supply circuit such that the two motor windings are respectively connected to the two-phases of the power supply circuit. The power supply circuit is configured and operated to provide voltages that result in achieving an optimum phase angle between the motor winding currents during starting and running of the motor. The power supply circuit is further configured and operated to provide variable speed operation of the single-phase motor while maintaining a constant volts/hertz ratio at all operating speeds. The power supply circuit is additionally configured and operated to enable operation of the motor in either direction of rotation.

Referring now to the drawings, FIG. 2 diagrammatically illustrates a two winding, single-phase induction motor 150 coupled to a two-phase power source provided in accordance with an embodiment of the present invention. Motor 150 includes two stator windings, i.e., a main winding 152 and a start winding 154 that are commonly connected at a common winding node 156. Windings 152 and 154 of motor 150 are connected across a first phase voltage V.sub.P1 and a second phase voltage V.sub.P2 of a two-phase power source. The two-phase power source is preferably configured and operated to provide phase voltages V.sub.P1 and V.sub.P2 with a desired phase angle therebetween, for example +90.degree., that results in optimum performance of motor 150.

A feature of the present invention is the ability to conveniently and inexpensively provide for motor direction reversal. As further explained with regard to the several disclosed embodiments, the present invention achieves this feature by reversing the phase relationship between first phase voltage V.sub.P1 and second phase voltage V.sub.P2. For example, changing the phase angle between V.sub.P1 and V.sub.P2 from +90.degree. (a leading relationship) to -90.degree. (a lagging relationship), or vice versa, will desirably reverse the motor direction. Of course, phase angles other than .+-.90.degree. could be considered depending upon optimum performance and efficiency conditions.

It is preferred that this phase reversal between V.sub.P1 and V.sub.P2 be performed after the rotor has ceased rotating to prevent unnecessarily high voltage, currents, and electromagnetic fields from being created in the motor, and to reduce the mechanical stresses caused by direction reversal.

Motor 150 is preferably provided as a conventional PSC motor that does not include a run capacitor. As a result, in accordance with the features of conventional PSC motors, winding 154 is provided with a smaller conductor size and has a greater number of winding turns than winding 152, so that the impedance of winding 154 is greater than that of winding 152. Another characteristic of conventional PSC motors is that during operation from an AC source, a larger voltage is applied across the start winding than across the main winding when the rotor rotates in its conventionally intended direction. This difference in winding voltages results, in part, from the connection of the run capacitor in series with the start winding. For example, with respect to a conventional 230 VAC PSC motor connected through a run capacitor to a single-phase 230 VAC, 60 Hz source, the voltage applied across the main winding would be approximately 230 V, while the voltage applied across the start winding could be on the order of 20% higher or 276 V.

In view of this difference between winding voltages, it is a feature of the illustrated embodiments of the present invention that the two-phase power source, to which motor 150 is connected, be configured and operated to provide V.sub.P2 >V.sub.P1. The relative magnitudes of phase voltages V.sub.P1 and V.sub.P2 are preferably selected to duplicate the voltages that would otherwise be experienced by the main and start windings of motor 150 if it was operated at its rated voltage and frequency as a conventional PSC motor including a run capacitor in series with the start winding. However, during variable speed operation of motor 150, the drive system of the invention varies the absolute magnitudes of voltages V.sub.P1 and V.sub.P2 to maintain a substantially constant ratio of volts/hertz for each winding while the relative magnitudes of the voltages expressed as a ratio between those voltages, i.e., V.sub.P2 /V.sub.P1, is maintained substantially constant, as more fully described below.

In accordance with the illustrated embodiments of the invention, the phase angle difference between the currents respectively flowing in windings 152 and 154, including the leading or lagging relationship, is controlled by operating the two-phase power source to generate the two-phase voltages V.sub.P1 and V.sub.P2 with the desired phase angle therebetween, rather than as a result of installing a run capacitor in series with the start winding of the conventional PSC motor.

A benefit obtained by such operation of motor 150 is the capability to maintain a selected phase shift between the winding voltages independent of motor rotational speed. This allows the motor to deliver a normally specified range of torque at any speed, including at zero speed corresponding to starting, so long as the air gap flux is held substantially constant. Since the voltages applied to the respective windings of motor 150 have different magnitudes, a different volts/hertz ratio is maintained for each of windings 152 and 154. As more fully described below, the power supply circuit for providing the two-phase power source is preferably configured to maintain a constant volts/hertz ratio for each motor winding for the full range of motor shaft speed.

A two-phase power source for supplying phase voltages V.sub.P1 and V.sub.P2 in accordance with the present invention can be provided in several different forms. For example, the power source can be provided as a two-phase alternator or two, single-phase alternators driven by a common shaft to provide an adjustable phase shift. Output voltage adjustment of the alternator(s) is accomplished by adjustment of the alternator excitation voltage. In the case of two single-phase alternators, the phase angle can be adjusted by angular adjustment of the respective alternator rotors on the common shaft.

The power source can also be provided as a "Scott connected" transformer configured to change three-phase power into two-phase power. Phase angle adjustment of the Scott transformer output is effected by changing winding taps. Frequency adjustment is effected by varying the frequency of the input voltage. However, these embodiments are not preferred for providing the direction reversing feature of the present invention.

It is preferred herein that the power source be provided as a power supply circuit, including an inverter, for electronically generating the two-phase power. FIG. 3 illustrates an induction motor drive system 200 constructed in accordance with an embodiment of the invention. System 200 is preferably constructed to include motor 150. System 200 includes line terminals 202 and 204 for connection to a line conductor and a reference or neutral conductor, respectively, of a single-phase AC power source. System 200 also includes a full wave rectifier circuit 206 including a first pair of diodes 208 and 210 connected in series with the polarity shown, and a second pair of diodes 212 and 214 connected in series with the polarity shown and in parallel with the first pair of diodes. Rectifier circuit 206 has a positive output terminal 220 and a negative output terminal 222.

In the preferred embodiment, a power factor controller 225 is also provided across terminals 220 and 222 which may be used to boost the DC voltage provided by full wave rectifier circuit 206 to a desirable level. For example, the power factor controller 225 could comprise an inductor (not shown), a controlled switch (not shown), and a diode (not shown) to boost the DC voltage to a desired level while providing sinusoidal input current waveforms to the power source. A capacitor 224 connected across the output terminals smooths the output voltage to a constant DC level. A tap node 226 between diodes 208 and 210 is connected to line terminal 202, and a tap node 228 between diodes 212 and 214 is connected to line terminal 204.

Rectifier circuit 206 operates in a well-known manner to rectify the single-phase AC voltage applied to line terminals 202 and 204 to provide an average voltage on its output terminals 220 and 22 which is approximately equal to the RMS single-phase voltage on line terminals 202 and 204 multiplied by 2.sqroot.2/.pi.. Alternatively, output terminals 220 and 222 can be connected directly (not shown) to the filter capacitor 224 to provide a DC voltage approximately equal to the RMS single-phase voltage on line terminals 202 and 204 multiplied by .sqroot.2.

System 200 further includes a three-phase inverter circuit 230 which is connected and operated to provide two-phase power to motor 150, in a manner more fully described below. Inverter circuit 230 includes three parallel connected bridge inverter legs 232, 234, and 236 each formed by two power switching devices connected in series.

Referring to FIG. 4(A), each switch S.sub.1 -S.sub.6 can be provided as a transistor, insulated gate bipolar transistor, F.E.T., G.T.0. device, or similar power switching device 240 connected with an accompanying anti-parallel conduction diode 242.

Most preferably, each switch S.sub.1 -S.sub.6 is provided as a sense FET 245 which provides overcurrent protection as shown in FIG. 4(B). During operation, situations such as motor overload, short circuit, or component failure, for example, can cause excessive currents to flow through switches S.sub.1 -S.sub.6. Sense FET 245 utilizes a portion of the intrinsic resistance of the sense FET in the "on" state to produce a current output signal 251 indicative of the current flowing through sense FET 245. Output 251 is provided to underlap and output drivers (see FIGS. 6(A), 6(B), and 12), and when the current output signal 251 exceeds a predetermined value, the underlap and drivers are disabled, thereby stopping the generation of switching signals by the underlap and output drivers to cause switches S.sub.1 -S.sub.6 to remain in their "off" state. In this way, overcurrent protection can be provided for three-phase inverter circuit 230. Most preferably, a sense FET Model No. IRC840 produced by International Rectifier Corp., of El Segundo, Calif., may be used.

Referring again to FIG. 3, bridge legs 232, 234, and 236 of inverter circuit 230 respectively include tap nodes 244, 246, and 248 for connection to motor 150 through a three-pole double throw switch 250. Tap node 244 is connectable to an end of winding 154, tap node 248 is connectable to an end of winding 152 and tap node 246 is connectable to common winding node 156. Tap nodes 244, 246, and 248 respectively correspond to the A, B, and C phase output nodes of inverter circuit 230 and are so labelled in FIG. 3. As more fully described below, switches S.sub.1 -S.sub.6 are operated to generate pulse width modulated (PWM) voltages at nodes 244, 246, and 248 that are defined with respect to negative output terminal 249 that also serves as a reference terminal R for defining the PWM voltages.

Switch 250 enables operation of motor 150 by connection to either the single-phase source or the output of inverter circuit 230. Switch 250 includes a first group of terminals a-b-c, a second group of terminals d-e-f, and a third group of terminals g-h-i. The first groups of terminals a, b and c are respectively connected to an end of winding 154, common winding node 156, and an end of winding 152. The second group of winding terminals d, e, and f are respectively connected to tap nodes 244, 246, and 248. With respect to the third group of winding terminals, terminals h and i are directly connected to line terminals 204 and 202, respectively, and a run capacitor 260 is connected between terminal g and line terminal 202.

When switch 250 is in a first position connecting terminals a, b, and c to terminals d, e, and f, respectively, motor 150 is operated in a two-phase mode from the output of inverter circuit 230. When switch 250 is in a second position connecting terminals a, b, and c to terminals g, h, and i, respectively, motor 150 is operated in a single-phase mode from the single-phase AC source. During the single-phase mode of operation, run capacitor 260 is connected in series between winding 154 and line terminal 202. Since motor 150 is preferably provided as a conventional PSC motor, capacitor 260 is preferably selected to have the run capacitor characteristics specified by the manufacturer of motor 150, so that motor 150 operates as a conventional PSC motor in the single-phase mode. Direction reversal is not practiced in the single phase mode of operation.

FIGS. 5(A) and 5(B) illustrate phasor diagrams for explaining how three-phase inverter circuit 230 can be connected and controlled to supply two-phase power to motor 150. In FIG. 5(A), the three voltage phasors V.sub.A, V.sub.B, and V.sub.C correspond to the voltages present at tap nodes 244, 246, and 248 of inverter circuit 230. As a result of the connection of inverter circuit 230 to motor 150 as illustrated in FIG. 3, a voltage V.sub.AB, that is the difference between voltage phasors V.sub.A and V.sub.B, is applied across winding 154 and corresponds to voltage V.sub.P2 in FIG. 2. A voltage V.sub.CB, that is the difference between voltage phasors V.sub.C and V.sub.B, is applied across winding 152 and corresponds to voltage V.sub.P1 in FIG. 2.

In accordance with the illustrated embodiment of the present invention, although inverter circuit 230 is a three-phase inverter circuit, it is not operated to produce balanced three-phase power. Thus, as seen in FIG. 5(A), phasor V.sub.C lags phasor V.sub.A by 180.degree. and phasor V.sub.B leads phasor V.sub.A by 105.degree.. Also, phasors V.sub.A, V.sub.B, and V.sub.C have the same magnitude. In FIG. 5(A) voltage V.sub.AB leads voltage V.sub.CB by a phase angle of .THETA.=90.degree.. For direction reversal, voltage V.sub.AB is caused to lag voltage V.sub.CB as discussed above. The leading or lagging relationship between voltages V.sub.AB and V.sub.CB (i.e., whether .THETA. is positive or negative) can be selected by interchanging the magnitude and direction of phasors V.sub.A with V.sub.B while phasor V.sub.C is maintained at a 180.degree. lagging relationship with phasor V.sub.A. In other words, phasors V.sub.A and V.sub.B can be "swapped" to desirably achieve either a leading or lagging relationship between voltages V.sub.AB and V.sub.CB.

As described more fully below, in the embodiments illustrated in FIGS. 6(A) and 6(B), phasor V.sub.C is derived from the inversion of phasor V.sub.A and is therefore always 180.degree. phase shifted from phasor V.sub.A. As shown in FIG. 5(B), phasors V.sub.A and V.sub.B have been swapped. Phasor V.sub.C retains its 180.degree. phase shift with respect to phasor V.sub.A. In this case, voltage V.sub.AB now lags voltage V.sub.CB by a phase angle .THETA.=90.degree..

In accordance with the invention and in a manner more fully described below, the operation of switches S.sub.1 -S.sub.6 is controlled to produce voltages V.sub.AB and V.sub.CB that constitute the two-phase output of inverter circuit 230 and that are separated by the desired phase angle .THETA. that corresponds to a desired rotation direction and an optimum performance of motor 150.

As described above and in accordance with the illustrated embodiments of the invention, the magnitude of volta V.sub.AB, which is applied across start winding 154, is greater than that of voltage V.sub.CB, which is applied across main winding 152. Since phasors V.sub.A, V.sub.B, and V.sub.C are defined to have the same magnitude, the difference in magnitude between voltages V.sub.AB and V.sub.CB is controlled by the selection of the phase angles between the phasors V.sub.A and V.sub.B. As will now be apparent to those skilled in the art, by varying the phase angle between phasor V.sub.B and V.sub.A while holding the magnitude of V.sub.B and the relative phase angles of phasors V.sub.A and V.sub.C constant, the ratio of voltages V.sub.AB /V.sub.CB can be varied while the phase angle difference between voltages V.sub.AB and V.sub.CB remains constant at 90.degree.. Further, the variation of both the ratio of voltages V.sub.AB /V.sub.CB and the phase angle therebetween can be accomplished by varying the phase and magnitude of any two of the three phasors V.sub.A, V.sub.B, and V.sub.C. The operating frequency of inverter 230, and hence the speed of motor 150, is varied by varying the respective magnitudes and frequencies of all three phasors V.sub.A, V.sub.B, and V.sub.C in proportion to a desired operating frequency while holding the phase relationships between the respective phasors constant, in order to maintain a constant volts/hertz ratio for each of windings 152 and 154 and the desired phase angle .THETA. between the voltages applied to the respective windings.

FIGS. 6(A) and 6(B) respectively show block diagrams of first and second embodiments of a control circuit 300 for generating control signals for controlling the operation of switches S.sub.1 -S.sub.6 to produce voltages corresponding to phasors V.sub.A, V.sub.B, and V.sub.C as illustrated in FIG. 5(A), and with phase angle .vertline..THETA..vertline.=90.degree. between the voltages applied to the motor windings. In FIGS. 6(A) and 6(B), like elements (i.e., those with like reference numerals) generally perform similar functions. Accordingly, the following description will apply to the control circuits shown in FIGS. 6(A) and 6(B) with respect to like elements. FIG. 7 illustrates the waveforms of various signals occurring during operation of control circuit 300.

Referring now to FIGS. 6(A) and 6(B), circuit 300 includes a speed ramp rate circuit 302 connected to receive an externally generated speed control signal that represents a desired operating speed of motor 150. The speed control signal may be derived by sensing some parameter or characteristic of the system, in which motor 150 is incorporated, in order to automatically control the motor speed in response to that sensed information. Alternatively, the speed control signal may be varied by means of an appropriate manually adjustable control, such as a potentiometer. Circuit 302 converts the speed control signal into a DC voltage, in a normalized range of 0 to 5 volts, on its output 304. The specific construction of circuit 302 depends on the nature of the speed control signal. However, for example and without limitation, if the speed control signal is provided as a -0 to 5 VDC signal, then circuit 302 can be provided as an R-C network configured as a low pass filter.

The DC voltage output of circuit 302 is applied to an input of a voltage controlled oscillator (VCO) 306 that provides on its output 308 a square wave having a frequency proportional to the DC voltage applied to the input of VCO 306. For example, VCO 306 provides a square wave output that has a frequency of 0 Hz, 66.44 KHz and 122.88 KHz respectively corresponding to DC voltage inputs of 0, 2.5, and 5 volts. VCO 306 can be provided as a model no. AD654JN manufactured by Analog Devices, Inc. of Norwood, Mass.

Output 308 of VCO 306 is connected to an inverted clock input of a twelve bit binary counter 310. Only eleven counter output terminals, Q.sub.1 to Q.sub.11 of counter 310 are used. Counter 310 can be provided as a model no. MC74HC4040N manufactured by Motorola, Inc. of Phoenix, Ariz.

Circuit 300 also includes a 2K-by-8 bit EEPROM 320 connected to receive on its address inputs AD.sub.1 to AD.sub.11 the output of counter 310 provided on output terminals Q.sub.1 to Q.sub.11 EEPROM 320 has stored therein a look-up table containing digital data values that define sinusoidal waveforms W.sub.DA and W.sub.DB respectively corresponding to phasors V.sub.A and V.sub.B described above with respect to FIGS. 5(A) and 5(B). In particular, EEPROM 320 stores digital data for waveforms W.sub.DA and W.sub.DB in digitally weighted form broken into 1024 incremental pieces per sinusoidal cycle. The incremental data for digital waveforms W.sub.DA and W.sub.DB are alternately stored at successive address locations in EEPROM 320 so that successive address values generated by counter 310 cause EEPROM 320 to produce on its data outputs D.sub.1 to D.sub.11 digital data respectively describing incremental pieces representative of waveforms A and B. EEPROM 320 can be provided as a model no. NMC27C16Q manufactured by National Semiconductor, Inc. of Santa Clara, Calif.

Table 1, shown below, illustrates exemplary values of the incremental data stored in EEPROM 320 corresponding to waveforms A and B.

The left-hand column of Table 1 lists each successive seventeenth storage address of EEPROM 320 in hexidecimal form. The remaining columns of Table 1 list the waveform data values which are also in hexidecimal form. In particular, the sixteen data values to the right of an address correspond to the data value stored at that address and the fifteen data values respectively stored at the next fifteen successive addresses. For example, the data value BD is stored at address 00A0, while the data values FO, BD, FO, etc. are respectively stored at addresses 00A1, 00A2, 00A3, etc. Further, since data values for waveforms W.sub.DA and W.sub.DB are alternately stored at successive locations, the data values for waveform W.sub.DA are stored at addresses 00A0, 00A2, etc., while that data values for waveform W.sub.DB are stored at addresses 00A1, 00A3, etc.

Referring again to FIGS. 6(A) and 6(B), the data outputs D.sub.1 to D.sub.8 of EEPROM 320 are applied to a digital input of a multiplying dual digital-to-analog converter (DAC) circuit 330. DAC circuit 330 includes a DAC select input 332. DAC circuit 330 converts to analog form the digital data it receives from EEPROM 320 and updates the analog result on either of its outputs 334 or 336 depending on the binary value applied to its DAC select input 332.

Referring now particularly to FIG. 6(A), a logic inverting circuit 331 is provided between output Q.sub.1 of counter 320 and DAC select input 332 of DAC circuit 330 to provide selective direction reversal of the motor 150. Logic inverting circuit 331 may operate in a first direction mode or a second direction mode. In the first direction mode, the digital value Q.sub.1 output from counter 310 is inverted (i.e., converted from logical low to high, or high to low) and applied to the DAC select input 332. In the second direction mode, the digital value Q.sub.1 is directly passed through to the DAC select input 332. Logic inverting circuit 331 receives a direction control signal from an outside source to select its mode of operation. The outside source may be the user of the motor 150, or a controller directing the operation of a system in which the motor operates, for example. The direction control signal may be used to control the direction of the motor 150. For example, logic inverting circuit may be an operational amplifier with a gain of -1 constructed by an operational amplifier (not shown) with its output fed back to the inverting input terminal.

Since counter output Q.sub.1 is the least significant bit of the output of counter 310, DAC circuit 330 alternatively updates each analog output, corresponding to each successive digital data input, alternately on its outputs 334 and 336. In the illustrated embodiment in FIG. 6(A) with the logic inverting circuit 331 in the second direction mode, sinusoidal analog waveforms A and B, representative Of voltage phasors V.sub.A and V.sub.B, are respectively provided on outputs 336 and 334. If the inverting circuit 331 is in the first direction mode, sinusoidal analog waveforms A and B, representative of voltage phasors V.sub.A and V.sub.B are instead respectively provided on outputs 334 and 336.

FIG. 7 illustrates sinusoidal analog waveforms A and B provided on outputs 336 and 334 of DAC circuit 330 when logic inverting circuit 331 is in the second direction mode. In accordance with the illustrated embodiment and the data listed in Table 1, waveforms A and B have the same magnitude and a phase angle therebetween of 105.degree., as noted above with respect to phasors V.sub.A and V.sub.B (FIG. 5(A)). In accordance with the illustrated embodiment, if logic inverting circuit 331 is in the first direction mode, waveforms A and B have a phase angle therebetween of -105.degree. (not shown).

DAC circuit 330 is further connected to receive a voltage scaling signal on a V.sub.REF input 338. In accordance with the magnitude of voltage scaling signal, DAC circuit 330, by operation of its multiplying function, scales the magnitude of its analog output in proportion to the voltage scaling signal. The voltage scaling signal is generated by a level converter circuit 340 that is connected to receive the DC voltage provided on output 304 of circuit 302. Circuit 340 is configured to generate the scaling signal as a DC voltage in proportion to the desired operating speed of motor 150 so that a predetermined constant volts/hertz ratio can be maintained for the voltage applied to each winding of motor 150. As a result, for example, the analog sinusoidal waveform voltage output by DAC circuit 330 on either output 334 or 336 has a peak-to-peak magnitude approximately ranging from 0 volts to 2.5 volts respectively corresponding to output voltages on output 304 of circuit 302 ranging from 0 volts to 5 volts. Circuit 340 can be provided as a resistive divider circuit. DAC circuit 330 can be provided as a model no. AD7528 CMOS dual 8-bit buffered multiplying DAC manufactured by Analog Devices, Inc. of Norwood, Mass. That model DAC has two V.sub.REF inputs which are conductively tied together for practicing the illustrated embodiment.

Referring to FIG. 6(A), the analog waveforms B and A provided on outputs 334 and 336 of DAC circuit 330 are applied through conventional analog buffer circuits 350 and 352 to first inputs of comparator circuits 354 and 356. In accordance with this illustrated embodiment of the invention, voltage phasor V.sub.C has the same magnitude as phasors V.sub.A and V.sub.B but lags by 180.degree. phasor V.sub.A. Therefore, the output of buffer circuit 352 is inverted by an analog inverter circuit 358 to create an analog waveform C representative of phasor V.sub.C which is applied to a first input of a comparator circuit 360. Waveform C is illustrated in FIG. 7.

Referring now to FIG. 6(B), a second embodiment of a circuit for controlling the operation of the inverter circuit of FIG. 3 is disclosed therein. In FIG. 6(B), output Q.sub.1 from counter 310 is directly connected to DAC select input 332 of DAC 330. Furthermore, outputs 334 and 336 of DAC 330 shown in FIG. 6(B) are applied to a signal switching circuit 337 to provide selective direction reversal of motor 150 in this embodiment. Waveforms B and A are output from the signal switching circuit 337 to conventional analog buffer circuits 350 and 352, respectively.

Signal switching circuit 337 may be operated in a first direction mode or a second direction mode. A direction control signal applied to the signal switching circuit 337 may be used to select between such modes and provide the selected motor direction. In the second direction mode, waveform B provided on output 334 of DAC 330 is provided by signal switching circuit 337 directly to buffer circuit 350 and waveform A provided on output 336 of DAC 330 is provided by signal switching circuit 337 directly to buffer 352. In the first direction mode, waveform B provided on output 334 of DAC 330 is provided by signal switching circuit 337 to buffer 352 and waveform A provided on output 336 of DAC 330 is provided by signal switching circuit 337 to buffer 350. As a result, in the first direction mode, waveforms A and B are swapped with respect to buffers 350 and 352 and the circuit elements connected thereto.

Circuit 300 additionally includes a triangular wave generator 370 that generates a triangular waveform T on its output 372. The generated triangular waveform T can, for example, have a peak-to-peak magnitude that varies from 1.25 to 3.75 volts. The frequency of the triangular waveform T determines the frequency at which switching signals are generated for application to switches S.sub.1 to S.sub.6 of inverter circuit 200. Triangular wave generator 370 can be provided as a circuit including an operational amplifier and a comparator, having a construction well known in the art. Presently, a switching frequency of approximately 7.5 KHz is preferred. In view of the above, those skilled in the art will now recognize that generator 370 may be operated at other frequencies, depending upon particular applications, in order to achieve desirable results.

The triangular waveform T generated on output 372 of generator 370 is applied to a second input of each of comparators 354, 356, and 360. FIG. 7 illustrates triangular waveform T superposed on each of waveforms A, B, and C, as received by comparators 356, 354, and 360, respectively. For illustrating purposes only, waveform T in FIG. 7 has a frequency of approximately 1000 Hz. Each of comparators 354, 356, and 360 generates on its output either a high logic level voltage, e.g., 5 volts, or a low logic level voltage, e.g., 0 volts, according to whether the magnitude of the applied analog waveform voltage A, B, or C is respectively greater or less than the magnitude of triangular waveform T. As a result, the output of each comparator constitutes a pulse width modulated waveform.

The pulse width modulated waveforms output by comparators 354, 356, and 360 are respectively applied to underlap and output driver circuits 380, 382, and 384. Circuit 380 is configured to generate a first switch driving signal B+that has substantially the same form as the pulse width modulated waveform applied thereto and a voltage level sufficient to drive switching device S.sub.3 (FIG. 3). Circuit 380 also includes a logic inverter circuit for generating a switch driving signal B- that is the logical complement of waveform B+ and having a voltage level sufficient to drive switching device S.sub.4 (FIG. 3). Circuit 380 further includes underlap circuitry to insure that signals B+ and B- never cause the respective switching devices to which they are applied to turn on at the same time, thereby providing for the time delay between the logic level turn-off command and the actual interruption of current by the switching device.

Circuits 382 and 384 are configured in the same way as circuit 380 so that circuit 382 provides switch driving signals A+ and A- for respectively driving switches S.sub.1 and S.sub.2, and circuit 384 provides switch driving signals C+ and C- for respectively driving switches S.sub.5 and S.sub.6. FIG. 7 illustrates switches driving signals A+, B+, and C+.

In the operation of inverter circuit 230 and control circuit 300, circuit 300 generates switch driving signals A+, A-, B+, B-, C+, and C- which are applied to respectively drive switches S.sub.1 -S.sub.6, such that a switch is driven closed when the driving signal applied thereto assumes a positive logic value. As a result of the pulse width modulated (PWM) closures of switches S.sub.1 -S.sub.6, PWM voltages are applied to windings 152 and 154 of motor 150. For example, the operation of switches S.sub.1 and S.sub.2 results in generation at node 244 of a PWM voltage V.sub.AR that is applied to one end of winding 154. Similarly, the operation of switches S.sub.3 -S.sub.6 results in generation of PWM voltages V.sub.BR and V.sub.CR that are respectively applied to common winding node 156 and an end of winding 152. PWM voltages V.sub.AR, V.sub.BR, and V.sub.CR are illustrated in FIG. 7 and respectively correspond to previously described phasors V.sub.A, V.sub.B, and V.sub.C. It is noted that the magnitudes of PWM voltages V.sub.AR, V.sub.BR, and V.sub.CR are a function of the magnitude of DC voltage provided across filter capacitor 224.

Application of PWM voltages V.sub.AR, V.sub.BR, and V.sub.CR to motor 150 results in application of voltages V.sub.AB and V.sub.CB as PWM voltages, respectively across windings 154 and 152. FIG. 8(A) illustrates PWM voltages V.sub.AR, V.sub.BR, and V.sub.AB, while FIG. 8(B) illustrates PWM voltages V.sub.BR, V.sub.CR, and V.sub.CB. In accordance with the well known operation of pulse width modulated inverter circuits, the pulse width modulated voltage waveforms applied to motor 150 closely simulate the sinusoidal waveforms they correspond to.

In the disclosed embodiments, the frequency of the triangular wave generator 370, which corresponds to the frequency at which the PWM voltages are applied to motor 150, is 7.5 KHz, which provides satisfactory operation of the motor 150 because the currents flowing through the motor 150 more closely approximate a sinusoidal signal. While this "switching frequency" could be increased, the components of the PWM inverter, such as switches S.sub.1 -S.sub.6, may have an upper switching frequency limit at which their performance begins to degrade. Accordingly, an optimum switching frequency should be utilized to provide a good sinusoidal approximation for the voltages applied to the motor, without degrading the performance of the components of the circuit 200.

Additionally, the switching frequency could be adjusted within an acceptable range to reduce the previously discussed noise and vibration problems. Such noise and vibration may be caused by the harmonic frequency components supplied to the motor, including the switching frequency component. For example, the switching frequency component may cause annoying vibration in the laminated motor windings. The switching frequency could be adjusted to reduce this noise and vibration by selecting a frequency which produces minimal acoustical amplitude.

A low pass filter (not shown) may be effectively coupled with each motor winding, thereby providing to the winding a sinusoidal voltage in which the high frequency switching components are minimized to reduce the dV/dt stress on the motor. For example, a capacitor (not shown) could be shunted across each motor winding and an inductor (not shown) could be put in series with each motor winding to form the low pass filter. An example of a low pass filter used in a single phase power supply for a motor is provided in U.S. Pat. Nos. 4,706,180 and 4,651,079, both issued to F. E. Wills on Nov. 10, 1987, and Mar. 17, 1987, respectively, both of which are incorporated by reference herein.

The operating speed of motor 150 can be varied in response to the speed control signal. During such variation, a constant volts/hertz ratio is maintained for the voltage applied to each winding 152 and 154 of motor 150 by operation of the multiplying function of DAC circuit 330 in accordance with the voltage scaling signal provided by level converter circuit 340. Further, while the magnitudes of the voltages V.sub.AB and V.sub.CB applied to motor 150 are varied during variable speed operation, the ratio between those voltages, i.e., their relative magnitudes, is maintained constant since the relative phase angles between voltage phasors V.sub.A, V.sub.B, and V.sub.C are fixed by the data stored in EEPROM 320.

During starting of motor 150, assuming operation at rated speed is desired, the speed control signal can initially require such full speed operation, which would result in an initial application of full voltage across the motor windings. This in turn would result in large motor inrush currents. It would therefore be necessary to provide switches S.sub.1 -S.sub.6 with a current rating sufficient to accommodate such inrush currents. Alternatively, still assuming that operation at rated speed is desired, and in accordance with known inverter operating practices, the speed control signal could initially be generated to require a low speed corresponding to the rated slip of the motor, i.e., approximately 2-5% of rated speed, so that no more than rated current flows initially. The motor could then be accelerated to rated speed by steadily increasing the magnitude of the speed control signal.

FIGS. 9(A) and 9(B) illustrate as a control circuit 400 alternate embodiments of control circuit 300 illustrated in FIGS. 6(A) and 6(B). In FIGS. 9(A) and 9(B), like elements generally perform similar functions and the following description applies to both FIGS. 9(A) and 9(B) with respect to those like elements. Control circuit 400 has the same construction as that of circuit 300 except for the portion shown in FIGS. 9(A) and 9(B). A difference between circuits 300 and 400 is that the latter circuit stores in EEPROM 320 incremental digital data for a waveform W.sub.DC to enable independent generation of analog waveform C instead of its generation by inversion of analog waveform A.

A further difference is that the embodiments of FIGS. 9(A) and 9(B) generate voltage phasors V.sub.A, V.sub.B, and V.sub.C in phase quadrature. In other words, and contrary to their apparent location in FIGS. 5(A) and 5(B), phasors V.sub.A, V.sub.B, and V.sub.C are displaced from each other by integer multiples of 90.degree. as shown in FIG. 5(C). As explained below, the angular displacement between each of these phasors may vary, while remaining in phase quadrature, depending upon the direction of motor rotation selected.

Referring to FIGS. 9(A) and 9(B), control circuit 400 includes a multiplying quad digital-to-analog converter (DAC) circuit 402 that is converted to receive the eleven bit digital data output from EEPROM 320. DAC circuit 402 includes a two bit DAC select input consisting of two select inputs 404 and 406. DAC circuit 402 converts to analog form the digital data it receives from EEPROM 320 and provides the analog output on one of its three outputs 408, 410, or 412 depending on the two bit binary value applied to its DAC select inputs 404 and 406. DAC circuit 402 is also connected to receive the voltage scaling signal on a V.sub.REF input 420 to perform a multiplying function in the same manner and for the same purpose as previously described for DAC circuit 330. DAC circuit 402 can be provided as a model no. AD7225 quad 8-bit DAC manufactured by Analog Devices, Inc. of Norwood, Mass. That model DAC has four V.sub.REF inputs which would be conductively tied together for practicing the illustrated embodiment.

Circuit 400 also includes a timing circuit 430 for generating a sequence of two bit signals. Timing circuit 430 includes an AND gate 432 which is connected to receive outputs Q.sub.1 -Q.sub.11 of counter 310, with outputs Q.sub.1 and Q.sub.2 being inverted prior to application to AND gate 432, by circuitry not shown. Circuit 430 also includes flip-flop circuits 434 and 436 each having a reset input R connected to receive the sum output signal provided by AND gate 432. The output signal of AND gate 432 is also applied to a reset input R of counter 310. Each of flip-flops 434 and 436 is also connected to receive on an inverted clock input C the signal provided on output 308 of VCO 306. An inverted data output Q of each of flip-flops 434 and 436 is applied to an AND gate 440 the output of which is applied to a data input D of flip-flop 434. A data output Q of flip-flop 434 is connected to a data input D of flip-flop 436 and provided as a first output 435 of timing circuit 430. A data output Q of flip-flop 436 is provided as a second output 437 of timing circuit 430.

In the operation of control circuit 400, timing circuit 430 generates as output the repeating sequence of two bit logic signals 00, 10, 01, 00, 10, 01, . . . as first output signal 435 corresponding to the first bit of the pair and second output signal 437 corresponding to the second bit of the pair. Referring now particularly to FIG. 9(A), therein is shown a logic inversion circuit 407 which receives the first and second output signals 435 and 437 from the timing circuit 430. Logic inversion circuit 407 provides two output signals to DAC select inputs 404 and 406. Logic inversion circuit 407 may be operated in a first direction mode or a second direction mode, as selected by the direction control signal input thereto to control the motor direction in this embodiment. In the first direction mode, the first output signal 435 from the timing circuit 430 (provided by the Q output from flip-flop 434) is provided to DAC select input 404, and the second output signal 437 from the timing circuit 430 (provided by the Q output from flip-flop 436) is provided to DAC select input 406.

In the second direction mode, the first and second output signals from the timing circuit are logically manipulated to cause DAC 402 to "swap" the signals provided on outputs 408 and 412. In other words, the second direction mode causes the signal that would have been provided on output 410 in the second direction mode to be provided on output 412, and causes the signal that would have been provided an output 412 in the second direction mode to be provided on output 408.

For example, assume DAC 402 provides a signal on output 408 upon the input of logic signal "10" to the DAC select inputs 404 and 406, and DAC 402 provides a signal on output 412 upon the input of logic signal "01" to the DAC select inputs 404 and 406. Then, in the first direction mode, logic inversion circuit 407 would simply pass through logic signals "10" and "01" from timing circuit 430 to DAC select inputs 404 and 406. However, in the second direction mode, logic inversion circuit 407 would output logic signal "01" to DAC select inputs 404 and 406 upon receiving logic signal "10" from the timing circuit 430. Similarly, logic inversion circuit 407 would output logic signal "10" to DAC select inputs 404 and 406 upon receiving logic signal "01" from the timing circuit 430. With regard to the remaining logic signal "00", which would select output 410, the logic circuit 407 would pass through this signal in both the first direction mode and the second direction mode. Logic inversion circuit 407 may comprise a digital data selector to output the desired logic signals upon receiving the first and second output signals 435 and 437 and the direction control signal.

FIG. 9(B) provides a fourth embodiment of a circuit according to the present invention. In FIG. 9(B) first and second outputs 435 and 437 of timing circuit 430 are provided to DAC select inputs 404 and 406, respectively. Further, a switching circuit 413 is provided which receives outputs 408, 410, and 412 from DAC 402. Switching circuit 413 also receives a direction control signal to select the motor direction for this embodiment. Switching circuit 413 may operate in a first direction mode or a second direction mode depending upon the direction control signal input thereto. In the first direction mode, output 410 is provided to buffer 416, output 408 is provided to buffer 414, and output 412 is provided to buffer 418. In the second direction mode, output 410 is provided to buffer 416, output 408 is provided to buffer 418, and output 412 is provided to buffer 414, thereby swapping outputs 408 and 412.

The embodiments shown in FIGS. 9(A) and 9(B) preferably operate in accordance with the phasors and voltages shown in FIG. 5(C). Referring now to FIG. 5(C), assume phasors V.sub.A, V.sub.B, and V.sub.C shown therein represent the voltages generated by circuit 400 in the first direction mode. Further, resultant volta V.sub.AB is shown leading voltage V.sub.CB by phase angle .THETA. in FIG. 5(C). Upon direction reversal, phasors V.sub.A and V.sub.C will be swapped with the result that the voltage V.sub.AB will lag (not shown V.sub.CB by phase angle .THETA. in the second direction mode. As will now be evident to one of ordinary skill, phase angle .THETA. may be adjusted by varying the magnitude of phasor V.sub.B, for example.

FIG. 10 illustrates plots of the signal generated by VCO 306 and the first and second output signals 435 and 437 generated by timing circuit 430 for application to the logic inversion circuit 407, relative to a common time axis. In FIG. 10, the signal generated by VCO 306 has a frequency of 122.76 KHz. As a result, DAC circuit 402 is controlled to sequentially provide on its outputs 408, 410, and 412, in analog form, the incremental data for analog waveforms A, B and C. Since in the illustrated embodiment EEPROM 320 is provided with an approximate 2K byte storage capacity, the digital data for waveforms W.sub.DA, W.sub.DB and W.sub.DC is stored in digitally weighted form broken into 682 incremental pieces per sinusoidal cycle. The sum signal applied to the reset input R of counter 310 causes the counter to reset its output after sequencing through a range of 2046 addresses of EEPROM 320.

FIG. 11 illustrates an induction motor drive system 500 constructed in accordance with another embodiment of the invention. System 500 is preferably constructed to include previously described motor 150. System 500 includes line terminals 502 and 504 for connection to a line conductor and a reference or neutral conductor, respectively, of a single-phase AC power source. System 500 also includes a full wave rectifying voltage doubler circuit 510 consisting of capacitors 512 and 514 and diodes 516 and 518. Capacitors 512 and 514 are connected in series with a neutral terminal node 520 therebetween connected to both line terminal 504 and motor common winding node 156. Diode 516 is connected, with the polarity shown in FIG. 11, between line terminal 502 and a positive output terminal node 522 of circuit 510. Similarly, diode 518 is connected with the polarity shown between line terminal 502 and a negative output terminal node 524 of circuit 510.

With respect to the operation of rectifying circuit 510, during each positive half-cycle of the applied single-phase AC power source, current flows through diode 516 to charge capacitor 512 when the voltage on line terminal 502 is positive relative to the voltage on line terminal 504. During the negative half cycles of the voltage on line terminal 502, current flows through diode 518 to charge capacitor 514. The circuit elements of circuit 510 are preferably selected so that the DC voltages across capacitors 512 and 514 are equal in magnitude, although opposite in polarity. As a result, capacitors 512 and 514 constitute a balanced DC voltage source having positive output terminal node 522, negative output terminal node 524 and neutral terminal node 520. The magnitude of the volt either output terminal node 522 or 524 is approximately equal to the RMS single phase voltage on line terminal 502 multiplied by .sqroot.2.

System 500 further includes an inverter circuit 530, consisting of half-bridge inverter legs 532 and 534, for generating a two-phase voltage. Inverter leg 532 consists of power switching devices 536 and 538 connected in series across output nodes 522 and 524 of circuit 510. Inverter leg 534 consists of power switching devices 540 and 542 connected in series across output nodes 522 and 524 and thus in parallel with inverter leg 532. Each of devices 536, 538, 540, and 542 are of the same type described above with respect to devices S.sub.1 -S.sub.6 shown in FIGS. 3, 4(A), and 4(B), and each includes an anti-parallel conduction diode as shown in FIG. 11.

System 500 additionally includes a double-pole, double throw switch or relay 550 to enable operation of motor 150 by connection to either the single-phase line source or the output of inverter circuit 530. Switch 550 includes a first pair of terminals a-b, a second pair of terminals c-d and a third pair of terminals e-f. The first pair of terminals a and b of switch 550 are respectively connected to center-tap terminal nodes 552 and 554 of inverter legs 532 and 534. The third pair of terminals e and f of switch 550 are respectively connected to windings 152 and 154 of motor 150. The second pair of terminals c-d are connected to line terminal 502, terminal c being connected directly and a run capacitor 560 being connected between terminal d and line terminal 502.

When switch 550 is in a first position connecting terminals a and b to terminals e and f, respectively, motor 150 is operated in a two-phase mode from the output of inverter circuit 530. When switch 550 is in a second position connecting terminals c and d to terminal e and f, respectively, motor 150 is operated in a single-phase mode from the single phase AC source. During the single-phase mode of operation, run capacitor 560 is connected in series between winding 154 and line terminal 502. Since motor 150 is preferably provided as a conventional PSC motor, capacitor 560 is preferably selected to have the run capacitor characteristics specified by the manufacturer of motor 150, so that motor 150 operates as a conventional PSC motor in the single-phase mode.

In accordance with the present invention, inverter circuit 530 is operated to provide a two-phase pulse width modulated output for application to motor 150 during the two-phase mode of operation. Switching devices 536, 538, 540, and 542 are operated by switch driving signals A+, A-, B+, and B-. Control circuit 300 illustrated in FIGS. 6(A) and 6(B) can be used to generate the necessary driving signals to control the operation of inverter circuit 530. However, it would be necessary to store in EEPROM 320 information corresponding to analog waveforms A and B that have between them a desired phase angle, e.g., 90.degree., for operation of motor 150 within system 500. That is, since the voltages generated at tap nodes 552 and 554 are directly respectively applied to windings 152 and 154, the phase angle between those voltages must be the desired phase angle for operation of motor 150. It would also be necessary to modify circuit 300 to provide analog waveforms A and B with the different magnitudes which correspond to conventional operation of motor 150, as described above. One way to achieve this is by not tying together the V.sub.REF inputs of the above noted model no. AD7528 DAC, applying the voltage scaling signal output by level converter 340 directly to the V.sub.REF input corresponding to waveform B and also applying the voltage scaling signal output of level converter 340 through an attenuator to the other V.sub.REF input. The attenuator can be provided as, for example, a resistance divider circuit or an operational amplifier circuit with a gain of less than 1.0. If the attenuation is performed to provide a ratio between the voltages applied to the respective V.sub.REF inputs that corresponds to the desired ratio between the voltages applied to the motor windings, then DAC 330 will generate waveforms having magnitudes with the same ratio by operation of its multiplying function.

Further, it is noted that if control circuit 300 is used to control the operation of inverter circuit 530, the portions of circuit 300 provided for generating drive signals C+ and C- are not required.

FIG. 12 illustrates a preferred control circuit 600 for controlling the operation of inverter circuit 530. Circuit 600 includes a sinusoidal wave generator 602 for generating a sine wave voltage signal on an output 604 and a cosine wave voltage signal on an output 606 Generator 602 includes a speed control input 608 for receiving a motor speed control signal, such as described with respect to circuit 300, for varying the frequency of the generated sine and cosine waves in accordance with a desired motor speed. Generator 602 can be provided as a model no. 4423 signal generator manufactured by Burr-Brown, Inc. of Tucson, Ariz.

Outputs 604 and 606 from the wave generator 602 are provided to switching circuit 607 which receives a direction control signal to select the motor direction for this embodiment. Switching circuit 607 provides a first output to high pass filter 610 and a second output to high pass filter 612. Switching circuit 607 may operate in a first direction mode on a second direction mode, depending upon the direction control signal input thereto. In the second direction mode, switching circuit 607 connects output 604 from the wave generator 602 to high pass filter 610 and output 606 to high pass filter 612. In the first direction mode, switching circuit 607 connects output 604 to high pass filter 612 and output 606 to high pass filter 610.

In order to maintain a constant volts/hertz ratio for the voltage generated by inverter circuit 530 as the motor speed is varied, the voltage signals produced by generator 602 on outputs 604 and 606 are passed through switching circuit 607 to high pass filters 610 and 612. High pass filters 610 and 612 comprise series connected capacitors 614 and 616, and shunt connected resistors 618 and 620. Each high pass filter 610 and 612 has a 6 dB/octave roll-off characteristic and a cut-off frequency above the upper limit of the operating frequency range for motor 150. As a result, the sine and cosine wave voltage signals produced by generator 602 are respectively attenuated by high pass filters 610 and 612 in an amount inversely proportional to the frequency of the sine/cosine wave signals. The 6 dB/octave roll-off characteristic provides a 1:1 inverse relationship between changes in frequency and voltage. In this way, the amplitude of the sine and cosine wave voltage signals, at the outputs of high pass filters 610 and 612, respectively, vary with and are directly proportional to, frequency changes in generator 602, thereby achieving a constant volts/hertz ratio for each filter output. Exemplary characteristics of capacitors 614 and 616 and resistors 618 and 620, corresponding to the above-noted high pass filter characteristics, are 0.1 .mu.F for each of capacitors 614 and 616, and 10K.OMEGA. for each of resistors 618 and 620.

Still referring to FIG. 12, resistor 618 is preferably provided as a potentiometer with an adjusting arm 622. Arm 622, and hence the output of high pass filter 610, is connected to a first input of a comparator circuit 630. The output of high pass filter 612 is connected to a first input of a comparator circuit 632. Circuit 600 further includes a triangular wave generator 640 connected to apply its triangular wave signal output to respective second inputs of both comparators 630 and 632. Generator 640 can be the same as generator 370 described above with respect to FIGS. 6(A) and 6(B). Each comparator 630 and 632 generates on its output either a high or a low logic level voltage according to whether the magnitude of the applied sinusoidal or cosinusoidal signal is greater or less than the magnitude of the applied triangular wave. Thus, the comparison performed by each comparator 630 and 632 and the resulting output is substantially the same as described above with respect to comparators 354, 356, and 360 and illustrated in FIG. 7, i.e., each comparator generates a pulse width modulated control signal. The switching frequency of the triangular wave generator 640 may preferably be adjusted as discussed above with regard to triangular wave generator 370 in the embodiments shown in FIGS. 6(A), 6(B), 9(A), and 9(B).

The outputs of comparators 630 and 632 are respectively applied to underlap and output driver circuits 650 and 652 each of which having substantially the same construction and function as circuits 380, 382, and 384 described above with respect to FIGS. 6(A) and 6(B). As a result, circuit 650 provides switch driving signals A+ and A- for respectively driving switches 536 and 538 (FIG. 11). Similarly, circuit 652 provides switch driving signals B+ and B- for respectively driving switches 540 and 542 (FIG. 11).

Provision of resistor 618 as a potentiometer allows adjustment of a setpoint of filter 610 to enable a reduction of the magnitude of the sine wave voltage signal, generated by generator 602, relative to the cosine wave voltage signal. This in turn results in generation of signals, by comparator 630 and circuit 640, that cause inverter circuit 530 to generate a pulse width modulated voltage, for application to main winding 152, that has a smaller magnitude than the voltage generated for application to winding 154. As a result, the above described feature of the illustrated embodiments whereby V.sub.P2 >V.sub.P1 (FIG. 2), is implemented.

In the operation of motor drive system 500, the system is operable in either the two-phase or single-phase mode according to the position of switch 550, as described above. During operation in the two-phase mode, control circuit 600 generates switch driving signals A+, A-, B+ and B- which are applied to respectively drive switches 536, 538, 540, and 542 such that a switch is driven closed when the driving signal applied thereto assumes a positive logic value. As a result of the pulse width modulated closures of the switches, a two-phase pulse width modulated voltage is generated by inverter circuit 530 and applied to windings 152 and 154 of motor 150. As seen in FIG. 11, the voltage generated by each inverter leg 532 and 534 is defined with respect to neutral terminal node 520. Further, in accordance with the manner in which the switch driving signals are generated, the phase angle difference between the voltages applied to the windings of motor 150 is 90.degree.. Additionally, during operation in the two-phase mode, drive system 500 enables variation of the speed of motor 150, in accordance with the speed control signal, while maintaining a constant volts/hertz ratio for each of the voltages generated for application across windings 152 and 154 of motor 150. Considerations for starting motor 150 in the two-phase mode are the same described with respect to inverter circuit 200. Direction reversal can be conveniently achieved by applying an appropriate direction control signal to switching circuit 607, while maintaining optimum efficiency and performance of motor 150.

In the single-phase mode, motor 150 can be operated directly from the single-phase AC source as a conventional PSC motor. As a result, motor 150 remains available for operation even when inverter circuit 530 is not available to drive the motor.

While drive system 500 including control circuit 600 has been described as providing a 90.degree. phase angle difference during operation in the two-phase mode, the invention is not so limited. Circuit 600 can be modified to substitute a sinusoidal wave generator for generator 602 that generates two waves that have any desired phase angle difference between them. As described above, the phase angle between the sinusoidal waves generated by generator 602, or a generator substituted therefor, is the phase angle between the voltages applied to the windings of motor 150.

While the relative magnitudes of the voltages generated by inverter circuit 230 (FIG. 3) are controlled by the phase angle between phasors V.sub.A and V.sub.B, the invention is not so limited. The relative magnitudes of the voltages can also be controlled by storing in EEPROM 320 data for waveforms having different magnitudes that result in the desired relative magnitudes between the voltages generated for application to motor 150. This technique can also be used in the case where control circuit 300 is adapted to generate the switch driving signals for controlling the switching devices of inverter circuit 530 (FIG. 11).

It is additionally noted that the above described modification to circuit 300 to enable its control of inverter circuit 530, in particular the conditioning of the level converter output to provide different voltages for application to the V.sub.REF inputs of the DAC circuit, can also be implemented for the control of inverter circuit 230 by any of control circuits 300 or 400. In such an implementation, the digitally stored waveforms would all have the same magnitude and would have a 90.degree. phase angle difference between the digital waveforms corresponding to waveforms A and B, and between the digital waveforms corresponding to waveforms B and C.

While illustrated embodiments of the invention have been described in which a constant volts/hertz ratio is maintained for each winding of motor 150, the invention is not so limited. With respect to certain motor driven loads, e.g., fans, the magnitude of the load varies with rotational speed. For such loads it may be more efficient to vary the volts/hertz ratio for each motor winding as a predetermined function of rotational speed. With respect to motor drive systems 200 and 500 when operated under control of control circuit 300, such variation of the volts/hertz ratio can be accomplished by modifying level converter circuit 340 to generate the scaling signal in accordance with the predetermined function of rotational speed.

The illustrated embodiments of the present invention provide convenient direction reversal. It should be understood that the present invention, in its broader aspects, is not intended to be limited to the particular signal manipulations disclosed to achieve direction reversal.

While the present invention overcomes the above described problems experienced in the operation of HVAC systems, it will now be apparent to those skilled in the art that the invention can be practiced in other system applications to likewise realize the advantages of the invention.

The foregoing description of preferred embodiments of the invention has been presented for purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed, and modifications and variations are possible in light of the above teachings or may be acquired from practice of the invention. The embodiments were chosen and described in order to explain the principles of the invention and its practical application to enable one skilled in the art to utilize the invention in various embodiments and with various modifications as are suited to the particular use contemplated. It is intended that the scope of the invention be defined by the claims appended hereto, and their equivalents.