Precision Ranging System

The present invention relates generally to reflected signal ranging systems and, more particularly, to an electromagnetic ranging system employing a hybrid of pulsed and continuous wave techniques to achieve very high ranging accuracy.

Present day radar techniques for determining the distance or range to a target have serious shortcomings when extremely high accuracy is required. The inability to give repeatable and highly accurate results has heretofore meant that radar ranging techniques have been ruled out for whole classes of applications which could be conveniently handled using radar techniques, if the accuracy could be improved.

As an example, consider the problem of measuring water height in a river. The classical solution is to place a measuring stick in the river and periodically check it to see where the water line crosses the stick. The classical technique requires a person to periodically check the water height, making this solution infeasible for large scale geodetic surveys. If radar techniques could achieve the desired accuracy, then water height could be quite easily measured by placing a radar transceiver on a bridge or other fixed structure and by measuring the water height as a function of range between the transceiver and the water's surface. The data could then be broadcast to a nearby geodetic station for effectively continuous monitoring of the water height. Regretfully, present day radar is not accurate enough to measure distances on the order of inches or fractions of inches. Hence, radar techniques have not been available to solve the exemplary water height problem.

The present invention provides a precision ranging system using radar techniques which achieves accuracies never before achieved with conventional radar techniques. The invention employs a combination of short pulse ranging and continuous wave processing techniques to measure the range of a noncooperative target with an absolute accuracy of less than plus or minus 0.015 feet or better. According to the invention, a short pulse or burst of electromagnetic energy is transmitted and the return from the target is envelope detected. The magnitude of the return pulse is sensed and fed to a level controller in the transmitter to ensure that the return pulse maintains a constant amplitude. The return pulse is then threshold detected and converted to a square wave which is in turn down converted. The phase of the down converted signal is then measured and yields the range under measurement.

By ensuring that the amplitude of the received pulse remains constant, the invention ensures that the slope of the leading edge of each received burst is constant. Thus, by maintaining the slope constant, the threshold crossing point is held in a constant relationship relative to the transmitted pulse, yielding extremely high accuracy. During the down conversion process, the phase of the return signal is preserved while the time scale is expanded by several orders of magnitude for much greater precision. The invention is virtually immune from range errors due to returns from distributed clutter (such as rain, fog or aerosols) or from objects in the detection pattern side lobes or from multipath distortion. The invention is not limited to any specific portion of the electromagnetic spectrum and can be used with either single or dual antenna (optic) transceivers and with direct or heterodyne detection schemes.

According to the invention, a hybrid technique is employed which achieves the advantage of short pulse, leading edge detection ranging systems in avoiding the clutter and side lobe problems, and also achieves the advantages of continuous wave ranging systems which are not limited in absolute accuracy by currently achievable rise times. Accordingly, a method is provided for determining the range to a target comprising the steps of reflecting a first electromagnetic signal from the target, receiving the first signal and generating a second electromagnetic signal at an energy level determined by the strength of the first received signal. The method continues by reflecting a second signal from the target, receiving the second signal and determining a phase difference between the generated second signal and the second received signal. Finally, the range to the target is determined in accordance with the phase difference. Further, in accordance with the inventive method, the second received signal is down converted by mixing or multiplying with a reference signal of slightly different frequency to produce sum and difference signals, (the difference signal being at a much lower frequency than the carrier). The difference signal, so produced, is used in determining the phase difference.

The invention further provides an apparatus for determining the range to a target comprising a transmitter having a means for providing oscillations at a controlled amplitude and a gating means for modulating the oscillations. The invention further comprises a receiver having means for providing a control signal based on the signal strength of received signals, for controlling the amplitude of the oscillations produced by the transmitter. The receiver further comprises a threshold detecting circuit and a down converter means for providing a shifted frequency signal having a phase portion. The receiver further comprises a means for comparing the phase portion of the shifted signal with the phase portion of a reference signal and for thereby determining the range to the target.

For a more complete understanding of the invention, reference may be had to the following specification and to the accompanying drawings.

FIG. 1 is a diagrammatic view of an exemplary application of the invention;

FIG. 2 is a waveform diagram useful in illustrating the principles of the invention;

FIG. 3 is a schematic diagram of the invention; and

FIG. 4 is a series of diagrammatically simplified waveforms of signals occurring at various indicated points in the circuit of FIG. 3.

With reference to FIG. 1, the invention is illustrated in an exemplary application of measuring the level of water in a river. The radar transceiver of the invention is indicated generally at 10, a distance or range R above the water surface 12. In the exemplary application, transceiver 10 may be mounted above the water surface on a bridge 14. The direct path between transceiver 10 and the water surface 12 is shown in solid line, while a spurious multipath signal is shown in dotted line. The multipath signal, included here to illustrate one of the problems which this invention overcomes, results from multiple reflections. In a continuous wave (CW) system, multipath signals add to the main return, causing phase distortion and ranging errors. It is often difficult to compensate for these errors in a CW system, since the multipath signal is at the same frequency as the main return. Similar range errors are caused by flase returns from objects in the detection pattern side lobes, such as the bridge embankments and by distributed clutter, such as rain, dust or fog between the transceiver and the water surface.

To overcome the problems caused by the addition of extraneous returns from nontarget objects, multipath reflections and clutter, the present invention utilizes a short pulse transmission. FIG. 2 illustrates the envelope of the short pulse transmission with the time scale exaggerated somewhat to better illustrate the rise time of the envelope. More specifically, FIG. 2 illustrates a group of pulsed signal envelopes plotted by signal strength or amplitude (a) as a function of time (t). It will be understood that the waveforms of FIG. 2 are in practice placed upon or used to modulate a carrier frequency. The invention is not limited to any specific portion of the electromagnetic spectrum and is, therefore, useable with carriers at both radio and microwave frequencies and at optical frequencies.

More specifically, FIG. 2 illustrates a transmitted pulse 16 and a series of return pulses 18, 20 and 22. Note that the return pulses 18 and 20 occur a time phi later than the transmitted pulse 16. This time phi represents the time it takes for the pulse or burst of electromagnetic energy to propagate from the transceiver, to the target, and back. Stated differently, phi is the phase difference between the transmitted and return pulses. Given the speed of propagation, the range to the target can be readily determined from this time phi. Return pulse 22 is illustrated as occuring at some time greater than the return pulses 18 and 20. Return pulse 22 may, for example, result from a multipath signal which has undergone several reflections (and, hence, greater transit time) than the main return pulses 18 and 20. Hence, return pulse 22 occurs later in time.

Although exaggerated somewhat for illustration purposes, the transmitted pulse 16 has a leading edge 24 with a positive finite slope. In practice, pulsed waveforms are generated by gating a carrier on and off. Due to physical limitations in the gating circuit, the rise time of leading edge 24 is a finite time. With presently available electronics, it is possible to gate a carrier on with a rise time of 2 nanoseconds from base 26 to peak 28 (see rise time r in FIG. 2). A similar finite fall time is seen in trailing edge 30. The rise time and fall time are, thus, determined by the physical limitations of the electronic gating circuitry. The actual duration of the overall pulse is determined by the length of time the gate remains open. Hence, it will be understood that the shape of the waveforms illustrated in FIG. 2 are merely exemplary and for the purpose of illustrating the principles of the invention and should not be construed as limiting the scope of the claims.

The invention is well adapted for use with noncooperative targets. Generally speaking, noncooperative targets cannot be readily controlled by the radar operator. The water surface 12 is a good example of a noncooperative target inasmuch as the wave motion and river currents affect the reflective surface in often unpredictable ways which cannot be readily controlled. The wave motion and currents produce a scintillation, causing the return pulse to vary in amplitude to a considerable extent. Return pulses 18 and 20 illustrate different amplitudes as might occur due to target scintillation. In comparing the return pulses 18 and 20, it is important to recognize that the rise time r is the same for both and also the same as the rise time of transmitted pulse 16. Notably, however, the slopes of the respective leading edges of pulses 18 and 20 are not the same, since pulse 18 attains a greater amplitude than pulse 20. As another consequence of the difference in amplitudes, the leading edge of pulse 18 crosses a predetermined threshold line 32 sooner than does the leading edge of pulse 20. Pulse 18 crosses the threshold at point X, whereas pulse 20 crosses the threshold at point Y, a slight time later. By comparison, pulse 22 is significantly attenuated and does not cross threshold line 32 at all. Recall that the phase difference or time delay phi between the transmitted pulse and the received pulse is used to compute range R. The time difference between threshold crossing X and threshold crossing Y thus represents error which, if not controlled, degrades ranging accuracy.

The present invention eliminates the timing errors which result from the variance in slope of return pulse leading edges. The invention compensates for this error by automatically adjusting the level of the transmitted pulse so that the slope of the return pulse remains constant. With reference to FIG. 3, the circuit for implementing the invention is illustrated. The invention comprises a transmitter section 34 and a receiver section 36. The transmitter section comprises an oscillator 38 for generating a carrier frequency. The output of oscillator 38 is fed through a level control circuit 40 which is in turn coupled to modulator 42. Modulator 42 drives antenna 44. Although the invention may be implemented for a wide range of carrier frequencies, the invention may be practiced using a CW Gunn oscillator producing a carrier frequency at nominally 32 GHz. Level control circuit 40 may be implemented as a variable attenuator consisting of 2 PIN switches in series. Modulator 42 may also be implemented using a third PIN switch and driven with a modulation of 2 w+2 dw. A representative output waveform is illustrated to the immediate right of antenna 44. The illustrated waveform comprises gated carrier oscillations at a repetition frequency of 2 w+2 dw. Stated differently, the period between pulses is 2 pi/(2 w+2 dw). Modulator 42 forms the leading and trailing edges of the pulse train with the rise time of the leading edge and fall time of the trailing edge determined by the electrical characteristics of the modulator circuit.

The receiver 36 comprises antenna 46 coupled to detector circuit 48. Detector circuit 48 may employ either direct or heterodyne detection schemes; the invention is practical with both. Generally to the right of antenna 46 is a waveform depicting the reflected or received signal exemplary of the signal at point A on the circuit diagram of FIG. 3. The waveform at point A is also shown in FIG. 4, line A. As indicated in FIG. 3, the reflected signal is of essentially the same character as the transmitted signal and has the same repetition frequency. Although not explicity illustrated in FIG. 3, the amplitude of the pulses or bursts of energy at the receiver antenna 46 would normally be somewhat lower than of the amplitude of the transmitted signal. This is due to the energy loss in transmission through the medium and reflection from the target.

The output of detector 48 is illustrated diagrammatically at line B in FIG. 4. In this case, the leading edge 24 and trailing edge 30 of each pulse is illustrated by vertical lines. It will be understood, however, that the rise and fall times are finite and, hence, the leading end trailing edges have a slight slope which is imperceptable in FIG. 4. The output of detector 48 is split at junction 50, with the output from the junction being fed through amplifier 52 to level control circuit 40 and also fed to threshold detector circuit 54. The signal fed through amplifier 52 controls the attenuation or gain of level control circuit 40 in accordance with the strength of the reflected signal which is received by antenna 46. The signal from amplifier 52 thus provides closed loop feedback control of the amplitude of the transmitter carrier, before the carrier is modulated. Preferably, the amplitude of the transmitted carrier is adjusted so that the received signal is on the order of twice the amplitude of the threshold provided by threshold detector 54. For reference purposes, the threshold level is indicated by threshold line 32 in FIG. 4, line B.

Threshold detector 54 produces an output spike or pulse each time the leading edge 24 of the detected waveform crosses the threshold 32. Line C of FIG. 4 illustrates the output of threshold detector 54. This output signal is fed to a flip flop circuit 56 which produces a square wave output, changing state with each spike or pulse from the threshold detector 54. This square wave is shown in line D of FIG. 4 and will be seen to comprise a frequency one-half of the repetition frequency of the transmitted waveform. In other words, the square wave output of flip flop 56 has a frequency of w+dw. Note that the modulation signal applied to modulator 42 has a frequency of 2 w+2 dw.

As will be explained momentarily, the invention utilizes a down-conversion technique for measuring the phase (transit time) of the received pulse which is in turn used to compute the target range. In order to effect down-conversion, the signal from the flip flop 56 is mixed or multiplied with a reference signal whose frequency is slightly offset from or detuned with respect to the frequency of the flip flop output. This mixing or multiplication produces sum and difference frequencies. If the difference or detuning between the flip flop output signal and the reference signal is sufficiently small, then the difference frequency can be made quite low. This is an advantage, since at any given phase angle difference, lower frequencies produce longer and, hence, more easily measured delay times. This phenomenon is discussed more fully below.

In order to produce the necessary reference signal, at a frequency slightly detuned relative to the modulation repetition frequency, frequency synthesizers 58 and 60 are provided. Frequency synthesizer 58 produces a signal of a predetermined frequency of 2 w. Frequency synthesizer 60 produces a signal at a second predetermined frequency of 2 dw. The outputs of synthesizers 58 and 60 are combined in multiplier 62 to produce a sum component 2 w+2 dw, which is fed to modulator 42 to provide the modulation signal. The outputs of frequency synthesizers 58 and 60 are also fed to a pair of divide by 2 circuits 64 and 66, respectively, which in turn provide signals at frequencies of w and dw, respectively. The w reference signal is multiplied with the output from flip flop 56 in multiplier 68 to produce sum and difference frequencies at the output thereof. This output is then filtered in filter 70 to extract the difference component, i.e., the dw portion of the output from flip flop 56. Note that the dw portion of the flip flop output is derived from the reflected signal received in antenna 46. Thus, the dw component contains phase information indicative of the range to and from the target. The dw reference signal from divide by 2 circuit 66 comes directly from the frequency synthesizer and is used as a reference standard against which the dw signal from filter 70 is compared in phase comparator 72. The output of phase comparator 72 is a signal representing the phase difference between dw reference standard and dw received signal. This phase difference yields the range to the target.

The illustrated circuit of FIG. 3 is set up so that the w signal is applied from divide by 2 circuit to multiplier 68, and the w+dw signal is applied to multiplier 68 from flip flop 56. The invention may also be implemented whereby the frequency synthesizers provide a reference of w+dw to multiplier 68 and whereby the transmitted signal is modulated at 2 w so that the flip flop output produces a signal at a frequency w. In either case, multiplier 68 is fed with signals from which the phase difference dw may be extracted.

To demonstrate the operation of the invention, it will be assumed that oscillator 38 is producing a carrier frequency at 32 GHz. It will further be assumed that the carrier is being modulated or gated on and off by modulator 42 to produce a train of repetitive pulses, each pulse having a duration of 5 ns. It will further be assumed that the modulation repetition rate is 6.006 MHz., i.e., the individual gated pulses or bursts of energy occur approximately 166.5 ns apart. Referring to FIG. 3, the modulator 42 is supplied a modulating signal of 2 w+2 dw, which is a representation of the modulation signal in radian frequency (i.e., radians per second). To convert from radian frequency in radians per second to frequency in cycles per second or Hertz, recall that 1 cycle per second (Hertz) is equal to 2 pi radians per second. Thus, in the present example where the modulation repetition frequency is 6.006 MHz., w is 6 pi radians per second and dw is 0.006 pi radians per second. For convenience, frequency will be stated in Hz. throughout the remainder of this example.

The gated signal is transmitted via antenna 44 into free space, where it propagates at the speed of light toward the target (in this case water surface 12). The transmitted signal reflects from the water surface to provide a return signal which is picked up by receiver antenna 46. For purposes of understanding the invention, it will be assumed that any wave motion at water surface 12 is several orders of magnitude slower than the 6.006 MHz. repetition frequency. If this is the case, then the target may be considered as effectively stationary for a great number of transmitted pulses. The surface of water in a river, for example, may have a wave motion which varies at a rate on the order of 1 KHz. Hence, the repetition frequency of the transmitted waveform is greater than 1,000 times faster than the wave motion, and at least 1,000 pulses may be reflected from the water surface before the surface has a chance to change appreciably.

The received signal can be expected to be attenuated due to losses incurred during propagation and incurred at the reflective boundary. Typically the return signal strength can vary by a factor of 1,000 to 10,000. The received signal is detected in detector 48 and the detected signal is fed through amplifier 52 to level controller 40 to provide closed loop feedback control of the carrier before modulation. If the return signal level is high, level controller 40 attenuates the carrier supplied to modulator 42; if the return signal level is low, less attenuation (or even amplification) is provided by level controller 40. In effect, the level controller circuit is responsive to the signal strength of the return signal (i.e., the received signal) to ensure that the received signal remains at a constant signal level or amplitude. Because the modulation repetition frequency is several orders of magnitude higher than the target motion, the circuit tends to lock onto the appropriate signal amplitude and maintain that amplitude constant.

It is important to recognize that signal level control is effected before the carrier is modulated. As explained above, present day electronic circuitry tends to introduce phase shift (time delays) which are inherent to the physical limitations of the components used. This includes electronic circuits for boosting or attenuating signal levels. Due to inherent limitations in the electronic components, the phase shift (time delay) of a signal attenuating or amplifying circuit changes as the degree of attentuation or gain is changed. Hence, the placement of a level control circuit or automatic gain control circuit after modulator 42 would add phase error to the system and would not achieve the result obtained by this invention.

The benefit of maintaining a constant return signal amplitude may be seen in FIG. 2. As explained above, even the finest electronic gating circuits require a finite rise time to go from a closed gate state to an open gate state. For purposes of understanding the invention, it will be assumed that the rise time of modulator 42 is 2 ns. from base 26 to peak 28. Comparing return pulses 18 and 20, both pulses will require 2 ns. to reach peak. However, because return pulse 18 is of a higher amplitude than return pulse 20, return pulse 18 crosses threshold 32 more quickly than return pulse 20. By maintaining the return signal amplitude at a constant level, the invention ensures that all return pulses for a given target range will cross threshold 32 at precisely the same time. The error or uncertainty depicted by the different threshold crossing positions X and Y is, thus, eliminated. Preferably, the return signal strength is maintained at a predetermined fixed level above threshold, twice threshold being presently preferred. One benefit which emerges from the use of threshold detected short pulse signals is the elimination of error due to clutter and multipath signals. These extraneous returns from objects other than the target are typically much lower in amplitude than the main return. Hence, they are readily rejected as being below threshold. Pulse 22 illustrates this in FIG. 2.

Once the return signal has been detected, its output appears generally as illustrated on line B of FIG. 4. The detected signal is compared with threshold level 32, and each leading edge threshold crossing generates a spike shown on line C of FIG. 4. These threshold crossings are used to generate a continuous wave (CW) square wave signal in flip flop 56. Flip flop 56 changes state from positive to negative to positive with each threshold crossing. Hence, the output of flip flop 56, shown on line D of FIG. 4, is a CW signal at a frequency of one-half the modulation repitition frequency. Thus in this example, the CW square wave signal produced by flip flop 56 is at a frequency of 3.003 MHz. Because the CW square wave signal is generated from threshold crossings of the received waveform, this signal conveys information relating to the target range.

In order to extract the target range information, the circuit employs a down conversion technique, a form of coherent integration. It will be recalled that the modulation repetition frequency (6.006 MHz.) was characterized as comprising two components, namely, 2 w and 2 dw. These components are generated by frequency synthesizer 58, supplying the 2 w component, and frequency synthesizer 60, supplying the 2 dw component. These components are mixed in multiplier 62 and the sum component is used to supply the modulation signal. Thus, frequency synthesizers 58 and 60 represent stable reference frequencies, which contain no target range information. The 2 w output of synthesizer 58 is divided by 2 in circuit 64 to provide a reference signal at frequency w, in this case 3.0 MHz. The 3.0 MHz. reference signal and the 3.003 MHz. received signal are multiplied in multiplier 68 and the difference component dw is extracted using filter 70. The difference component dw is at a frequency of 0.003 MHz., i.e., 3 kHz. The dw signal produced by filter 70 still contains the phase information of the return signal received at antenna 46. This phase information is indicative of the target range. Stated differently, the phase of the return signal is preserved in the down conversion process.

In order to extract the phase information indicative of range, the dw reference signal supplied by frequency synthesizer 60 and divide by 2 circuit 66 is compared with the down converted dw signal in phase comparator 72. The dw reference signal is derived directly from frequency synthesizer 60 and does not vary as target range varies. In contrast, the dw signal derived from the return signal received at antenna 46 includes a phase shift caused by the time it took the transmitted signal to propagate to and from the target. By comparing or subtracting the reference dw signal from the down converted dw signal, the phase difference between the two is extracted. This phase difference yields the target range according to the formula phi=2 R/c, where R is the range to the target and c is the velocity of light. In the down conversion process, the phase information, originally carried by the 6.006 MHz. pulse train, is preserved as the signal is converted to a 3 kHz. frequency. The down conversion process results in considerably greater measurement accuracy, as a one degree phase variation at 3 kHz. is 2,000 times longer than a one degree phase variation at 6 MHz. In other words, this is a coherent integration process in which phase information is integrated over an interval appreciably longer than the period between transmitted bursts.

While the invention has been illustrated in conjunction with a water height problem, it will be recognized that the principles of the invention may be employed in a much wider range of applications. The invention may, for example, be useful in determining the level of oil in a super tanker. In addition, the invention may find utility in the construction and manufacturing industries and other disciplines where accurate measurements are required. As stated above, the invention is useable over a wide range of frequencies throughout the electromagnetic spectrum, including optical frequencies. In selecting the modulation frequency, consideration should be given to the rate at which the noncooperative target can be expected to change its position. Preferably, the modulation repetition rate should be on the order of 1,000 times greater than the target movement rate, although the invention will work at both lower and higher modulation repetition frequencies than this. In selecting the amount of reference signal detuning, i.e., in selecting dw, it will be recognized that the smaller dw is made, the lower the down conversion frequency becomes; and, hence, the longer the coherent integration time becomes. At extremely long integration times, the movement of the noncooperative target may present a problem, and at shorter integration times (larger values of dw) accuracy is degraded. Hence, the selection of the appropriate down conversion frequency depends upon the nature of the problem, the range involved, the accuracy desired, and the anticipated rate of movement of the target.

While the invention has been illustrated and described in its presently preferred embodiment, it will be understood that the invention is capable of modification without departing from the scope of the invention as set forth in the appended claims.