Power Supply for Arc Lamps

This invention provides a current controlled AC/DC power supply for arc lamps, such as HMI lamps, mercury vapor lamps, sodium vapor lamps, and the like. Such lamps are used in theatrical productions, on cinematographic stages, for the production of TV shows, in industrial applications, for lighting sporting events and for street and outdoor lighting in general, to name only a few applications. When used in cinematographic, theatrical, and TV applications and sometimes when used in industrial or sporting applications, the lamp selected must have a correct light spectrum characteristic (or color temperature), which often means that it must have the same light spectrum (color temperature) as the sun so that colors appear natural. In such cases, the current supplied by the power supply to the lamp must be carefully controlled in order to provide precise regulation of lamp color temperature.

The prior art power supplies have a number of drawbacks. They tend to be bulky, expensive and slow to ignite or re-ignite the lamp being powered. In the case of power supplies used for theatrical work, cinematographic work or in a TV studio, the power supplies should be preferably portable (and, in the case of the prior art, they were, at best, semi-portable because the power supplies were quite heavy), they should generate flicker free light and need to be able to ignite and re-ignite the lamp quickly. In industrial applications, the ability to re-ignite a lamp quickly can also be very important. In an effort to save energy costs, many factories have switched from traditional lamps to arc lamps due to their greater energy efficiency. Power outages can occasionally occur, however, and since the power supplies used to power such lamps have required a comparatively long time to restart or re-ignite the lamps, the factory can be without light for a considerable length of time. If it takes more than a few minutes to restart the lamps, then the loss of production at the factory can outweigh the savings from using such lamps.

HMI lamps and other similar lamp types should not be operated on DC (Direct Current) because DC causes erosion of the electrodes resulting in rapid destruction of the bulb. If the lamp is powered by a sinusoidal AC (Alternating Current) waveform, the erosion problem is overcome, but the resulting light emitted varies sinusoidally resulting in the phenomenon known as flicker. Those skilled in the art realize that flicker is undesirable, especially in cinematographic applications. If a square waveform is utilized to power the lamp then both the erosion problem and the flicker problem are overcome. U.S. Pat. No. 4,485,434 teaches how to generate a squarewave using a bridge circuit.

Readily available power sources supply sinusoidal AC. For example, the 120 volt 60 Hertz power available in American homes and industry is sinusoidal AC. On cinematographic stages DC power has traditionally been available. It is well known to convert sinusoidal AC to DC by means of a simple rectifier and filter device. A converter device is then used to obtain the proper current and voltage for the lamp.

Preferably, a power supply for arc lamps should be able to be powered from either AC or DC sources of wide voltage ranges. In particular, the power supply should be functional even when the input voltage is less than the voltage required to ignite and run the arc lamp. Moreover, the power supply should be light weight, cost effective to manufacture and yet provide sufficient power resources to quickly ignite or re-ignite the arc lamp.

Briefly, and in general terms, the instant invention provides a power supply for an arc lamp, which has a DC converter with an input and output. The DC converter is capable of increasing the voltage received at its input and supplies the increased voltage to its output. A current sensing circuit is provided for controlling the amount of current delivered to the output. An output "H" bridge is coupled to the output of the DC converter for generating a squarewave.

In the disclosed embodiment, the bridge circuit preferably includes four switches, preferably MOSFETs, which are turned on and off so that the output load on the bridge (the lamp) receives power in one polarity or the other alternatively. The converter preferably includes a transformer driven by a plurality of transistors, also preferably MOSFETs. Bridge configurations and push/pull arrangements are disclosed. The transformer output is then rectified and filtered to create a DC source for the "H"bridge. The transistors of the converter are turned on and off by a pulse width modulator control circuit in order to vary the DC output power in accordance with lamp requirements. The lamp current is sensed and supplied to an amplifier which then supplies a signal to the pulse width modulator control circuit forming a control loop to keep the lamp voltage and current essentially fixed. This allows the lamp color temperature to be accurately regulated and the lamp to be operated without varying intensity level regardless of input power variations. Further, a wide range of either AC or DC power sources may be utilized as the original power input for the device. The output voltage can be either higher or lower than the source voltage to the power supply. The power supply, even when sized to power a 4000 watt arc lamp, can be easily carried by one person.

The novel features which are believed to be characteristic of the invention are set forth in the appended claims. The invention itself, however, both as to its construction and its method of operation and use, together with the objects and features thereof, will be best understood from the following detailed description of a number of embodiments, when read in conjunction with the accompanying drawing, wherein:

FIG. 1A is a circuit diagram of a first embodiment of a DC converter device;

FIG. 1B is a circuit diagram of a second embodiment of a DC converter device;

FIG. 1C is a circuit diagram of a third embodiment of a DC converter device;

FIG. 1D is a circuit diagram of a fourth embodiment of a DC converter device;

FIG. 1E is a circuit diagram of a fifth embodiment of a DC converter device;

FIG. 1F is a circuit diagram of a sixth embodiment of a DC converter device;

FIG. 1G is a circuit diagram of a seventh embodiment of a DC converter device;

FIG. 2A is a circuit schematic of the gate drive control circuit for the DC converter of FIGS. 1A through 1G;

FIG. 2B shows modifications to the gate drive control circuitry of FIG. 2A, which modifications are used with the DC converter embodiment of FIG. 1B;

FIG. 2C shows modifications to the gate drive control circuitry of FIG. 2A, which modifications are used with the DC converter embodiment of FIG. 1C;

FIG. 3 is a circuit schematic of an "H" bridge which is connected to the output of the converters of FIGS. 1A through 1G;

FIG. 4 is a circuit schematic of an embodiment of the control circuit for the "H" bridge circuit of FIG. 3; and

FIG. 5 is a schematic diagram of an arc lamp head.

FIG. 1A is a schematic diagram of a first embodiment of a DC converter. Incoming AC or DC is applied via Inputs 1 & 2, a circuit breaker CB, and a fuse F to a full wave diode rectifying bridge 1. The polarity of incoming DC applied at Inputs 1 & 2 is not particularly important since rectifying bridge 1 will correct the polarity, if required. Rectifying bridge converts the incoming AC to DC (if the input is AC) or merely passes incoming DC. The DC output from the bridge 1 appears at a bus 3, 4, and is smoothed, if necessary, by a filter 2A comprising a choke inductor and a capacitor.

The DC passing or converted by rectifying bridge 1 is thereafter converted to AC by an oscillator circuit which includes MOSFETs (Metal Oxide Silicon Field Effect Transistors) 5-8. The AC voltage is stepped up by an autotransformer 9 and thereafter reconverted back to DC, which appears at an output bus 19, 20.

The four MOSFET switches 5, 6, 7 and 8 of the oscillator circuit are arranged in a bridge configuration in which only one pair of MOSFETs are gated on and therefore conducting at any given time (that is, at most, only one pair of MOSFETs, either MOSFETs 5 and 6 or 7 and 8 are conducting at any given time) causing current to alternate in the winding of an autotransformer 9. The conduction by the MOSFET switches 5-8 is pulse width modulated at a high frequency (preferably greater than 15 KHz so as to be inaudible and more preferably about 30 KHz) by controlling the gates of MOSFET switches 5-8 by a gate drive transformer 10 which is driven a control circuit which will be subsequently described with reference to FIG. 2A. At 30 KHz, gate drive transformer 10 is a convenient means of controlling the gates of MOSFETs 5-8 with the pulse width modulated signal available at nodes 25 and 26. Those skilled in the art will appreciate that other means of driving the gates can be used, including using optologic devices in a manner similar to that used (and subsequently described) with respect to MOSFETs 50-53 in FIG. 3.

Autotransformer 9 steps up the voltage received at its input and the resulting alternating current flows through and is rectified by diodes 11 and 12 connected at the output of autotransformer 9. A filter, in this case a "T" filter comprising an inductor 14, capacitor arrangement 15 and inductor 16, filters and smooths the DC generated by diodes 11 and 12.

Capacitor arrangement 15 is preferably a 1000 mfd 300 VDC capacitor 15A in series with a 5 ohm resistor 15B, which in turn are connected in parallel with a film capacitor 15C having high ripple current handling capabilities. Inductor 14 is preferably a 300 MicroHenry choke and inductor 16 is preferably a 100 MicroHenry choke. The size of the capacitor 15A is preferably relatively large while the size of the inductors is preferably relatively small. The reason for this relationship will be discussed subsequently.

During the time MOSFETs 5-8 are not conducting, current can be supplied to the filter 14, 15, 16 through an additional MOSFET 17 and diode 18. DC flows from output 19 of the DC converter to the input of an "H" bridge, which will be subsequently described with reference to FIG. 3. The DC returns via return 20 and then flows through a current sensing resistor 21 back to bus 4. The voltage drop generated across the current sensing resistor 21 is applied to the gate drive control circuit of FIG. 2A and, as will be seen, is used to control the pulse width modulation applied to the MOSFET gates 5-8 through transformer 10. Since the voltage drop across sensing resistor 21 is proportional to the current being supplied by the power supply, the pulse width modulation of MOSFETs 5-8 by the gate drive circuitry of FIG. 2A effectively controls the amount of current delivered by the power supply to the lamp. As previously mentioned, controlling the current to the lamp means that its color temperature is being controlled.

MOSFET switch 17 is gated on when the output voltage of the DC converter is higher than its input voltage and off when the output voltage is less than its input voltage. Current is pumped to the output of the converter when MOSFET switch 17 is on (i.e. when the output voltage is higher than the input voltage) during portions of the flyback of autotransformer 9 thereby increasing the efficiency of the circuit. Thus, MOSFET switch 17 reduces the amount of power required to be converted by the bridge MOSFETS 5, 6, 7 and 8 during normal running of the lamp when gated on or allows operation of the lamp at reduced voltage levels during warm up when gated off. As will be discussed subsequently, MOSFET 17 also provides additional power during the re-ignition of a arc lamp, thereby increasing the speed by which such lamps can be re-ignited by the power supply.

The input voltage to the DC converter can be as low as 90 volts and it will still function properly. Lower voltages can be accommodated, if desired, by changing the winding ratio of autotransformer 9 to yield a higher voltage step up. The winding ratio of autotransformer is preferably 2:1:2, but these ratios can, of course, be varied. The maximum voltage which can be accommodated is determined by the ability of the various components to withstand higher voltages. Either AC or DC can be applied to the Inputs 1 and 2. Thus, the power supply is capable of using either AC or DC in a wide range of possible voltages as its source of power.

FIGS. 1B through 1G are schematic diagrams of additional embodiments of a DC converter. Those skilled in the art will appreciate that the embodiments of FIGS. 1B through 1G bear certain similarities to the embodiment of FIG. 1A and therefore components which perform essentially the same function as heretofore described bear the same identification numerals. Components whose functions have changed somewhat are shown with a prime after their identification number and these components will be described in the following description.

Turning to FIG. 1B, there is shown a circuit schematic of a second embodiment of a DC converter. In this second embodiment, MOSFETS 6', 7' and 8' are arranged in a push/pull arrangement with autotransformer 9. In operation, MOSFETs 6', and 8', are alternatively switched on to drive current through the autotransformer 9 onto the load. MOSFET 7' is on continuously when the unit is in normal operation, i.e., the lamp to which the "H" bridge of FIG. 3 is connected is hot and the output voltage from the power supply is high. When the output voltage must be low, MOSFET 7' is switched on and off at a rate twice that of either MOSFET 6'or MOSFET 8'. That is, the pulse width modulation signal which is used to drive MOSFETs 6' and 8' is also used to drive MOSFET 7' such that MOSFET 7' is on when either MOSFET 6' or 8' is on. By using pulse width modulation control coupled to nodes 23', 25 and 26, the output voltage can be controlled to be lower than that of the input voltage to the power supply.

As in the case with the fifth MOSFET 17 described with reference to FIG. 1A, MOSFET 7' in the embodiment of FIG. 1B is switched on when the output voltage is approximately equal to the input voltage.

In operation, when MOSFET 6' conducts, current flows from MOSFET 7' (which is also conducting) into autotransformer 9 and through MOSFET 6' to the input return bus 3. Current also flows through diode 11 to the load. At the end of the duty cycle for MOSFET 6', MOSFET 6' turns off and the current stops flowing through it. However, the load current can still flow through diodes 11 or 12 as required. MOSFET 8' turns on at the next duty cycle period and current is reversed in autotransformer 9 and flows from MOSFET 7'into autotransformer 9 and through MOSFET 8' to the input return. It also flows through diode 12 and to the load. The turns ratio of the autotransformer of FIG. 1B is preferably 1:1:1:1, but these ratios can, of course, be varied.

The control circuit for MOSFETs 6', 7' and 8', which will be subsequently described with reference to FIGS. 2A and 2B, senses the current and voltage requirements and adjusts the duty cycles of the MOSFETs accordingly.

Turning now to FIG. 1C, this is a circuit diagram of a third embodiment of the DC converter. This embodiment is similar to the embodiment of FIG. 1B, but in this case MOSFET 17 is employed in the same manner as it is in FIG. 1A. Thus, MOSFET 17 is switched on whenever the output voltage of the power supply is approximately equal to its input voltage. MOSFET 7' can be controlled in the same manner as it is with reference to FIG. 1B or it can merely be switched on only when either MOSFET 6' or MOSFET 8' are also switched on. The reason for this is that in the embodiments of FIGS. 1A and 1C, MOSFET 17 supplies additional current through the filter 14, 15, 16 during the time the MOSFETs 5-8 are not conducting (in the case of FIG. 1A) or during the time MOSFETs 6' and 8' are not conducting (in the case of FIG. 1C). This function of MOSFET 17 is generally handled by MOSFET 7' in the embodiment of FIGURE 1B; however, instead of supplying the extra current at the outputs of diodes 11 and 12 as is done in the case of FIGS. 1A and 1C, the extra current is supplied via the autotransformer 9 in the embodiment of FIG. 1B. This can be a drawback if the autotransformer 9 has too high of an impedance -- it could interfere with the ability of the power supply to re-ignite a lamp since the desired surge of current might not be available. See the discussion near the end of this patent pertaining to lamp re-ignition.

Thus, the embodiment of FIG. 1B is probably less efficient compared to the embodiments of FIGS. 1A or 1C; however, on the other side of the coin, the embodiment of FIG. 1B is somewhat less expensive to construct since it uses a lesser number of MOSFETs. The winding ratio of the autotransformer 9 is again preferably 1:1:1:1, but these ratios can, of course, be varied.

FIG. 1D is a schematic diagram of a fourth and presently preferred embodiment of the DC converter. In this embodiment, MOSFETs 6' and 8' operate in push/pull fashion, but instead of using an autotransformer 9, a transformer 9'with isolated secondary is employed. MOSFETs 6' and 8' are connected to the primary winding which has a center tap connected to the supply bus 4. The secondary of transformer 9' is coupled via diodes 11 and 12 to filter 14, 15, 16. A center tap on the secondary side of transformer 9' is coupled to the return of bus 4 via a diode 13 and is also coupled to MOSFET 17. Again, MOSFET 17 is gated on whenever the output voltage exceeds the supply voltage and MOSFETs 6' and 8' are controlled by the pulse width modulation control circuit of FIG. 2A.

In operation, a voltage is induced in the secondary of transformer 9' which is higher than the input voltage which is applied through MOSFET 17. As in the case of the previously described embodiments, MOSFET 17 is turned off to allow the output voltage of the supply to decrease below its input voltage, when required. When MOSFET 17 is turned off, diode 13 connects the secondary center tap to return bus 4 allowing energy to flow in the secondary.

A possible variation of this embodiment would be to pulse width modulate MOSFET 17 at twice the rate of MOSFETs 6' and 8' to cause the output voltage to drop below the input voltage.

As those skilled in the art would appreciate, the turns ratio of the transformer determines the maximum output voltage. We prefer to use a transformer with turns ratio of 1:1.8 (primary to secondary).

FIG. 1E depicts a schematic diagram of a fifth embodiment of the DC converter. This embodiment is quite similar to the fourth embodiment, except that the MOSFET 17 is coupled via diode 18 to the outputs of commonly connected diodes 11 and 12 as opposed to being coupled at the center tap of the secondary of transformer 9'. As in the case of the preceding push/pull embodiments, MOSFET 6' and 8' are pulse width modulated to control the output voltage and current. MOSFET 17 is switched on during normal hot lamp operation, that is, when the output voltage of the power supply exceeds the input voltage. When the output voltage needs to be lower than the input voltage, MOSFET 17 is switched off. The winding ratio of transformer 9' is preferably at least 1:2.

FIG. 1F depicts a sixth embodiment of the DC converter which is similar to the embodiment of FIG. 1A in that a full bridge series of MOSFETs 5-8 are used at the input to transformer 9', but transformer 9' in lieu of being an autotransformer has an isolated secondary similar to the embodiments of FIGS. 1D and 1E. By using the full bridge arrangement of MOSFETs, this avoids the necessity for a center tap on the primary of transformer 9'. The winding ratio is preferably 1:1.8.

FIG. 1G depicts yet another embodiment of the DC converter, which is generally similar to the embodiment of FIG. 1F, up to the input of transformer 9'; but the output of the transformer 9' and the connection of MOSFET 17 is made similar to the embodiment of FIG. 1E. The winding ratio is preferably 1:2 or greater.

FIG. 2A is a schematic diagram of an embodiment of the gate drive control circuit. This circuit is preferably powered via supply lines 31 and 32 with stabilized D.C. voltage sources which are isolated from each other and from the main power circuits. Indeed, a power supply having preferably four isolated 15 Volt D.C. outputs is connected as follows: One output is connected at lines 3 & 4: a second output is connected at lines 31 & 4; a third output is connected at lines 32 & 24; and the fourth output is connected at lines 64 & 20 (see FIGS. 3 & 4).

An amplifier Integrated Circuit (IC) 25 receives a portion of the voltage drop generated across current sensing resistor 21 (FIGS. 1A through 1G), and amplifies it. The portion received is controlled and adjusted by a voltage divider 46 which includes a pot 45. The output of IC 25 is applied to a pulse width modulating IC 26, the outputs 27, 28 of which are inverted and buffered by buffer ICs 29 and 30. The outputs of buffer ICs 29, 30 drive the primary winding of gate drive transformer 10 for the embodiments of FIG. 1A, 1F and 1G. The outputs of buffer ICs 29 and 30 drive the MOSFETs 6' and 8' directly in the embodiments of FIGS. 1B-1E. In either case the current supplied by the power supply is controlled by changing the period of time the pairs of MOSFETs 5 & 6 and 7 & 8 (in FIGS. 1A, 1F and 1G) or the MOSFETs 6' and 8' (in the other figures) are on, i.e., by changing the width of the pulses provided by the gate drive circuitry to the gates of the MOSFETs.

The preferred type numbers (model numbers) and manufacturers of ICs 25 & 26, and indeed of all the major ICs used in the disclosed power supply are listed in Table I.

The voltage on bus 3, which is the input voltage to the DC converter, is applied via a voltage divider formed by resistors 40, 41 to one input of a comparator IC 35. The divider supplies a portion of the voltage on bus 3 to IC 35, the portion being within the normal input range of comparator IC 35. Similarly, the voltage outputted by the DC converter at node 22 is divided by resistors 42, 43 and applied to the other input of IC 35. The state of the comparator IC then indicates whether the input voltage is higher or lower than the output voltage of the DC converter. The output of comparator IC 35 is applied via a buffer IC 38 to an optologic isolator IC 39. The output of the isolator IC 39 is amplified by a buffer IC 44 and applied to the gate 24 and source 23 of MOSFET transistor 17 (FIGS. 1A and 1C-1G) so that it is turned on or off in response to the comparative levels of the input and output voltage of the power supply. As previously discussed, MOSFET 17 is gated on when the output voltage of the DC converter is greater than its input voltage and off when the output voltage is less than the input voltage.

FIG. 2B depicts another embodiment of the gate drive control circuit. This embodiment is similar to the embodiment of FIG. 2A, but in this case the output from the optologic isolator IC 39 is controlled not only by the results of the comparison done at IC 35, but is also controlled according to the state of buffer ICs 29 and 30. This embodiment may be used in connection with the second embodiment of the DC converter shown in FIG. 1B to control the states of MOSFETs 6', 7' and 8'. For example, and with respect to FIG. 1B, it will be recalled that MOSFET 7' is turned on whenever either MOSFET 6' or MOSFET 8' is turned on. This function is accomplished by the connections from buffer ICs 29 and 30 through OR summing junction 35A to invertor 38. Also, MOSFET 7' in FIG. 1B is also turned on when the output voltage exceeds the input voltage of the power supply and this function is controlled by the state of comparator IC 35 as it, too, is connected via OR gate 35A to buffer 38.

With respect to the embodiment of FIG. 2C, this circuit is also similar to the embodiment of FIG. 2A. This control circuit can be used to control MOSFETs 6', 7', 8' and 17 of FIG. 1C. Naturally, MOSFET 7' in FIG. 1C must be switched on whenever MOSFETs 6' or 8' are on, and this is accomplished through the action of OR gate 35A (the OR summing juction 35A of FIG. 2B and the OR gate 35A of FIG. 2C can be used interchangably). The output of OR gate 35A is connected via an invertor 38A, an optologic IC 39A and a buffer IC 44A to node 23' (which controls the gate of MOSFET 7' ). Otherwise, the circuit of FIG. 2C is the same as the circuit of FIG. 2A.

The schematic diagram of "H" bridge circuit is shown in FIG. 3. MOSFET transistor switches 50, 51, 52 and 53 are alternately turned on and off in pairs so that the DC converter output current applied at bus 19, 20 is caused to flow in alternating directions through the lamp head 95 (FIG. 5) which is connected at output terminals 54 and 55. MOSFETs 50 and 51 cause the lamp current to flow in one direction and MOSFETs 52 and 53 cause it to flow in the other direction. The gates of the MOSFETs are driven by buffer ICs 56, 57, 58 and 59. Buffer ICs 56 and 59, which drive the lower MOSFETs 51, 53 in the "H" bridge, are driven alternatingly at nodes 60, 61 directly from a bridge control circuit which will subsequently be described with reference to FIG. 4. The buffer ICs 57, 58 for the upper MOSFETs 52, 50 are driven by optologic isolators ICs 62 and 63. The bias power for buffer ICs 57 and 58 and the output side of the optologic isolator ICs 62 and 63 is preferably derived from the fourth output of the aforementioned power supply at node 64. This bias power is stored in capacitors 65 and 66 which are charged through diodes 67 and 68 and current limiting resistors 69 and 70. This charging action occurs when the corresponding lower MOSFET is switched on. The optologic amplifier devices 62, 63 are alternatingly driven by the "H"bridge control circuit of FIG. 4.

An output clamping circuit is connected across terminals 54 and 55. It includes a capacitor 80, diodes 81 and 82 and a bleeder resistor 83. This clamp circuit protects MOSFETs 50-53 from voltage transients and spikes which can and will occur at terminals 54 and 55. Such spikes arise from the fact the arc lamp 100 typically is installed in a head 95 (FIG. 5) which has inductive components therein which generate voltage spikes when driven with a squarewave. A positive going spike is shunted to capacitor 80 by diode 81 or 82. The charge on capacitor 80 is maintained by coupling capacitor 80 to capacitor 15 (FIG. 1) via node 22 and resistor 83. Positive going spikes will charge capacitor 80 to a higher potential than that which normally exists on capacitor 15, but resistor 83 will discharge the difference before the next spike occurs.

Turning now to FIG. 4, which is a schematic diagram of the control circuit for the "H" bridge of FIG. 3, this control circuit has a timer IC 73 which drives a flip-flop IC 74. The frequency outputted by IC 73, nominally 60 Hz, is set by resistors and capacitor arrangement 75. The outputs of IC 74 are connected to inverting buffer ICs 76 and 77. These outputs and power supply lines 64, 20 are coupled to the optologic isolators ICs 62 and 63 of FIG. 3 via control busses 71 and 72. Thus, the outputs of the inverting buffer ICs 76, 77 drive the various inputs of the "H" bridge circuit, that is, the inputs of buffer ICs 56 and 59 and the inputs of optologic amplifier ICs 62 and 63 (FIG. 3).

The output applied to the lamp at terminals 54, 55 is therefore 60 Hz squarewave. Since the turn off times of the MOSFETs 50-53 is longer than there turn on times, the output at terminals 54, 55 momentarily shorts at each transition. This shorting can help to reduce voltage spikes which are generated when the current is quickly switched in an inductive load such as head 95. Spikes which still occur are handled by the previously described clamp circuit. The maximum positive and negative voltage of the squarewave is equal to the voltage on bus 17, 18 (less the voltage drops across the conducting MOSFET switches 50, 51 or 52, 53).

Assuming that the power supply is connected to a 120 volt source (either AC or DC) and to an arc lamp head 95 such as that depicted in FIG. 5, the voltage which must be generated to efficiently ignite the arc lamp 100 is on the order of 230 volts (as an AC squarewave, preferably modified as previously discussed). At this point, MOSFETs 5-8 or MOSFETs 6' and 8' as the case may be, will be oscillating and MOSFET 17 (MOSFET 7' in the case of FIG. 1B) will be gated on to pump additional power into the lamp. At ignition, the output voltage will drop to approximately 30 volts, causing MOSFET 17 to turn off (and cause MOSFET 7' in the case of FIG. 1B to switch on and off in time with MOSFETs 6' and 8'.) The output voltage from the power supply will rise to approximately 120 volts in 30 to 40 seconds. As the output voltage of the power supply exceeds its input voltage, MOSFET 17 (MOSFET 7' in the case of FIG. 1B) is again turned on by its gate control circuit and MOSFET 17 (MOSFET 7', FIG. 1B) will then again supply additional power to the arc lamp. The voltage will continue to rise to a steady state condition where the output voltage is approximately 210 volts. The time required to ignite a 4000 watt arc lamp (model Daymax DMI 4000 manufactured by ILC) and come to full output voltage is approximately 90 seconds or less. Those skilled in the art will appreciate that this is much faster than with prior art power supplies.

If the arc lamp 100 is de-energized or becomes extinguished, and while it is still physically hot (i.e. it only was recently de-energized or extinguished), the arc lamp 100 can be brought back to full power (and therefore full intensity light) in approximately ten seconds. In approximately ten seconds, capacitor 15 is recharged to the maximum voltage available from the power supply, which will likely be 235 volts or greater. At the same time, MOSFET 17 is on, thereby permitting the power supply to supply the necessary surge of current to re-ignite the arc lamp which is supplied when the re-ignite switch 90 is briefly closed and thereby connecting the voltage at output terminals 54 and 55 through a resistor 91 to an ignition coil in head 95. The charge stored in capacitor 15 is dumped very quickly into the lamp, causing it to quickly re-ignite. Since the inductance of inductor 16 is relatively small, it offers little impedance to the surge of current provided by capacitor 15. The lamp 100 will promptly re-ignite and come to full power if the user waits approximately 10 seconds before momentarily closing switch 90. Those skilled in the art will appreciate that prior art supplies typically require more than one minute to re-ignite an arc lamp.

The semiconductor devices used in the FIGURE include both CMOS and TTL type devices and therefore those skilled in the art will appreciated that appropriate level shifting devices will also have to be employed. Alternatively, the devices can be changed to use, for example, exclusively CMOS or TTL devices.

The invention has been described in connection with a number of embodiments and certain modifications have been mentioned. Further modifications will become apparent to those skilled in the art. Therefore, the invention is not intended to be limited to the disclosed embodiments, except as required by the appended claims.