Operational Amplifier Having Low DC Current Input Circuit

The present invention relates in general to operational amplifier circuits and is particularly directed to a circuit configuration for effectively minimizing the DC current drawn by the input of the amplifier.

The electrical transmission links by way of which input terminals of an operational amplifier are coupled to upstream circuitry normally include resistive current paths, thereby resulting in an IR drop across such paths which alters or shifts the actual voltage levels expected to be seen at the amplifier input terminals. It is desirable, therefore, to minimize the DC input current drawn by the amplifier and thereby improve its performance. One scheme for reducing the input current employs a Darlington-connected transistor pair, with the input transistor of the Darlington pair being biased at a very low current level. Unfortunately, this low level biasing approach dramatically reduces the signal response of the circuit for large magnitude inputs.

More particularly, with attention directed to FIGS. 1 and 2 of the drawings, there are shown respectively a schematic illustration of a Darlington-connected pair of bipolar transistors Q1 and Q2 coupled between an input terminal IN and a bias current sink 42, and the topological integration of the transistors Q1 and Q2 in a dielectric isolated region 20 in a semiconductor wafer.

The Darlington pair is comprised of an input NPN transistor Q1, the base 12 of which is connected via link 10 to an input terminal IN. The collector 11 of transistor Q1 is connected in common with the collector 21 of NPN transistor Q2 and to a collector bias link 31. The emitter 13 of transistor Q1 is coupled via a link 32 to the base 22 of transistor Q2. The emitter of transistor Q2 is coupled via a link 34 to a current sink I2'. The connection of the emitter of transistor Q1 and the base of transistor Q2 is denoted by node a and is coupled over link 33 to a current sink (e.g. diode-connected transistor) 42. Links 10, 32, 33 and 34, which are typically formed of conductive interconnect material (e.g. aluminum, doped-polysilicon) disposed atop a dielectric insulator layer (e.g. SiO.sub.2), are illustrated schematically in FIG. 2 to simplify the drawings. Typically, the current sink 42 is disposed in an island region 30 separate and isolated from the island region 20, as shown topologically in FIG. 2. As a result, the link 33 between node a and the current sink region 42 has a substantial length and passes over or is adjacent to a ground track or grounded semiconductor material, to induce a parasitic capacitance C.sub.p between link 33 and ground.

When a large (negative) transient is applied to the input terminal, the response at the emitter 32 (node a) of transistor Q1 will be affected by the voltage developed across parasitic capacitance C.sub.p, which must be discharged. Because, the DC current level for accomplishing this task has been set at a very low level (I.sub.1 ' corresponding to the current sink 42 connected between node a and ground), the response time of the circuit is significantly impaired.

An alternative approach to the low level bias scheme shown in FIGS. 1 and 2 is to substitute a junction field effect transistor (JFET) in place of the bipolar input transistor. At high input temperature operation (+125.degree. C.) however, the JFET limits the input current to only several nanoamperes and its incorporation into the bipolar integrated circuit structure requires an expensive increase in processing complexity.

In accordance with the present invention, the above described drawbacks of conventional proposals for minimizing the DC input current drawn by an operational amplifier are effectively eliminated by a new and improved circuit configuration which incorporates a bias control transistor for node a of the Darlington-connected transistor pair into the same semiconductor region in which the Darlington-connected transistor pair is formed. As a result, the conductive link for interconnecting the emitter of the input transistor to the bias transistor is shortened considerably and maintained in a substantially spaced-apart relationship with respect to ground track or substrate bias. Moreover, all portions of the surface of the dielectrically isolated island are effectively bootstrapped with the (common mode) input signal, whereby the parasitic capacitance from node a to ground is effectively eliminated. This enables the collector current of the input transistor to be reduced significantly relative to the collector current flowing through the output of the Darlington pair (on the order of 1:50 compared with the approximately 1:7 ratio of the configuration of the prior art shown in FIGS. 1 and 2).

To accommodate for device characteristic variations among separate wafer processing runs, a self-balanced bias circuit, containing a transistor pair matched to the bias and Darlington output transistors, is employed as the current source of the operational amplifier. As a result, the ratio of the collector currents of the Darlington-connected pair can be controlled irrespective of saturation currents or sheet resistivity of bias resistances employed for the bias control transistor (connected to node a) and for the self-balanced bias circuit, respecitvely.

FIG. 1 is a diagrammatic illustration of a portion of an operational amplifier circuit showing the parasitic capacitance encountered in a prior art configuration;

FIG. 2 is a diagrammatic illustration of the integrated circuit topology of the circuit diagram of FIG. 1;

FIG. 3 is a diagrammatic illustration of one symmetrical half of an operational amplifier circuit in accordance with the present invention;

FIG. 4 is a diagrammatic illustration of the integrated circuit topology of the operational amplifier circuit diagram of FIG. 3; and

FIG. 5 is a diagrammatic illustration of a self-balanced bias circuit for controlling the bias current for the circuit of FIG. 3.

Referring now to FIG. 3, there is diagrammatically illustrated one-half of an operational amplifier circuit in accordance with the present invention, the wafer topology for which is illustrated in FIG. 4. The other half of the operational amplifier circuit is identical to the configuration shown in FIG. 3 and includes a set of transistors symmetrically interconnected relative to mirror line 3--3' through which current source I.sub.1, diodes 101-103, resistor R1 and current sink I.sub.s are colinear between a positive bias supply line +V and a negative bias supply line -V.

In FIGS. 3 and 4, NPN bipolar transistors Q1 and Q2 are connected as a Darlington pair just as are transistors Q1 and Q2 in the configuration shown in FIG. 1, discussed above. Also, transistors Q1 and Q2 are integrated in the same dielectrically isolated island 20 as shown in FIG. 4. The base 52 of input transistor Q1 is coupled over input link 10 to an input terminal IN. The collector 51 of transistor Q1 is coupled over link 75 to a node 90 to which a link 76 from collector 61 of transistor Q2 is coupled. Node 90 is also coupled to the emitter 93 of an NPN bias transistor Q4 the collector 91 of which is connected to a bias voltage terminal 95 and the base 92 of which is connected to a node 100 to which an input current source I.sub.1 and the anode of a diode 101 are connected. Input current source 50 (to be discussed in detail below in conjunction with the description of FIG. 5) supplies a bias current I.sub.1 for each of the symmetrical halves (only one being shown in FIG. 3) of the operational amplifier.

The emitter 53 of transistor Q1 is coupled over link 54 to node a and to the base 62 of transistor Q2. The emitter 63 of transistor Q2 is coupled over link 82 to the collector 71 of a bias PNP transistor Q3, the base 72 of which is coupled over link 81 to a node 80 and the emitter 73 of which is coupled over link 55 to node a. Link 82 is coupled via link 83 to node 70, to which one end of a resistor R1 is coupled. The other end of resistor R1 is coupled to node 80. Also, diodes 102 and 103 are coupled in series with diode 101 between nodes 100 and 80. Each of diodes 101, 102 and 103 is preferably formed of a diode-connected bipolar transistor, having the same geometry as NPN transistors Q1, Q2 and Q4, for providing a triple voltage drop (3 V.sub.be) between nodes 80 and 100 to match the base-emitter voltage drops through transistors Q1, Q3 and Q4 are to provide temperature compensation therefor. A bias current (I.sub.s) sink 60 is coupled between node 80 and a negative voltage supply link -V, as described above.

As shown in FIG. 4, NPN transistors Q1 and Q2 are formed in the same dielectrically isolated well 20, the major portion of which is N type material (corresponding to the collector regions of transistors Q1 and Q2). At the right-hand portion of island region 20, as viewed in FIG. 4, the island is overlapped with a P type region 30' which forms the P type collector of transistor Q3. The respective N base region 72 and P emitter region 73 of transistor Q3 are formed in the P region 30' so that, in effect, each of transistors Q1, Q2 and Q3 is formed in the same dielectrically isolated island region. The P type collector material of the island region 30' forms a PN junction 59 with N type collector region of island region 20. By virtue of transistor Q4 and the connection of the collector 71 of transistor Q3 through current source 60 to negative supply line -V, PN junction 59 is effectively back-biased. Transistor Q4 maintains the N collector island region 20 at a higher potential than the P collector of island region 30', so that all surface portions of the island 20-30' are effectively boot-strapped with the input (common mode signal). As is the case with FIG. 2, connection links 54 and 55, which respectively interconnect the emitter 53 of transistor Q1 to the base 62 (Node a) of transistor Q2 and node a to the emitter 62 of bias transistor Q3, are shown in schematic form in FIG. 4 to simplify the drawing. Unlike link 33 of the prior art configuration shown in FIG. 2, however, interconnect link 55 does not pass over a grounded portion or a ground link of the integrated circuit, so that the unwanted parasitic capacitance C.sub.p described above in conjunction with the description of FIGS. 1 and 2 is effectively eliminated. By the elimination of the parasitic capacitance, the collector current of the input transistor Q1 of the Darlington pair can be reduced significantly relative to the collector current of transistor Q2.

In a typical Darlington-coupled pair, the ratio of collector currents of the input and output transistors is typically on the order of 1:300. In the configuration shown in FIGS. 1 and 2, however, because of the need of a current sink 42 to discharge the parasitic capacitance C.sub.p, the effective ratio of collector currents is reduced drastically (on the order of 1:7). By incorporating the bias transistor Q3 into the same dielectrically isolated island with transistors Q1 and Q2 in accordance with the present invention, however, the ratio of the collector currents in transistors Q1 and Q2 can be increased to a value on the order of 1:50, nearly an order of magnitude improvement over the prior art.

The ratio of the collector currents in transistors Q1 and Q2 is controlled by the voltage drop across resistor R1 and may be defined as:

The collector currents flowing through transistors Q2 and Q3 may be defined as:

Substituting equations (2) and (3) into equation (1) there is obtained:

Therefore, the magnitude of Current I.sub.1 can be defined as:

In setting the behavioral characteristics of the operational amplifier shown in FIGS. 3 and 4, the ratio of collector currents of transistors Q2 and Q3 is established at a prescribed value. Because of processing variables between successive runs of integrated circuit processing from wafer-to-wafer, the supply current I.sub.1 must be established as a function of the saturation currents of transistors Q2 and Q3. For accomplishing this objective, a self-balanced bias circuit shown in FIG. 5 is employed to generate the current I.sub.1.

In the diagrammatic illustration shown in FIG. 5, current sources I.sub.2 and I.sub.3 correspond to the desired collector currents for transistors Q2 and Q3 of the circuit of FIG. 3. Current source I.sub.2 is connected between a positive voltage supply link (+V) and a link 135 which is coupled to the collector of an NPN transistor Q5 and the base 132 of an NPN transistor Q7. Current source I.sub.3 is coupled between positive supply link 130 and a link 125 which is coupled to the base 122 of transistor Q5 and the emitter 113 of PNP transistor Q6. The collector 111 of transistor Q6 is coupled over link 126 to a negative voltage supply link (-V). Similarly, the emitter 123 of transistor Q5 is coupled over link 136 to the negative supply link (-V). It will be noted that transistors Q5 and Q6 are of the same polarity and geometry and interconnected in the same manner as transistors Q2 and Q3 of the circuit shown in FIG. 3. Because transistors Q5 and Q6 are formed as part of the operational amplifier they are effectively matched with transistors Q2 and Q3 with respect to saturation currents. A negative feedback loop for controlling the current in transistor Q5 and Q6 is formed of NPN transistor Q7 the collector 131 of which is coupled over link 145 to a diode-connected current source transistor Q8, the collector current of which corresponds to the supply current I1. The emitter 133 of transistor Q7 is coupled via a diode (connected transistor) 141 to one end of a resistor R2 and to link 146 to which the base 112 of transistor Q6 is coupled. Like diodes 101, 102 and 103 in the circuit of FIG. 3, diode 141 provides balancing voltage drop (V.sub.BE) and temperature compensation between the emitter 133 of transistor Q7 and the base 112 of transistor Q6. The other end of resistor R2 is coupled to the negative supply link (-V). A separate diode-connected transistor Q9 is coupled in parallel with transistor Q8 and represents the current source I.sub.1 for node 100 in the circuit of FIG. 3.

In operation, transistor Q5 and current source I.sub.2 behave as a common emitter voltage amplifier, while transistor Q7, resistor R2 and transistor Q6 form a buffer and unity gain feedback loop. With the values of current sources I.sub.2 and I.sub.3 being set at a prescribed ratio, then, because of the feedback loop through transistor Q7 between the collector 121 of transistor Q5 and the base 112 of transistor Q6, current I.sub.1 will be adjusted to prescribed value, so as to maintain the current flowing through transistors Q1 and Q2 at the ratio intended by sources I.sub.2 and I.sub.3.

More specifically, if the collector current flowing through transistor Q5 exceeds the value supplied by current source I.sub.2, the voltage at the base of Q7 will be lower than it would be if the collector current of Q5 were equal to I.sub.2. This will cause the voltage at the base of Q5 to be lower, by virtue of the unity gain connection between the base of Q7 and the base of Q5, than it would be if the collector current of Q5 were equal to I.sub.2. This will tend to reduce the collector current of Q5. This interaction will persist until the collector current of Q5 is brought into equality (to within the base current of Q7) with I.sub.2, and maintained there. This, in turn, maintains the current I.sub.1 at the proper value for maintaining the prescribed voltage drop I.sub.1 R.sup.1 discussed above. Conversely, for a decrease in the current flowing through transistor Q5, there will be an increase in the base voltage at transistor Q7, thereby increasing the potential at the base 122 of transistor Q5 and increasing the current flow therethrough, thus causing a decrease in the collector current of Q5, to maintain the proper ratio between currents I.sub.2 and I.sub.3. This feedback balancing action ma be defined by equilibrium equations:

For the circuit shown in FIG. 5, the magnitude of current I.sub.1 may be defined in the same manner that equation (5) was derived, namely

Equating expressions (5) and (8), there is obtained

By equating the values of resistances R1 and R2, expression (9) may be reduced to

Thus, the ratio of the collector currents flowing through transistors Q1 and Q2 can be controlled irrespective of their saturation currents or the resistivity of resistances R1 and R2. Of course, it will be appreciated that it is necessary to constrain the ratio I.sub.2 /I.sub.3 to ensure that I.sub.2 I.sub.sp >I.sub.3 I.sub.sn for the full range of I.sub.sp and I.sub.sn that will be encountered.

As will be appreciated from the foregoing description, by incorporating the input bias transistor into the same dielectrically isolated island and effectively boot-strapping all components of the surface of the island to the common mode input signal, the parasitic capacitance at the emitter of the input transistor is effectively eliminated, so that the response time of the operational amplifier circuit to common mode excitation is significantly enhanced. A reduction in input device bias level of nearly an order of magnitude over that of the prior art is obtained. Moreover, by integrating a self-balancing bias circuit containing a pair of current control transistors effectively matched to the bias and output transistors of the input circuit of the amplifier, the ratio of the collector currents of the Darlington pair can be controlled irrespective of their saturation currents and the actual values of the bias resistances for a particular wafer processing run. In other words, there is effectively matching of the circuit configuration and of the physical and operational parameters of both the input circuit and the bias circuit. It should also be realized that in the circuit configuration shown and described, the polarization of the components (e.g. transistor Q1 is shown as an NPN transistor) and bias potentials therefor are not limited to those depicted. Where the polarities of the transistors are reversed, bias potentials are correspondingly inverted.

While I have shown and described an embodiment in accordance with the present invention, it is to be understood that the same is not limited thereto but is susceptible to numerous changes and modifications as known to a person skilled in the art, and I therefore do not wish to be limited to the details shown and described herein but intend to cover all such changes and modifications as are obvious to one of ordinary skill in the art.