CMOS Bandgap Voltage Reference Apparatus and Method

This invention relates to voltage references, and more particularly, to bandgap voltage references implemented in complementary metal-oxide-silicon integrated circuit technology.

A stable voltage reference immune to temperature and power supply variations is required for high performance analog components produced today. For example, the conversion accuracy of signals from analog to digital and back again in precision coders/decoders (CODECS) is directly dependent on the accuracy of an internal reference. Typically the internal reference is a voltage reference which must be tolerant of power supply voltage variations and noise as well as temperature variations. To keep the cost of CODECS as low as possible, the internal reference should be as physically small as possible and allow for precision trimming of the output voltage, V.sub.BG, if necessary.

A common solution to the internal voltage reference requirement is a circuit known as a bandgap voltage reference. Ideal bandgap voltage references have a predetermined output voltage substantially invariant with variations in temperature by combining the positive temperature coefficient of a thermal voltage (V.sub.T or kxT/q, discussed below), generated by differing voltage drops across forward-biased PN junctions at different current densities, with the negative temperature coefficient of the voltage drop across a forward-biased PN junction (V.sub.BE). In practice, by adding a multiplied thermal voltage, V.sub.T, to V.sub.BE, an output voltage with a predetermined temperature coefficient can be created. Typically, the predetermined temperature coefficient is chosen to be substantially zero.

An exemplary bandgap voltage reference of the prior art is shown in FIG. 2. This simplified bandgap reference 200 is susceptible to power supply voltage (V.sub.DD) variations and noise. A more elaborate variation of this bandgap reference having less susceptibility to power supply voltage variations is discussed in "Analysis and Design of Analog Integrated Circuits", second edition, by P.R. Gray and R.G. Meyer, 1984, pp. 733-737. In FIG. 2, a voltage V.sub.t, which is proportional to V.sub.T, is generated across resistor 201. The temperature coefficient of V.sub.t is also proportional to V.sub.T and is scaled as V.sub.t is scaled to V.sub.T. As will be discussed in more detail in the Detailed Description, below, if each bipolar transistor 203.sub.1 -203.sub.n is substantially identical to bipolar transistor 202 and substantially identical current flows into transistor 202 as into combined transistors 203.sub.1 -203.sub.n, then the current density in transistor 202 is n-times that in each transistor 203.sub.1 -203.sub.n. The differing current density results in a different voltage drop across transistor 202 than across transistors 203.sub.1 -203.sub.n. The difference in voltage drops, designated here as V.sub.t, is forced to appear across resistor 201. To do so, the voltage on the emitter of transistor 202 (node N) must be the same as the voltage on nodeN'. To make the voltage on nodes N and N' the same, fieldeffect transistors (FETs) 204, 205 form a current mirror (not numbered) and FETS 206, 207 form a regulator means (not numbered) which, when coupled to the current mirror, maintains equal output voltages on the source terminals of FETs 206, 207 coupling to nodes N and N', respectively. If FET 206 and FET 207 are the same size with equal current flowing through them from corresponding FETs 204, 205, the gate-to-source voltages of FETs 206, 207 will be the same, resulting in identical voltages on nodes N, N', the outputs of the regulator means. Hence, the difference in the voltage drops across transistors 202 and transistors 203.sub.1 -203.sub.n, V.sub.t, appears across resistor 201.

Since the voltage V.sub.t is dropped across resistor 201, the current through resistor 201, i.sub.t, corresponds to the thermal voltage V.sub.T and has the same temperature coefficient as V.sub.t less the temperature coefficient of resistor 201. More particularly, the temperature coefficient of the current i.sub.t has the temperature coefficient of V.sub.t less the temperature coefficient of resistor 201. In practice, the temperature coefficient of the resistor 201 is much less than the temperature coefficient of V.sub.t. FET 208, responsive to FET 205, mirrors current i.sub.t to produce current I.sub.t. Current I.sub.t passes through resistor 209 to paralleled bipolar transistors 210.sub.1 -210.sub.n, which generate the forward PN junction voltage, V.sub.BE. Transistors 210.sub.1 -210.sub.n correspond to transistors 203.sub.1 -203.sub.n and are substantially the same size. The voltage V.sub.BE adds with the voltage drop across resistor 209 in response to the current I.sub.t, resulting in the bandgap voltage V.sub.BG approximately equaling:

where R.sub.201, R.sub.209 are the resistances of resistors 201, 209, respectively. The where R.sub.201, R.sub.209 pectiThe typcal resistance of resistor 209 is six times that of resistor 201, with resistor 209 consisting of six resistors in series, each resistor having the same resistance as resistor 201.

This bandgap voltage reference 200 suffers from large area requirements due to the 2n+1 bipolar transistors (transistors 203.sub.1 -203.sub.n and 210.sub.1 -210.sub.n and seven resistors (resistors 201 and 209) necessary for the proper generation of the V.sub.t and V.sub.BE voltages. Further, this arrangement suffers from low power supply noise immunity resulting from the FETs 206, 207 of the regulator means having different drain-to-source voltages: the drain voltage of FET 207 is the power supply voltage (V.sub.DD) less a gate-to-source voltage (.apprxeq.1 volt) of FET 205, and the drain voltage of FET 206 is approximately 0.7 volts plus a gate-to-source voltage V.sub.BG (.apprxeq.1 volt). For a five volt power supply, the resulting approximate drain-to-source voltages are 3.3 volts for FET 204, 1 volt for FET 205, 1 volt for FET 206 and 3.3 volts for FET 207. The finite output resistances of the FETs 204, 205, 206, 207 with different drain-to-source voltages cause slight differences in current to flow through FETs 206, 207 and consequently, different voltages on the nodes N, N'. This difference in voltage deviates V.sub.t from the true voltage difference in voltage drops across transistor 202 and transistors 203.sub.1 -203.sub.n. Also, the accurate mirroring of the current i.sub.t by FET 208 is compromised by the finite output resistance thereof. The inaccuracy of current mirroring FETs 206, 207, 208 degrades the performance of the bandgap voltage reference by deviating the output voltage V.sub.BG, and the desired temperature coefficient thereof, from what is desired. The differences in current through FETs 206, 207 and the inaccurate current mirroring by FET 208 are dependent on the power supply voltage V.sub.DD. As disclosed on page 735 of the above-identified reference, the current mirror is compounded and the FETs 206, 207 are made as large as possible to reduce the output resistances thereof in an attempt to reduce the effect the power supply voltage has on the reference. However, different drain-to-source voltages on the FETs remain and, consequently, the immunity of the reference to power supply voltage variations still suffers. Another undesirable result of compounding the current mirrors limits the required power supply voltage (V.sub.DD) to greater than five volts, a common power supply voltage used in integrated circuits.

A new bandgap voltage reference has been invented which requires considerably smaller chip area than bandgap voltage references of the prior art. Further, the output voltage of the new bandgap voltage reference can be easily trimmed for a precision output voltage independent of resistors used therein. This has been achieved generally by having a thermal voltage reference including at least one current mirror having a control current input and output; a regulator means, coupled to the control current input and the output of the current mirror, for providing at least two outputs with substantially the same voltages thereon; a first bipolar transistor coupled to a first one of the outputs of the regulator means; and a plurality of paralleled bipolar transistors coupled to a second one of the outputs of the regulator means through a first resistor; a means, responsive to at least one of the current mirrors, for generating a current to an output node, the current being proportional to current passing through the current mirror; and a second resistor, coupling between the output node and the first bipolar transistor, for generating a voltage proportional to the current from the means. With the output current from the means being proportional to temperature and the bipolar transistors being all of substantially the same size, the voltage on the output node is the output voltage from the bandgap voltage reference and is the sum of the voltage across the second resistor and a voltage developed from the first bipolar transistor.

Further, a means and method for reducing the sensitivity of a bandgap voltage reference to variations in the power supply voltage has been invented. This result has been achieved generally by having a differential amplifier for sensing voltages where a current mirror and a regulator means couple together in the thermal voltage reference portion of the bandgap reference, and a variable voltage power source, responsive to the differential amplifier, for powering the bandgap voltage reference. The voltage from the variable voltage power source is varied in response to the differential amplifier until the sensed voltages are substantially equal.

The foregoing features of this invention, as well as the invention itself, may be more fully understood from the following detailed description of the drawings, in which:

FIG. 1 is a schematic diagram of the improved CMOS bandgap voltage reference with a voltage pre-regulator in accordance with one embodiment of the invention; and,

FIG. 2 is a simplified schematic diagram of a bandgap voltage reference of the prior art.

Referring to FIG. 1, the improved bandgap voltage reference 10 and voltage regulator 11 are shown in schematic form. The voltage regulator 11 will be discussed in more detail below, but it is sufficient to state here that the regulator 11 provides a predetermined voltage V.sub.A derived from the power supply V.sub.DD to power the bandgap voltage reference 10. It is noted here that the designations V.sub.DD and V.sub.A refer to power sources and the voltages thereof and may be used interchangeably.

The bandgap voltage reference 10 has a first current mirror (FETs 20, 21) driving a regulator means (FETs 22, 23) to supply currents I.sub.N1, I.sub.N2 to nodes N1 and N2, respectively. FET 20 is substantially matched to FET 21 and FET 22 is substantially matched to FET 23 so that currents I.sub.N1 and I.sub.N2 are, for purposes here, substantially equal. Further, by having substantially equal currents I.sub.N and I.sub.N2, and with FET 22 substantially the same as FET 23, the gate-to-source voltages for FETs 22, 23 are substantially the same. Hence the voltages on nodes N1 and N2 are substantially the same and form the outputs of regulator means. Current from node N1 (I.sub.N1) passes through resistor 24 and bipolar transistors 25.sub.1 -25.sub.n, whereas current from node N2 (I.sub.N2) and from resistor 26 (I.sub.PTAT) together passes through bipolar transistor 27. For the moment neglecting the current contribution to node N2 from resistor 26 (I.sub.PTAT), which will be discussed in more detail below, and noting that the electrical characteristics and temperatures of transistor 27 and each of the transistors 25.sub.1 -25.sub.n are substantially identical, a voltage, V.sub.t, is dropped across resistor 24 which is proportional to the thermal voltage, V.sub.T. As discussed above in connection with the bandgap reference of the prior art (FIG. 2), due to substantially equal voltages on nodes N1, N2 and a different current density in each of transistors 25.sub.1 -25.sub.n from that in transistor 27 resulting from, for purposes here, essentially identical current flowing in transistor 27 and through the combined transistors 25.sub.1 -25.sub.n, the voltage V.sub.t, across resistor 24 is given by the equation:

where n is the number of transistors 25.sub.1 -25.sub.n. Recognizing that:

where k is Boltzmann's constant (1.3805.times.10.sup.-23 J/.degree.K), T is the temperature in degrees Kelvin, q is the electrical charge of an electron (1.6021.times.10.sup.-19 C). It is noted that the thermal voltage V.sub.t is proportional to absolute temperature, i.e., it has a positive linear temperature coefficient, hence V.sub.t is also proportional to absolute temperature. It therefore follows that the current I.sub.N1, through resistor 24, is also dependent on absolute temperature. More particularly, the temperature coefficient of the current I.sub.N1 has the temperature coefficient of V.sub.t less the temperature coefficient of resistor 24. In practice, the temperature coefficient of the resistor 24 is much less than the temperature coefficient of V.sub.t. Further, since the current mirrors (FETs 20, 21, 22, 23) assure that the current I.sub.N1 is substantially the same as the current I.sub.N2, the current I.sub.N2 is also dependent on absolute temperature:

where R.sub.24 is the resistance of resistor 24.

When including the current into node N2 from resistor 26, I.sub.PTAT, V.sub.t then becomes:

FET 28 proportionally mirrors the current I.sub.N1 through cascode connected FET 29 to produce current I.sub.PTAT. The size of FET 28 is M times the size of FETs 20 or 21, magnifying the current I.sub.N1 by a factor of M. FET 29, being cascode connected with FET 28, assists in the accurate magnification of the current I.sub.N1. The gate of FET 29 is coupled to a bias voltage generator 30 providing a bias voltage, V.sub.B. Bias voltage generator 30 will be discussed in more detail below, but it sufficient to state here that the voltage V.sub.B is chosen so that the voltage on the drain of FET 28 is substantially the same as the voltage on the drains of FETs 20, 22 to improve the accuracy of the magnification of the current I.sub.N1 by FETs 28, 29. Since I.sub.PTAT /I.sub.N1 =M, then:

The current I.sub.PTAT passes through resistor 26 to generate a voltage equal to the product of the resistance of resistor 26, referred to here as R.sub.26, and the current I.sub.PTAT. This voltage is added to the voltage across the transistor 27, V.sub.BE, discussed above, to develop the bandgap reference voltage V.sub.BG :

Substituting for the current term I.sub.PTAT the above, then:

If R.sub.24 =R.sub.26, then the effects of the resistors 24, 26 can be ignored, making the temperature coefficient of the bandgap voltage V.sub.BG virtually dependent only on the combined multiplied thermal voltage and the forward biased PN junction voltage V.sub.BE. Using exemplary values of M=5, R.sub.24 =R.sub.26, V.sub.BE =0.65 volts and the exemplary values given above, V.sub.BG becomes approximately 1.15 volts with a practically zero temperature coefficient at room temperature. This bandgap voltage reference design has the advantage of allowing another degree of freedom to the designer over the bandgap voltage reference of the prior art (FIG. 2): the ability to set the contribution of the thermal reference to the bandgap output voltage by the current magnification factor M as well as scaling the resistance ratio of the resistors 24, 26. Hence with a one-to-one ratio of the resistances of resistors 24, 26, precision adjustments of the bandgap output voltage V.sub.BG can be easily done by trimming resistor 24 or resistor 26 until the desired output voltage is attained.

To reduce susceptibility of the bandgap voltage V.sub.BG with variations and noise in power supply voltage V.sub.DD, voltage regulator 11 pre-regulates the power supply voltage V.sub.DD to generate V.sub.A, thereby isolating the bandgap voltage reference from V.sub.DD. V.sub.A is chosen such that the voltage on the drains of FETs 20, 22 is substantially the same as the voltage on the drains of FETs 21, 23. An additional advantage of pre-regulating V.sub.A, so that the drain-to-source voltages of FET 20 and FET 21 are substantially the same (also making the drain-to-source voltages of FET 22 and FET 23 substantially the same), is that the currents I.sub.N1 and I.sub.N2 are better matched, improving the desired accuracy of the reference. FETs 31, 32 sense the difference between the voltage on the drains of FETs 20, 22 and that on the drains of FETs 21, 23. Further, as will be explained below, FETs 31, 32 are typically substantially the same size as corresponding FETs 20, 21. FETs 33, 34 mirrors current from FET 31 to combine with the current from FET 32 to node N3. The result is a differential amplifier, with a single-ended output on node N3, responsive to the difference in voltages at the interconnections of the current mirror of FETs 22, 23 and the regulator means of FETs 20, 21. Current source 35 supplies current exceeding the necessary current to power the shown circuits, and other circuitry (not shown) that requires the regulated voltage V.sub.A, with the excess current diverted to V.sub.SS by FET 36. Although not shown here, current source 35 provides current proportional to the current necessary to power the bandgap reference by mirroring and magnifying the current through FET 33. FET 36 responds to the voltage on node N3, the output of the differential amplifier formed by FETs 31, 32, 33, 34. A closed loop is thereby formed which regulates the voltage V.sub.A by FET 36 shunting varying amounts of current from current source 35 to keep the drain-to-source voltages of FETs 22, 23 substantially the same. With exemplary threshold voltages V.sub.th of FETs 20, 21, 31, 32 of 1 volt, the voltage V.sub.A is approximately 3 volts. The regulation of voltage V.sub.A improves the overall performance of the bandgap voltage references by essentially eliminating deviations in the desired output voltage, V.sub.BG, and the desired temperature coefficient thereof, due to inaccuracies in the mirroring and magnification of the currents I.sub.NI, I.sub.N2 into current I.sub.PTAT.

Bias voltage generator 30 provides the proper bias voltage, V.sub.B, to have the voltage on the drain of FET 28 substantially the same as voltage on the drains of FETs 20, 22 (and, consequently, the same voltage as on the drains of FETs 21, 23) to assure accurate magnification of the current I.sub.N1. FETs 38 and 39 are diode-coupled to drop the voltage V.sub.A to the proper bias voltage V.sub.B. By having the sizes of FETs 38, 39 substantially 1/M the sizes of corresponding FETs 28, 29 and with substantially the same current flowing through FETs 38, 39 as in FETs 20, 22, the voltage on the drain of FET 38 will be substantially equal to the voltage on the drains of FETs 20, 21. Consequently, because the size and current flowing through FET 28 is M times that of corresponding FET 39 and the voltage on the sources of FETs 29 and 39 are substantially the same, the gate-to-source voltage of FET 39 must then be the same as the gate-to-source voltage of FET 29, yielding the necessary bias voltage V.sub.B.

To assure that the current flowing through FETs 38, 39 is substantially the same as the current flowing in FET 20, FET 40 mirrors the current flowing in FET 31. As stated above, the size of FET 31 is substantially the same as the size of FET 20. Having the size of FET 40 substantially the same size as FET 33 allows the current flowing in FET 40 to accurately mirror the current flowing in FET 20. Hence, the bias voltage V.sub.B from voltage generator 30 assures that the drain voltage of FET 28 is substantially the same voltage as on the drains of FETs 20, 22.

Although N- and P-channel FETs and PNP bipolar transistors are shown, it is understood that the N- and P-channel FETs can be interchanged and NPN bipolar transistors can be substituted for PNP transistors, with corresponding change in polarity of V.sub.DD and V.sub.SS, with no significant change in the performance of the claimed bandgap voltage reference. Further, it is understood that NPN transistors can be used in place of the shown PNP transistors with the suitable reconfiguration of the transistors. In addition, although conventional current mirrors are shown, it is understood that other types of current mirrors could be substituted, such as Wilson current mirrors. It is also understood that scaling the size of a particular FET can be accomplished by simply enlarging the width of the FET. It is preferable, however, to achieve scaling by paralleling multiple FETs to achieve the desired size.

Having described the preferred embodiment of this invention, it will now be apparent to one of skill in the art that other embodiments incorporating its concept may be used. If is felt, therefore, that this invention should not be limited to the disclosed embodiment, but rather should be limited only by the spirit and scope of the appended claims.