Speed Control System for DC Motor

The present invention relates to a motor control system, and more particularly to a circuit for controlling the rotational speed of a DC motor.

The rotational speed of a DC motor is proportional to the counterelectromotive voltage. Accordingly, the principle for a speed control of a motor is to change a motor driving current in accordance with variation of a motor load and to counteract, by controlling a voltage between terminals of the motor, the variation of a voltage drop across an internal resistance of the motor caused by the change of the motor driving current. Thus, the counterelectromotive voltage of the motor becomes constant, and the rotational speed thereof is stabilized against load variation.

A speed control circuit of a DC motor is based upon the above-mentioned operating principle. However, the control circuit according to prior art has a drawback that the variation of a bias current due to the changes of a power source voltage and/or an operating temperature causes the counterelectromotive voltage of the motor to vary with the result of change in a rotational speed.

Therefore, an object of the present invention is to provide an improved motor speed control circuit.

Another object of the present invention is to provide a control system for stabilizing a speed of a motor against not only variation in a motor load but also variations in a power source voltage and an operating temperature.

A motor speed control system according to the present invention comprises a DC motor connected between a power source terminal and a first terminal, a resistor connected between the power source terminal and a second terminal, a resistance divider connected between the first and second terminals, a third terminal connected to a dividing point of the resistance divider, a first transistor having a collector-emitter path connected between the first terminal and a reference terminal, a second transistor having a collector-emitter path connected between the second terminal and reference terminal, an error amplifier having first and second input terminals and an output terminal which is connected to the bases of the first and second transistors, means coupled between the first input terminal of the error amplifier and the first terminal for generating a reference voltage thereacross, and means for connecting the second input terminal of the error amplifier to the third terminal.

The reference voltage circuit is provided on the side of the first terminal and is thus disconnected from the second terminal. Accordingly, the counterelectromotive voltage of the motor is stabilized against the variation in a bias current flowing into the reference voltage generating means, so that the rotational speed of the motor is kept constant.

The above and other objects, advantages and features of the present invention will be more apparent from the following description taken in conjunction with the accompanying drawings, in which

FIG. 1 is a circuit diagram showing a prior art control system;

FIG. 2 is a circuit diagram showing a first embodiment of the present invention;

FIG. 3 is a circuit diagram showing the circuit of FIG. 2 in detail;

FIG. 4 is a circuit diagram showing another example of the construction of the reference voltage circuit and bias current source shown in FIG. 1;

FIG. 5 is a circuit diagram showing a second embodiment of the present invention;

FIG. 6 is a circuit diagram showing a third embodiment of the present invention; and

FIG. 7 is a circuit diagram showing one application of the present invention.

Before describing the embodiments of the present invention, a prior art motor speed control circuit will be described with reference to FIG. 1 to facilitate the understanding of the present invention.

In FIG. 1, a first transistor 6 for motor driving has its collector-emitter path connected between a first terminal 1 and a reference terminal 4 which is grounded, and a second transistor 7 has its collector-emitter path connected between a second terminal 2 and the reference terminal 4. The second terminal 2 is connected to one input terminal of an error amplifier 9 through a reference voltage circuit 8, and the other input terminal of the error amplifier circuit 9 is connected to a third terminal 3. An output from the error amplifier circuit 9 is applied in common to the bases of the transistors 6 and 7. A DC motor 10 is connected between a power source 15 and the first terminal 1, and a resistor 13 is connected between the power source 15 and second terminal 2. A resistor 11 is connected between the second and third terminals 2 and 3, and a resistor 12 is connected between the first and third terminals 1 and 3. These resistors 11 and 12 constitute a voltage divider. A bias current I.sub.B is applied from an current source 16 to the reference voltage circuit 8 which hence generates a reference voltage thereacross.

In this arrangement, a voltage V.sub.M between terminals of the DC motor 10 is expressed by the following equation (1):

wherein R.sub.M is the internal resistance of the DC motor 10; I.sub.M is a current flowing through the DC motor 10 (i.e., a motor driving current); and E.sub.a is a counterelectromotive voltage of the DC motor 10.

Since the error amplifier circuit 9 operates to control the base voltages of the transistors 6 and 7 so as to equalize the voltages at the two input terminals thereof, the voltage difference between the terminals 2 and 3 becomes equal to the reference voltage V.sub.REF generated by the reference voltage circuit 8. Assuming that the resistance value of the resistor 11 is R.sub.1, a current I.sub.1 flowing through the resistor 11 is expressed by the following equation (2): ##EQU1##

Since an input current of the error amplifier circuit 9 is substantially zero, the current I.sub.1 also flows through the resistor 12. Accordingly, the potential difference V.sub.2-1 between the terminals 2 and 1 is expressed by the following equation (3): ##EQU2## wherein R.sub.2 represents the resistance value of the resistor 12.

The current I.sub.1 further flows through the resistor 13. A collector current I.sub.7 of the transistor 7 and a bias current I.sub.B produced by the current source 16 also flow through the resistor 13. Therefore, the voltage drop V.sub.13 across the resistor 13 is expressed by the following equation (4):

wherein R.sub.3 is the resistance value of the resistor 13.

A collector current I.sub.6 of the transistor 6 becomes equal to the sum of the motor driving current I.sub.M and the current I.sub.1, and is expressed by the following equation (5):

Since the collector current I.sub.7 of the transistor 7 is not supplied to the motor 10, the collector current I.sub.7 is designed to be 1/K (K=positive value) of the collector current I.sub.6 of the transistor 6, as expressed by the following equation (6), for purpose of reducing power consumption.

Conversely, the resistance value R.sub.13 of the resistor 13 is designed to be K times as high as that of the internal resistance R.sub.M of the motor 10, as represented by the following equation (7):

Thus, the following equation (8) is obtained from the above equations (2)-(7). ##EQU3##

Accordingly, as is clear from the results of comparison between the equations (1) and (8), the counterelectromotive voltage Ea of the motor 10 is expressed by the following equation: ##EQU4##

Thus, the counterelectromotive voltage Ea is made constant against the variation in the motor current I.sub.M. Namely, even when a load of the motor 10 varies, the rotation speed thereof is kept constant.

However, a relatively large change occurs in the first member KI.sub.B R.sub.M of the right side of the equation (9) due to the variation in the voltage of the power source 15 and/or an operating temperature of the control system, so that the rotation speed of the motor varies. More specifically, the voltage of the power source 15 varies due to a ripple component when it is obtained by rectifying an AC voltage. When a battery is employed as the power source 15, its voltage decreases in accordance with power consumption. The operating temperature varies dependently upon the environmental conditions and further varies due to the heat generation of the transistors 6 and 7. Such variations of power source voltage and/or operating temperature cause the change in the bias current I.sub.B of the bias current source 16. The change in the bias current I.sub.B can be made small, but it is amplified K times, as is apparent from the equation (9). Since the collector current I.sub.7 of the transistor 7 is not used for driving the motor 10, the value of K is considerably increased to reduce the power consumption. Consequently, a large variation occurs in the counterelectromotive voltage Ea due to the change of the bias current I.sub.B, and the rotation speed of the motor cannot be kept constant.

Referring to FIG. 2, there is shown a first embodiment of the present invention, in which the same constituents as those shown in FIG. 1 are denoted by the same reference numerals to omit their further description. In this embodiment, the reference voltage circuit 8 is connected between the first terminal 1 and the non-inverting input terminal (+) of the error or differential amplifier circuit 9, and a terminal 5 connected to the power source 15 is provided as a fifth terminal to disconnect the bias current source 16 from the second terminal 2. The bias current source 16 is connected between the terminal 5 and the reference voltage circuit 8. Accordingly, the bias current I.sub.B is supplied from the power source 15 without flowing through the resistor 13, and then flows into the transistor 6 through the reference voltage circuit 8. As a result, the equations (2) to (5) shown previously are rewritten to the following equations (10) to (13), respectively. ##EQU5##

From the equations (10) to (13), (6) and (7), a voltage V.sub.M between the terminals of the DC motor 10 is expressed by the following equation (14): ##EQU6## Thus, the counterelectromotive voltage Ea is represented as follows: ##EQU7## This voltage Ea is constant irrespective of the variation of the motor current I.sub.M. Namely, a stabilized rotational speed of the motor against the load variation is obtained. Moreover, the first member on the right side of the equation (15) does not contain the proportional constant K. Accordingly, even when the bias current I.sub.B varies slightly due to the variation of the voltage of the power source 15 and/or operating temperature, such slight variations are not amplified with K times. As a result, the rotation speed of the motor is also stabilized against the variations in the power source voltage and the operating temperature.

FIG. 3 shows a detailed circuit construction of the embodiment of FIG. 2, wherein the transistors 6 and 7, the reference voltage circuit 8, the error amplifier 9 and the bias current source 16 are fabricated as a monolithic semiconductor integrated circuit device 100. In order to obtain the relationship of the equation (6), the transistor 6 is formed of a plurality of unit transistors 6-1 to 6-N, and their collector-emitter paths are coupled in parallel, their bases being connected in common. The size of each of unit transistors is equal to the size of the transistor 7. The transistors 6-1 to 6-N and 7 have emitter resistors 6R-1 to 6R-N and 7R, respectively. The reference voltage circuit 8 consists of three NPN transistors Q.sub.5 to Q.sub.7 and three resistors 8R.sub.1 to 8R.sub.3 which are connected as shown in the drawing. This structure is known as a so-called "band gap regulator", which generates a reference voltage of about 1.2 V thereacross. This reference voltage does not change substantially irrespective of the variation in the voltage of the power source 15 and the operating temperature. The error amplifier 9 has two PNP transistors Q.sub.1 and Q.sub.2 connected in a differential form, the emitters of which are connected to a constant current source 9-1 through the respective emitter resistors 9R.sub.1 and 9R.sub.2. The base of the transistor Q.sub.1 is connected to the terminal 3 as an inverting input terminal (-), and the base of the transistor Q.sub.2 is connected as a non-inverting input terminal (+) to the mode between the bias current source 16 and the reference voltage circuit 8. The collector of the transistor Q.sub.2 is connected to the reference terminal 4 through a diode D.sub.1 and a resistor 9R.sub.3, and further to the base of an NPN transistor Q.sub.3. The emitter of the transistor Q.sub.3 is connected to the reference terminal 4 through a resistor 9R.sub.4, and the collector thereof is connected to a constant current source 9-2 and the base of an NPN transistor Q.sub.4. The collector of the transistor Q.sub.4 is connected to the terminal 5, and the emitter thereof is connected to the reference terminal 4 through a resistor 9R.sub.5. The emitter of the transistor Q.sub.4 is led out as an output terminal of the error amplifier 9 and is connected in common to the bases of the transistors 7 and 6-1 to 6-N.

Another circuit construction of the reference voltage circuit 8 and the bias current source 16 is shown in FIG. 4. In this circuit, the change in the bias current I.sub.B is made small against the variations of the power source voltage and operating temperature. More specifically, the bias current source 16 has a band gap regulator consisting of transistors Q.sub.10 to Q.sub.12 and resistors 16R.sub.1 to 16R.sub.3, and the regulator is connected in series between the terminals 5 and 4 via a diode-connected transistor Q.sub.13 and a constant current source 16-1. A transistor Q.sub.14 and a resistor 16R.sub.4 are further provided as shown in the drawing. Therefore, assuming that the voltage of the band gap regulator is represented by V.sub.B, the bias current I.sub.B takes a value of V.sub.B /R.sub.164 (R.sub.164 being a resistance value of the resistor 16R.sub.4), so that the change of the bias current I.sub.B with respect to the variation in the power source voltage and the operating temperature is reduced to an extremely low level. The reference voltage circuit 8 consists of a single resistor 8R. Accordingly, the reference voltage V.sub.REF takes a value of ##EQU8## (R.sub.8 being a resistance value of the resistor 8R), and it is stabilized against the variation of the power source voltage and the operating temperature. If the resistance values R.sub.8 and R.sub.164 are designed to be R.sub.8 =R.sub.164, a band gap voltage V.sub.B appears across the resistor 8R.

The transistor Q.sub.13 may be omitted. In this case, a diode is provided to be connected in series to the resistor 8R in the reference voltage circuit 8.

FIG. 5 shows a second embodiment of the present invention, in which the same constituents as those shown in FIG. 2 are denoted by the same reference numerals to omit their explanation. In this embodiment, an NPN transistor 17 is additionally provided. The collector-emitter path of this transistor 17 is inserted between the terminal 2 and the collector of the transistor 7, and the base of the transistor 17 is connected to the node between the reference voltage circuit 8 and the bias current source 14. Accordingly, the collector potential of the transistor 7 differs from the collector potential of the transistor 6 by a level that is equal to a voltage difference between the reference voltage V.sub.REF and a base-emitter forward voltage of the transistor 17. In other words, the collector-emitter voltage of the transistor 7 is rendered close to the collector-emitter voltage of the transistor 6, so that the transistor 7 is matched with the transistor 6 to obtain an accurate value of the proportional constant K.

FIG. 6 shows a third embodiment of the present invention. The error amplifier 9 has a substantially identical construction as that shown in FIG. 3, and therefore includes the constant current source 9-1 between the terminal 5 and a node to which the emitters of the transistors Q.sub.1 and Q.sub.2 are connected in common through resistors 9R.sub.1 and 9R.sub.2. In order to expand the input dynamic range of the error amplifier 9, therefore, it is preferable to supply a lowered level of the voltage at the terminal 3 to the inverting input terminal (-) of the error amplifier 9. For this purpose, a transistor 60 and a constant current source 50 are provided as a level-shift circuit. The terminal 3 is connected via the base-emitter path of the transistor 60 to the inverting input terminal (-) of the error amplifier 9. Accordingly, the voltage at the terminal 3 is level-shifted by a base-emitter forward voltage of the transistor 60, and then applied to the inverting input terminal (-) of the error amplifier circuit 9. The constant current source 50 maintains the voltage between the base and emitter of the transistor 60 at a constant level. The bias current source 16 has a band gap regulator consisting of transistors Q.sub.40 to Q.sub.42 and resistors R.sub.40 to R.sub.42 and connected in series between the terminals 4 and 5 via a constant current source 40. The band gap voltage V.sub.B is supplied to a series circuit consisting of the base-emitter path of a transistor Q.sub.43 and a resistor R.sub.43. The transistors Q.sub.44 and Q.sub.45 constitute a current mirror circuit in which the ratio of input current to output current is set to 1, and a collector current of the transistor Q.sub.43 is supplied as an input current for the current mirror circuit. Accordingly, the bias current I.sub.B from the bias current source 16 takes a value of (V.sub.B -V.sub.BE43)/R.sub.43, wherein V.sub.BE43 is the voltage between the base and emitter of the transistor Q.sub.43. The reference voltage circuit 8 is formed of a single resistor R.sub.50, the resistance value of which is set to the same level as that of the resistor R.sub.43. Therefore, a voltage drop (V.sub.B -V.sub.BE43) occurs across the resistor R.sub.50. Since the voltage between the base and emitter of the transistor Q.sub.43 and that between the base and emitter of the transistor Q.sub.60 are substantially equal, the voltage between the terminals 3 and 1 is maintained at the level of the band gap voltage V.sub.B. This voltage is stabilized against the vairations of the power source voltage and operating temperature. Moreover, the change of the bias current I.sub.B become very small.

FIG. 7 shows an application of the present invention, which can change the rotation speed of the motor 10. A control circuit 300 can employ any one of the structures shown in FIGS. 1, 3, 5 and 6. It is understood from the equation (15) that the counterelectromotive voltage Ea of the motor 10, i.e. the rotational speed thereof, can be switched from one value to another value by changing the resistance value of at least one of the resistors 11 and 12. In the circuit shown in FIG. 7, the resistance value of the resistor 12 is changed. To this end, a series circuit of a resistor 201 and the collector-emitter path of a transistor 200 is connected in parallel to the resistor 12. The base of the transistor 200 is connected to the power source 15 through a resistor 202 and a switch 203. When the switch 203 is off, the transistor 200 is in a cut-off state, so that only the resistor 12 is connected between the terminals 1 and 3. When the switch 203 is turned on to energize the transistor 200, the resistor 201 is connected in parallel to the resistor 12, so that the resistance value between the terminals 1 and 3 becomes small. This causes the rotation speed of the motor to become low.

The internal resistance R.sub.M of the motor 10 may become large as the rotational speed of the motor increases. If the resistance value of the resistor 13 as a torque control resistor is set constant nevertheless the internal resistance R.sub.M has become large, the variation of the rotational speed of the motor increases. Therefore, when the rotation speed of the motor is increased by changing the resistance value of at least one of the resistors 12 and 11, it is preferable to increase the resistance value of the resistor 13.

The present invention is not limited to the abovementioned embodiments and application, but may be modified and changed without departing from the spirit and scope of the present invention.