Antenna Feed Network Employing Over-Coupled Branch Line Couplers

1. Field of the Invention

The invention relates generally to antenna feed systems, particularly in the microwave frequency bands.

2. Description of the Prior Art

A wealth of antenna coupling technology is extant in the prior art. The text "An Antenna Engineering Handbook." Henry Jasik. Editor, McGraw-Hill 1961 (first edition) describes much of this prior art. In addition, the text "Radar Handbook", Merrill Skolnick, Editor (McGraw Hill 1970) presents an overview of the state of this art at the time of its publication. That reference also serves to acquaint the reader with microwave system aspects requiring and utilizing circuits of the class to which the invention belongs.

In the prior art, stripline implementations of antenna multi-element feeds have frequently required rather complex trace cross-overs or connections between more than one plane of conductive traces. Single plane cross-overs inherently produce undesired coupling and the cost of manufacture of multi-plane devices is relatively high. The manner in which the invention contributes to this art to provide a simpler, more easily manufactured stripline feed network will be evident as this description proceeds.

Microwave stripline type components for related purposes are described in the patent literature, for example U.S. Pat. No. 4,460,877 entitled "Broad-band Printed Circuit Balun Employing Coupled-Strip All Pass Filters" and copending U.S. Pat. No. 4,578,652 issued Mar. 25, 1986 entitled Broadband Four-Port TEM Mode 180.degree. Printed Circuit Microwave Hybrid. Those disclosures show stripline implemented circuits in which the circuit traces are printed circuits held between parallel ground planes, as is the case for the invention. U.S. Pat. No. 3,518,688, discloses a low loss, high strength honeycomb arrangement for effecting the overall structure mechanically. That technique is applicable to the devices of the aforementioned patents and also to this invention.

The basis for the type of circuit coupling to form more than one beam, as employed in the invention, is examined in a paper included in the Transaction of IEEE, Vol. Ap-16, July 1968 pp. 436-440. The resulting feed concept has been referred to as a Lopez Feed.

In consideration of the disadvantages of the pertinent prior art, it may be said to have been the general object of the invention to provide a stripline antenna feed in which the conductive traces are in a single plane (without cross-overs in the plane) forming over-coupled branch-line couplers. The device is therefore relatively easily and inexpensively manufactured.

The single-plane printed circuit traces sandwiched between spaced ground planes according to the invention are folded in a plurality of trombone-like sections successively coupled by means of a plurality of discretely spaced, quarter-wave conductive cross-over traces forming individual directional couplers. One output of each coupled trace of each of the sections feeds a corresponding array element and another passes most of the power on to successive sections.

The circuit of the invention may be thought of as a plurality of four port directional couplers in stripline form, each directional coupler having an input port and a diagonal port into which most of the input RF power is diverted. Couplings between 10 db and 25 db can be easily provided, while a fourth port is terminated.

The details of a typical stripline implementation of the invention are presented hereinafter.

FIG. 1 is a basic illustration of the unique directional coupler according to the invention;

FIG. 2 is a schematic of the single plane traces on the center plane of a stripline device; according to the invention, for feeding "N" antenna elements of an array from an RF input port in a frequency scanning system;

FIG. 2A is a schematic of the single plane traces configured for a beam normal system;

FIG. 2B is a schematic of the single plane traces for forming two or more beams by feeding "N" antenna elements from two or more RF input ports;

FIG. 3 is a plot relating coupling to a corresponding antenna element as it relates to cross-over admittances; and

FIG. 4 is an isometric, partially sectioned view of a typical assembled stripline feed according to the invention.

Although it is to be understood that the circuit of the invention is entirely reciprocal, it will be described in terms of the transmitting mode.

Although the invention would be broadly applicable to any RF frequency, it is most practical in the microwave region where physical sizes are small. Also, the invention may be thought of as realizable in any TEM mode transmission line medium, it is most practical in stripline form.

Referring now to FIG. 1 showing a cross-over coupling section generally at 10, it will be assumed that port 1 is excited from a source of RF power (not shown) via a common input port. It is desired that most of the input power be coupled to port 3, since this power proceeds down the line to the further sections. An arbitrary portion of the input power is diverted (coupled) to port 2 of each section and from there is diverted to a corresponding antenna element (1 through N) as shown in FIG. 2. port 4 terminated in a characteristic impedance 24 of the lines is also shown in FIG. 2. Thus, the assembly of FIG. 1 comprised of cross-over directional coupler sections 10 arranged as shown in FIG. 2 operates as a transverse electro-magnetic (TEM) power distributor/combiner.

Most of the power at port 1 in FIG. 1 exits at port 3 and an arbitrary lesser amount of power is coupled to port 2. In a similarly labeled conventional branch-line hybrid most of the power would be coupled to port 2 from port 1 and a minimum of (-10 db) would appear at port 3.

Three one-quarter wave conductive cross-over traces 11, 12 and 13 are shown in FIG. 1, although the invention is not limited to that number of conductive cross-over traces. In fact, the use of four or more cross-over traces in each directional coupler section broadens the bandwidth of the circuit with little increase in complexity.

In FIG. 2 identified by the general reference character 10a, the three cross-over traces are identified as "a" (the two outside traces) and "c" (the center trace). If the input admittance at port 1 is arbitrarily identified as Y=1, and the admittances of the through lines are Yo=1, then relative or normalized admittances for the three conductive cross-over traces 11, 12 and 13 can be determined and the plot on FIG. 3 relates those admittances to the coupling from port 1 to port 2, i.e., the portion of the input power diverted to an antenna (radiator) element (1 through N). Table I following presents the data of FIG. 3 for particular couplings of the device ideal for use as an antenna feed for a frequency scanning radar system. Note that FIG. 2 is the frequency scanning configuration. The phase delay between adjacent elements is determined as a function of the electrical length from a point 15 and a point 14 and the corresponding points for other sections. The length of the conductive trace leading from port 4 of the directional coupler to the termination characteristic impedance 24 is arbitrary. The parallel spacing d of the conductive traces connecting port 3 of one directional coupler to port 1 of the following coupler should be at least "3w" for adequate decoupling (where "w" is the trace width).

However, FIG. 2A is the beam normal configuration wherein a plurality of meander line lengths 30, 31, 32, 33 produce a constant electrical phase at the antenna (radiator) elements (1-N) independent of frequency. FIG. 2A represents the beam normal configuration because the electrical path lengths from the RF input to each antenna (radiator) element (1-N) is constant. Note that FIG. 2A has a single RF input with each directional coupler terminating in the characteristic impedance 24. Each of the parallel sections of transmission lines are separated by the three one-quarter wave conductive cross-over traces 11, 12 and 13 with the output port (port 2) leading to the corresponding antenna (radiator) element (1-N) via one of the meander lines 30-33 employed for electrical phase matching.

The characteristic admittance of the circuit Y.sub.o is determined by well-known parameters including the trace width and the ground plane spacing as

where

YO=characteristic admittance,

f is a determinable function,

w is the trace width,

b is the ground plane spacing, and

.epsilon. is the dielectric constant of materials between the ground planes.

It therefore follows that the directional coupler branches "a" and "c" admittances can be established by predetermination of their trace widths accordingly for a give "b" and ".epsilon.."

It can be shown that the admittances of branches "a" and "c" do not affect the phase relationships between ports 2 and 3 of the directional coupler 10. It can also be shown that the port 2 signal amplitude (power) decreases as the excitation frequency varies from center frequency.

Another set of solutions is possible for the directional coupler parameters achieving the same power distribution but providing increasing signal amplitude at port 2 as the frequency varies from design center. The "a" and "c" branch normalized admittances for mid-range coupling values are according to Table II for this variation.

In the variation according to Table II, the phase of the port 2 signal lags that of the port 3 signal whereas in the Table I solution the phase of the port 3 signal lags that of the port 2 signal. This will be evident from FIG. 3.

In a further variation (see FIG. 2B) in which elements common to those of the preferred embodiment of FIGS. 2 and 2A are identified by a prime designation, two nearly independent beams can be realized by connecting the normally loaded isolated ports (ports 4) to a second beam-forming network. The two beam forming networks have at least two RF inputs employed as common ports (ports 1) receiving beam 1 and beam 2. Further, the plurality of output ports (ports 2) are connected discretely to the corresponding antenna (radiator) element (1-N) via the meander lines (30'-33') employed again for electrical phase matching. The plurality of four port directional coupler circuits are again implemented in transverse electro-magnetic mode each including parallel sections of transmission line with the first transmission line section being connected at one end to the RF source of excitation located at one of the common ports (ports 1). Also the first transmission line section is connectable to a second directional coupler port of a first directional coupler of the second beam forming network. The second beam forming network is then connected to a characteristic impedance termination point at the other end of the first transmission line section of the second beam forming network providing the fourth directional coupler port. Further, the three one-quarter wave conductive cross-over traces 11', 12' and 13' are positioned as to cross-over the parallel transmission line sections. The construction is as in FIGS. 2 and 2A in which the second directional coupler port is located along the first section of transmission line and the third directional coupler port is located along the second section of transmission line with the third port being diagonal to the first port. The output ports (ports 2) of the first beam forming network connect to the corresponding antenna (radiator) element (1-N) while the second port of the second beam forming network provides an output for connection to the fourth port of the first beam forming network. Finally the third port of each of the directional coupler circuits is connected to the first port of the succeeding directional coupler in each of the beam forming networks to provide RF excitation to each successive directional coupler while the first port of the first of the directional coupler circuits of each of the beam forming networks is connected to one of the common ports. The antenna element spacing is nearly arbitrary, .lambda./2 for example, but can be otherwise. The feed geometry can be such that radiating elements (1 through N) are connected directly at the stripline edge as suggested in FIG. 2, however, the outputs could be connected to coaxial cable or the like for antenna element connection if this is mechanically desirable.

Referring now to FIG. 4, the device of the invention is depicted with a pair of parallel conductive planes 16 and 17 each spaced "b" from the conductive cross-over traces 11, 12 and 13 on a dielectric sheet 18. As previously indicated, the honeycomb dielectric spacers of U.S. Pat. No. 3,518,688 (Stayboldt) can be provided in the spaces between 16 and 18 and between 17 and 18 as that patent reference shows. The conductive cross-over traces 11, 12 and 13 are applied to dielectric board 18 by known printed circuit methods. The conductive cross-over traces 11, 12, and 13 themselves are usually copper or one of its alloys.

To show how a coaxial input line 19 can be electrically connected to the device 10, a coaxial center conductor 22 is connected to the first port 1 in the coupler sections of FIG. 2. An outer conductive sleeve of the coaxial input line 19 would be connected to a plurality of ground planes as at 20 and 21. Of course a coaxial connector could be installed consistent with those connections if desired.

With an understanding of the invention, various modifications within the scope and spirit of the invention will suggest themselves to those of skill in this art. Accordingly, it is not intended that the scope of the invention should be regarded as limited by the drawings or the specifics of this description, these being intended to by typical and illustrative only.