Precommutated Current Source Inverter

The invention relates to current source inverters (CSI) in general, and more particularly to adjustable speed AC motor drives embodying the same. A current source inverter (CSI), as the name implies, is one where the variable frequency electrical output is a current wave rather than a voltage wave.

A CSI includes a large inductor on the DC-side and a three phase phase-controlled-rectifier (PCR) connected to the AC power line. The rectifier generates variable DC voltage which is converted to a variable current source by a large DC-link inductor. The thyristors of the inverter pass the current from the current source symmetrically to the three phases of the machine while generating a variable-frequency, six-stepped current wave in the CSI. The CSI has many advantages over the voltage-source inverter (VSI): fuseless protection, full four-quadrant regenerative capability with simple structure, rugged and reliable operation, etc. In spite of such advantages, the conventional CSI also has several limitations: lower operating frequency range, bulkiness, cost, high spike voltage at the machine terminals, etc. The shortcomings of the conventional CSI mainly arise from the existence of the leakage inductance of the machine. Energy exchange between the leakage inductance and the commutation circuit is necessary in order to obtain the change from phase to phase of the current of the machine at every instant of commutation. To accomplish this various methods have been used which, however, have the aforementioned limitations typical of the conventional CSI.

For instance, a well-known and widely used CSI is the auto-sequential commutated inverter (ASCI), the schematic of which is shown in FIG. 1. Six thyristors, six diodes and six capacitors are used in the ASCI. The capacitors are used both to commutate thyristors and to take over the energy stored in the leakage inductance of the machine. The diodes play a role in isolating the capacitors from the load and in helping to store energy for commutation.

In the ASCI, the operating range of the frequency and the spike voltages are a function of the load, the capacitance value and the machine leakage inductance. Since the capacitors are discharged and charged by the DC-current flowing from the current source during the time of commutation, the upper limit of the frequency of operation is limited by the light load condition and when the DC current is minimum. On the other hand, the capacitor receives its maximum voltage when at maximum load condition, when the DC current is maximum. In order to increase the operating frequency range, the capacitance value must be decreased and made as small as possible. However, the capacitance value should be increased and made as large as possible in order to decrease the peak capacitor voltage. The compromise resides in a choice of a capacitance value which makes the operating frequency somewhat lower and the voltage ratings of the thyristors and the diodes somewhat higher.

In order to cope with such problems, several modifications have been made in the circuitry. FIG. 3 hereinafter shows such a circuit with a diode, a resistor and an inductor connected in parallel with the main thyristor, whereby the capacitor discharging time is reduced and the operating frequency range is extended by creating with the inductance and the resistor an additional discharging path during the commutation interval. However, the efficiency of the inverter is reduced by the dissipation in the resistor of part of the commutation energy to prevent the capacitor voltage from building up too high as the commutating operation repeats itself.

In order to increase the inverter efficiency as well as to widen the operating range, clamp circuits have been used, as is shown in FIG. 4 hereinafter. In an ASCI with a clamp, the capacitors are used only for the commutation and the peak capacitor voltage is limited by the clamp circuit which absorbs the energy stored in the machine leakage inductance during the commutation interval. This type of circuit operates in a wide range of frequency and within a limited capacitor voltage. However, the clamp circuit tends to be bulky and expensive if a technique is used by which the energy is reconverted to in such a form that it can be passed back into the power line.

A new CSI is proposed having high inverter efficiency, wide range of operating frequency and low spike voltages (with greatly reduced total capacitance value) upon the semiconductor devices. There is also no need for a voltage clamp circuit. The invention resides in using pre-commutation in an ASCI system whereby, instead of merely discharging the capacitor energy, this energy is kept in reserve for the subsequent commutation cycle. The system according to the invention uses only one commutation capacitor which is used to commutate the main thyristors and to exchange energy between the leakage inductance of the machine and the commutation circuit. To that effect, positive and negative polarity commutation thyristors and six auxiliary thyristors are used to distribute the current to the respective AC phases in relation to the main thyristors. Moreover, the polarity of the capacitor voltage is reversed through an inductor by controlling one of two inversion thyristors respectively affected to the opposite polarities.

FIG. 1 shows an auto-sequential commutated inverter (ASCI) of the prior art;

FIG. 2 illustrates with curves the operation of the circuit of FIG. 1;

FIG. 3 is a prior art ASCI including ring around reactors and resistors;

FIG. 4 is a prior art ASCI including a clamp circuit;

FIG. 5 shows the basic pre-commutated current source inverter according to the present invention;

FIGS. 6A, 6B illustrate with curves the commutation cycle in the current source inverter of FIG. 5;

FIGS. 7A-7H show the current paths in the current source inverter of FIG. 5 for the successive time intervals A-G illustrated in FIGS. 6A and 6B; and

FIGS. 8 and 9 are two variants of the current source inverter of FIG. 5.

Referring to FIG. 1, a conventional auto-sequential commutated inverter is shown to include a diode (D.sub.1 -D.sub.6) in series with each thyristor (T.sub.1 -T.sub.6) for each pole (T.sub.1, T.sub.4), (T.sub.3, T.sub.6), (T.sub.5, T.sub.2) of a thyristor bridge in relation to respective nodal points J.sub.U, J.sub.V, J.sub.W with respective AC input lines U, V, W to a motor MT. In addition, commutating capacitors are interconnected for each polarity between the respective nodal points of the series networks formed by a thyristor and a diode. Thus, capacitor C.sub.15 is between (T.sub.1, D.sub.1) and (T.sub.5, D.sub.5), capacitor C.sub.13 between (T.sub.1, D.sub.1) and (T.sub.3, D.sub.3) and capacitor C.sub.35 between (T.sub.3, D.sub.3) and (T.sub.5, D.sub.5), and so on for the negative polarity. T.sub.A and T.sub.B are the positive and negative terminals.

Referring to FIG. 2, assuming conduction is between T.sub.1 and T.sub.2, thus along phases U and W while carrying a current I.sub.L, when thyristor T.sub.3 is being gated ON, capacitor C.sub.13 which was initially charged as shown in FIG. 1, will (1) turn OFF thyristor T.sub.1, and (2) start discharging from charge -V.sub.c following trajectory abcd, d corresponding to the new charge +V.sub.c of the capacitor after T.sub.3 is ON and T.sub.1 is OFF. The commutation is completed and the current path is through phase lines V and W at this final stage. When discharging, the capacitor reaches an operative point c (beyond the zero charge level marked by operative point b) owing to the motor voltage E.sub.o between phase lines. Then, from c to d takes place a resonant charging in the opposite polarity due to the effect of the line inductances L.sub.U, L.sub.V in conjunction with the capacitor C.sub.13. Diode D.sub.1 prevents the resonant portion (dotted line) from extending downward beyond point d. Thus, capacitor C.sub.13 has been first working as a commutation capacitor for the off-going thyristor T.sub.1, and it has absorbed energy taken over from the energy stored in the leakage inductance of the machine. The diodes (D.sub.1 -D.sub.6) play the role of isolating the capacitors from the load and they help store energy for commutation. The commutation period is defined by total time (t.sub.1 +t.sub.2), where t.sub.1 is the unloading time, and t.sub.2 the resonant charging time.

Referring to FIG. 3, a prior art bridge of six thyristors T.sub.1 -T.sub.6 connected between DC terminals T.sub.A, T.sub.B and the phase AC output lines U, V, W, is shown to include diodes D.sub.1 -D.sub.6 in series between the respective thyristors, the junction points J.sub.1 -J.sub.6 being common with the thyristors and J.sub.U, J.sub.V, J.sub.W with the output lines. Like in FIG. 1, commutation capacitors C join poles on each polarity side. In order to reduce the capacitor discharging time, and extend the frequency range of operation, additional discharging paths are provided with series network of inductance and resistor (L.sub.1, R.sub.1) with a connecting diode as shown between each of junction points (J.sub.1 -J.sub.6) and the associated DC terminal.

Referring to FIG. 4, another prior art bridge, much like FIG. 3, in that main thyristors (T.sub.1 -T.sub.6), diodes (D.sub.1 -D.sub.6) and commutation capacitors are used between poles, however, with the provision of a clamp circuit CLC connected with junction points J.sub.U, J.sub.V, J.sub.W. Due to the clamp circuit, the energy stored in the machine leakage inductance during the commutation interval is absorbed, thereby allowing wider range of operation and less capacitor voltage.

The present invention resides in making use of a redundancy occurring in the commutation process just described.

It appears from FIG. 2 that the major portion of the time interval t.sub.1 is in effect the discharging period of the capacitor energy stored during the previous commutation cycle and that it can be much longer than that required for recovering the thyristor. Interval t.sub.2 is the resonant charging period of the capacitor in relation to the machine leakage inductance when the phase-to-phase current change of the machine occurs. If the energy stored during the previous cycle can be put to use instead of being wasted, for instance by simply discharging it to supply current to the next phase of the machine which the current will flow through right after termination of the current flowing in the present phase, the capacitors will have been used more efficiently. An inverter operated in this manner is said to be a precommutated current source inverter (PCSI), since such operation occurs just prior to actual commutation of the main thyristor. Inverters so designed, namely using the concept of precommutation, have a number of advantages over the conventional CSI's.

A new precommutated current source inverter (PCSI) is proposed as illustrated in FIG. 5. Curves illustrating the commutation cycle are shown in FIG. 6. A single commutation capacitor C is used both to commutate the main thyristors and to exchange energy between the leakage inductance of the machine and the commutation circuit. Six auxiliary thyristors (A.sub.1 .about.A.sub.6), connected in a bridge between terminals ATA, ATB, are provided and connected by respective nodal points J.sub.1, J.sub.2, J.sub.3 to the three phases U, V, W of the machine, and connected to the main thyristors (T.sub.1 .about.T.sub.6) via the nodal points J.sub.U, J.sub.V, J.sub.W. The current flowing through the commutation capacitor (C) can flow through one of the three phases of the machine depending upon the control of the auxiliary thyristors. Two commutating thyristors (S.sub.1, S.sub.2) serially connected via a nodal point J.sub.S between DC terminals (T.sub.A, T.sub.B) are being connected to the terminals ATA, ATB of the auxiliary thyristor bridge through the commutating capacitor (C) and a nodal point J.sub.S '. Thus, the commutating capacitor C between the nodal point J.sub.S of S.sub.1 and S.sub.2 and the nodal point J.sub.S ' belongs to terminal ATA associated to auxiliary thyristors A.sub.1, A.sub.3, A.sub.5 for one polarity and to terminal ATB associated to auxiliary thyristors A.sub.4, A.sub.6, A.sub.2 for the other polarity, like those of the main thyristors T.sub.1 -T.sub.6. The upper side main thyristors (T.sub.1, T.sub.3, T.sub.5) are commutated by turning ON the upper commutating thyristor (S.sub.1) and, inversely, the lower side main thyristors (T.sub.2, T.sub.4, T.sub.6) are commutated by turning ON the lower side commutating thyristor (S.sub.2). Two serially-connected inversion thyristors (S.sub.3, S.sub.4) are connected in antiparallel to the (A.sub.1 -A.sub.6) auxiliary bridge, thus, across terminals ATA, ATB. The nodal point J.sub.L ' between S.sub.3 and S.sub.4 is connected to the nodal point J.sub.S of commutation thyristors S.sub.1, S.sub.2, through an inductor (L). Accordingly, S.sub.3 or S.sub.4 is connecting one end of the commutation capacitor C to one end of the inductor L, and the polarity of the capacitor voltage is being inverted through the inductor L by turning ON either the upper, or the lower, commutation thyristor (S.sub.4, S.sub.3).

Referring to FIGS. 6A and 6B, the operation of the circuit of FIG. 5 is as follows:

(i) During a precommutation stage, one of the auxiliary thyristors (A.sub.1 .about.A.sub.6) is turned ON together with one of the commutating thyristors (S.sub.1, S.sub.2) in order to create a current path to the next phase of the machine prior to the commutation of the main thyristors.

(ii) After such a period of precommutation and before the polarity of the capacitor voltage has changed, the other auxiliary thyristor (A.sub.1 -A.sub.6) which is in parallel with the main thyristor to be turned off is turned ON to commutate one of the main thyristors. The previously conducting main thyristor is now turned OFF by the remaining energy of the capacitor. This results in efficient commutation, considering the energy required here for commutation.

(iii) After a period which is sufficient to turn OFF the main thyristor, one of the inversion thyristors (S.sub.3, S.sub.4) is turned ON to cause a fast polarity reversal of the remaining capacitor voltage. This operation is especially important during no load operation when the source current is small. Owing to this, the circuit can operate over a wide range of frequency independently from load conditions.

(iv) After the polarity inversion of the capacitor voltage has been completed, the capacitor is resonantly charged until the current flowing through the main thyristor which is not being turned ON is equal to the source current.

Referring to FIGS. 6A and 6B and the time intervals (A-G) there shown, the operation of a commutation cycle will now be explained with more details:

For simplicity, it is assumed that the DC-link inductance is large enough to maintain constant current during the commutation cycle. The DC-source is assumed as in FIG. 5, to be composed of a phase controlled rectifier PCR so that with the DC-link inductors L.sub.A, L.sub.B it can be considered as a current source. The modes of operation during a commutation cycle are explained by the illustration of FIGS. 7A-7H showing the current paths when transferring conduction successively from one another through thyristors T.sub.1, T.sub.2, T.sub.3.

Thyristors T.sub.1 and T.sub.2 are conducting and the DC-source current flows through the phases U and W of the machine. The commutating capacitor C is initially charged, in the direction shown by FIG. 7A, to voltage -V.sub.o.

This is the precommutation interval. At time t.sub.o thyristor S.sub.1 and the auxiliary thyristor A.sub.3 are turned ON simultaneously. The phase V current i.sub.v gradually increases, while the phase U current i.sub.u gradually decreases during interval B (FIG. 6B). The time duration t.sub.b of this interval is controlled properly in relation to the load current to discharge the energy to a certain level V.sub.c1 (FIG. 6A) but not to discharge it too much, in order to retain enough energy to turn OFF the main thyristor T.sub.1 at the next interval (C). Interval B never appears in a conventional CSI. It is a characteristic feature of the precommutated current source inverter according to the invention.

At time t.sub.1 thyristor A.sub.1 is now turned ON, which turns OFF thyristor T.sub.1 due to the reverse bias voltage of the capacitor remaining from interval B. The DC-source current flows through the capacitor and through phases U and V. The duration of this interval C depends on the turn-OFF time requirement of the main thyristor T.sub.1. The voltage of capacitor C goes from V.sub.c1 to V.sub.c2 (FIG. 6A), while currents i.sub.u and i.sub.v (FIG. 6B) remain nearly constant.

The voltage polarity of Capacitor C did not change during interval C. If the DC-source current is small (light load condition), the magnitude of the capacitor voltage remains large and negative after the termination of interval C. By turning ON thyristor S.sub.3 at time t.sub.2, the capacitor voltage is now caused to rapidly reverse through the inductor L. This eliminates unnecessary charging time of the capacitor. Interval D enables operation of the inverter over a wide range of frequency, independently of the load condition.

After interval D, depending upon whether the capacitor voltage reaches or exceeds the emf voltage (V.sub.emf) between phases U and V of the machine or it does not, there will be either a single time interval E.sub.1, or an additional time interval E.sub.2 will be provided. The latter is skipped altogether if the capacitor voltage has reached V.sub.emf, when interval D expires (time t.sub.3 in FIG. 6B).

This interval starts at time t.sub.3 if the capacitor voltage exceeds the emf voltage V.sub.emf existing between the U and V phases of the machine, with current flowing through L. During interval E.sub.1, thyristor T.sub.3 is turned ON, thereby turning OFF A.sub.3. The DC-source current is bisected into two current paths (as shown in FIG. 7E); one path is through thyristor S.sub.1, capacitor C, thyristor A.sub.1 and the U-phase of the machine. The other path is through thyristor T.sub.3 and the V-phase of the machine.

This interval starts when the inductor current i.sub.L (FIG. 6A) becomes zero, and the capacitor voltage V.sub.c is still less than the emf voltage V.sub.emf between U and V phases of the machine. (Otherwise interval E.sub.2 is skipped.) The thyristor states and the current path for interval E.sub.2 are equal to those of interval C (FIG. 7C).

When the inductor current i.sub.L becomes zero and the capacitor voltage V.sub.c is greater than the emf voltage V.sub.emf between the U and V phases, interval F is initiated at time t.sub.4. During this interval, the U phase current i.sub.u gradually decreases to zero while the V phase current i.sub.v gradually increases and becomes equal to the DC-source current (FIG. 6B). Capacitor C is resonantly charged. The maximum peak voltage of the capacitor at the end of interval F occurs under maximum load condition and it is approximately proportional to the initial current value of the U-phase at the beginning of interval F. As shown in FIG. 6B, the initial (time t.sub.4) phase currents of interval F are approximately equal to the final currents of interval B (time t.sub.1) which are considerably less than the DC-source current due to the precommutation effect. This is the main reason why capacitor voltage V.sub.c is lower, in the current source according to the invention, as compared with the conventional ASCI circuit. When capacitor voltage V.sub.c is lower, the semiconductor voltage ratings in the inverter are also lower by the same amount. This results in lower cost.

At time t.sub.5 (FIG. 6B) thyristors T.sub.2 and T.sub.3 are turned ON and all other thyristors are OFF. This is the final state at the end of the commutation cycle.

The preceding sequence of operation described a commutation between thyristors T.sub.1 and T.sub.3. Similar sequences of events are required to commutate between T.sub.2 and T.sub.4, T.sub.3 and T.sub.5, T.sub.4 and T.sub.6, and T.sub.5 and T.sub.1 before the above-described commutation sequence is repeated.

There are other implementations of the invention.

For instance, FIG. 8 shows a modification of the circuit of FIG. 5 which eliminates the large dv/db build-up across the auxiliary thyristors, as illustrated upon thyristor A1 by the mode associated with time interval B in FIGS. 6B and 7B. To this effect, diodes D.sub.1 to D.sub.6 are serially connected with the auxiliary thyristors A.sub.1 -A.sub.6, respectively, and serial networks are provided including a diode and a resistor, namely (D.sub.11, R.sub.11) to (D.sub.16, R.sub.16) which are connected in antiparallel fashion via junction point J.sub.AP with the auxiliary thyristor bridge and the commutating capacitor C, respectively, to the respective nodal points between (A.sub.1, D.sub.1), (A.sub.2, D.sub.2) . . . (A.sub.6, D.sub.6).

Another variant is shown in FIG. 9. There, two capacitors C.sub.1 and C.sub.2 are used, instead of one commutating capacitor C and of the inductor L. Capacitor C.sub.1 is associated with commutating thyristors S1, S2, and capacitor C.sub.2 is associated with commutating thyristors S1', S2', in the same fashion as were C and L in FIG. 5. However, diodes D.sub.1, D.sub.2, associated with C.sub.1, and D.sub.3, D.sub.4 associated with C.sub.2, are mounted in antiparallel fashion with the auxiliary thyristor bridge (A.sub.1 -A.sub.6). With this configuration, a total capacitance value can be selected by control in three levels: C.sub.1, C.sub.2 and C.sub.1 +C.sub.2. Capacitor C.sub.1 represents the lower capacitance value; capacitor C.sub.2 has a value larger than C.sub.1, and the maximum capacitance is effective when both are used, namely (C.sub.1 +C.sub.2). These values are used, C.sub.1 for the lowest load condition, C.sub.2 for the middle load condition, and (C.sub.1 + C.sub.2) for maximum load. Such capacitance value adjustment is effected by controlling the commutation thyristors (S1, S2, S1', S2') individually, or concurrently, in relation to the load current (S1, S2) for the value C.sub.1 ; (S1', S2') for the value C.sub.2 and both (S1, S1', S.sub.2, S.sub.2 ') for the maximum value (C.sub.1 +C.sub.2). This results in a wide operative frequency range like with the capacitor voltage inversion approach used in the embodiment of FIG. 5.

It is observed that the effect of the operation of the auxiliary thyristors is to reduce the peak capacitor voltage, and hence the thyristor's voltage, which actually becomes a little more than one-half of what it could be otherwise. Moreover, no voltage clamp circuit is necessary, which would have been bulky and expensive. The reason that the voltage stress is reduced lies in that a larger than "normal" capacitor can be used, thanks to the precommutation process, since the larger the capacitor the more the voltage stress is lowered.