Stepper Motor Control Circuit

This invention relates to a stepper motor control circuit.

In known stepper motor control circuits, control of the current by means of a chopper operating at a frequency an order of magnitude higher than the maximum stepping rate has already been proposed. The circuits employed in the prior art have, however, been extremely complex and have usually employed output stages containing a large number of high power semi-conductor components.

In addition, although the problem of shock loading of the rotor by the stepping pulses to it has been considered previously, no completely satisfactory solution has been proposed.

In accordance with one aspect of the invention a stepper motor control circuit comprises a waveform generator circuit arranged to generate a pair of substantially stepless symmetrical periodic waveforms of constant amplitude and of frequency determined by a variable step rate demand signal received thereby, the waveforms being 90.degree. out of phase, a pair of pulse width modulation circuits arranged to drive the motor windings, each modulation circuit receiving one waveform as a current demand signal and a feedback signal related to the instantaneous current in the associated one of the motor windings.

The term "substantially stepless" is used herein to mean a waveform in which any step changes which occur are of very small magnitude compared with the amplitude of the waveform. Thus, for example, the waveform could be generated utilizing digital techniques, wherein the waveform amplitude is represented by a digital signal of at least 7 bits, so that a change in the digital signal represents a step in the wave form of only one one hundred and twenty eight of the waveform amplitude.

Preferably, each modulation circuit includes an output stage comprising a pair of semi-conductor switch elements which are transformer coupled to a driving stage of the modulation circuit. The output stage is preferably arranged as a push-pull stage operating between positive and negative power supply rails with the windings connected to an earth return through a current sensing resistor which provides the current feedback signal.

Preferably, the pulse width modulating circuits are arranged to operate at a fixed frequency at least an order of magnitude higher than the maximum stepping frequency.

Four examples of the invention are shown in

FIGS. 1 to 4 respectively of the accompanying drawings which are electrical circuit diagrams of the stepper motor control circuits.

Referring firstly to FIG. 1, the stepper motor which is being controlled has four windings, two of which, 10 and 11 are shown in the accompanying drawing. These windings are connected in parallel with one another, via a current sensing resistor R.sub.1 to ground at one end and to the junction of two inductors L.sub.1, L.sub.2 at the other end. The inductors L.sub.1, L.sub.2 are connected in series between the collector of a pnp transistor Q.sub.1 and the collector of an npn transistor Q.sub.2. Transistor Q.sub.1 has its emitter connected to a positive power supply rail 12 and transistor Q.sub.2 has its emitter connected to a negative power supply rail 13. The inductor L.sub.1 is magnetically linked to a winding L.sub.3 which is connected between ground and the anode of a diode D.sub.1, the cathode of which is connected to rail 12. Similarly inductor L.sub.2 is linked to a winding L.sub.4 connected between ground and the anode of a diode D.sub.2 which has its cathode connected to the rail 13. A capacitor K.sub.3 is connected between the junction of inductors L.sub.1 and L.sub.2 and ground.

Transistor Q.sub.1 is biased off by a resistor R.sub.3 connecting its base to the rail 12. The base of transistor Q.sub.1 is also connected to the anodes of two diodes D.sub.3 and D.sub.4 the cathodes of which are connected to opposite ends of two series connected secondary windings TRA.sub.1 and TRA.sub.2, the common point of which is connected via a resistor R.sub.4 to the rail 12. Similarly, transistor Q.sub.2 is biased off by a resistor R.sub.5 connected between rail 13 and the base of transistor Q.sub.2 which is also connected to the cathodes of two diodes D.sub.5 and D.sub.6 having their anodes connected to opposite ends of two series secondary windings TRB.sub.1 and TRB.sub.2, the common point of which is connected by a resistor R.sub.6 to the rail 13.

The transformer of which the windings TRA.sub.1 and TRA.sub.2 are the secondaries has two primary windings TRA3 and TRA4 which are both connected at one end to the collector of a pnp transistor Q.sub.3 the emitter of which is connected to a +12 V regulated supply rail 14. The other ends of windings TRA3 and TRA4 are connected to the anodes of two diodes D.sub.7 and D.sub.8 which have their cathodes connected respectively to the C.sub.2 and C.sub.1 output terminals of a switchmode pulse width modulation control integrated circuit 15 (type TL494MJ manufactured by Motorola Inc). A resistor R.sub.7 interconnects the anodes of the diodes D.sub.7 and D.sub.8.

Similarly two primary windings TRB3 and TRB4 associated with windings TRB1 and TRB2 are connected at one end to the collector of a pnp transistor Q.sub.4, the emitter of which is connected to the rail 14. Two diodes D.sub.9 and D.sub.10 connect the other ends of the windings TRB.sub.3 and TRB.sub.4 to the C.sub.2 and C.sub.1 terminals of i.c. 15. A resistor R.sub.8 interconnects the anodes of diodes D.sub.9 and D.sub.10.

The transistors Q.sub.3 and Q.sub.4 are controlled by a signal at a terminal A so that when this signal is high transistor Q.sub.3 is off and transistor Q.sub.4 is on and when the signal is low transistor Q.sub.3 is on and transistor Q.sub.4 is off. To this end transistor Q.sub.3 has its base connected by a resistor R.sub.9 to the rail 14 and by a resistor R.sub.10 to the terminal A. The transistor Q.sub.4 has its base connected by a resistor R.sub.11 to the rail 14 and by a resistor R.sub.12 to the collector of an npn transistor Q.sub.5 the emitter of which is connected to a ground rail 16 and the base of which is connected by a resistor R.sub.13 to the terminal A.

The integrated circuit 15 has its C.L. sense inverting input terminal connected by a resistor R.sub.14 to a terminal B and by a resistor R.sub.15 to the rail 16. The error amplifier non-inverting input terminal is connected by a resistor R.sub.16 to terminal B and by a resistor R.sub.17 to rail 16. The error amplifier inverting input terminal and the C.L. sense amplifier non-inverting input terminal are both connected to the junction between the current sensing resistor R.sub.1 and the windings 10 and 11. The compensation/PWM Comparator terminal is left open circuit and is not used. The dead time control terminal, the ground terminal and the E.sub.1 and E.sub.2 terminals are all connected to rail 16. A resistor R.sub.18 connects the RT terminal to rail 16 and a capacitor K.sub.1 connects the C.sub.T terminal to ground. These two components control the frequency of a clock oscillator incorporated in i.c. 15. The V.sub.REF and output control terminals of the i.c. 15 are connected together.

The error amplifier non-inverting input terminal of the i.c. 15 is also connected to the cathode of a diode D.sub.11 and C.L. sense amplifier inverting input terminal is connected to the anode of a diode D.sub.12. The anode of diode D.sub.11 and the cathode of diode D.sub.12 are connected together and also connected via a resistor 19 to the terminal A.

The terminals A and B are the output terminals of a waveform generation circuit which occupies the lower half of the drawing.

This waveform generation circuit receives an 8-bit digital word as an input, such word being derived in parallel form from a control computer (not shown), and representing the desired stepping frequency of the stepper motor. The respective bit inputs are connected by a simple d-to-a conversion network 8 to the inverting input of an operational amplifier A.sub.1 which has a feedback resistor R.sub.20 connected between its output and its inverting input. The non-inverting input of amplifier A.sub.1 is connected to the reference voltage output of the i.c. 15.

The inverting input of an amplifier A.sub.2 is connected by a resistor R.sub.21 to the output of amplifier A.sub.1 and by a resistor R.sub.22 to the output of amplifier A.sub.2. The non-inverting input of amplifier A.sub.2 is grounded via a resistor R.sub.23. Resistors R.sub.21 and R.sub.22 are of equal value so that amplifier A.sub.2 acts as a unity gain inverting amplifier providing an output signal of the same magnitude as the output of amplifier A.sub.1, but of opposite sign.

Two pairs of complementary f.e.t.s. Q.sub.6 and Q.sub.7, Q.sub.8 and Q.sub.9 are each connected in series between the outputs of amplifiers A.sub.1 and A.sub.2 the gates of f.e.t.s. Q.sub.6 and Q.sub.7 being connected together by two resistors R.sub.24 and R.sub.25 in series and the gates of f.e.t.s. Q.sub.8 and Q.sub.9 being connected together by two resistors R.sub.26 and R.sub.27. The voltage at the junctions of each of these pair of resistors determines which one of each pair of f.e.t.s. is conductive.

The junction between transistors Q.sub.8 and Q.sub.9 is connected via a resistor R.sub.28 to the inverting input of an operational amplifier A.sub.3 which has its non-inverting input grounded by a resistor R.sub.29 and its output connected by a feedback capacitor K.sub.2 to its inverting input. The amplifier A.sub.3 operates as an integrator, the output of which ramps linearly at a rate determined by the output voltage of amplifiers A.sub.1 and in a sense determined by which of the two f.e.t.s. Q.sub.8 and Q.sub.9 is conductive.

The output of amplifier A.sub.3 is connected by a resistor R.sub.30 to the non-inverting input of a comparator A.sub.4 which has its inverting input grounded. The output of comparator A.sub.4 is connected by two resistors R.sub.31 and R.sub.32 in series to its inverting input, the junction of these resistors being connected by two oppositely connected zener diodes D.sub.13, D.sub.14 to ground. The output of amplifier A.sub.4 is connected to the junction of resistors R.sub.26 and R.sub.27 so as to control the f.e.t.s. Q.sub.8 and Q.sub.9.

The amplifier A.sub.3 and the comparator A.sub.4 operate together as a triangular waveform generator with the output of amplifier A.sub.3 swinging between two fixed positive and negative voltage levels set by the zener diodes D.sub.13 and D.sub.14. The frequency of the waveform is directly proportional to the voltage. The maximum value of the stepping frequency is an order of magnitude less than the frequency of the chopper.

The output of amplifier A.sub.3 is connected to terminal B and is also connected by a resistor R.sub.34 to the inverting input of a comparator A.sub.5, the non-inverting input of which is connected to ground by a resistor R.sub.33. The output of comparator A.sub.5 is connected to terminal A and is also connected by a relatively high ohmic value resistor R.sub.35 to its non-inverting input. This ensures sharp switching of the comparator between its two output states, the comparator A.sub.5 operating as a zero-crossing detector.

Assuming that the zener diodes D.sub.13 and D.sub.14 have breakdown voltages of 6.8 V, the output of amplifier A.sub.3 ramps between -7.5 V and +7.5 V. The output of comparator A.sub.5 is -12 V when the A.sub.3 output is positive and +12 V when it is negative. The effect of resistors R.sub.14 to R.sub.17 and R.sub.19 and diodes D.sub.11 and D.sub.12 is to determine which one of inputs of the i.c. 15 receives a real demand signal in the range of 0 to .+-.75 mV and which one receives a false signal, the magnitude of which is increase by current flowing through resistor R.sub.19.

Considering a cycle of operation which commences when the output of amplifier A.sub.3 is at -7.5 V and rising, the signal at terminal B will be -7.5 V and that at terminal A, +12 V. As a result the signal at the inverting input of the c.l. sense amplifier of i.c. 15 will be -75 mV whereas that at the non-inverting input of the error amplifier will be positive. The internal "high-wins" diode gate in i.c. 15 ensures that only the error signal corresponding to the difference between the -75 mV input signal and the voltage across R.sub.1 has any controlling effect. The i.c. 15 operates to route fixed frequency variable duration pulse through the E.sub.1 C.sub.1 and E.sub.2 C.sub.2 path within the i.c. 15. Since transistor Q.sub.4 is on as a result of the output of comparator A.sub.5 being +12 V, transformer TR.sub.B is pulsed so that transistor Q.sub.2 is pulsed on and off appropriately. The current level in the winding 10 and 11 is regulated in this way to make the voltage across resistor R.sub.1 -75 mV.

As the output of amplifier A.sub.3 rises the current in resistor R.sub.1 is progressively reduced. When the output of amplifier A.sub.3 crosses zero, the output of comparator A.sub.5 changes to -12 V so that the signal at the inverting input of the C.L. sense amplifier falls instantaneously to a relatively large negative value whereas that at the non-inverting input of the i.c. error amplifier falls to OV. At the same time transistor Q.sub.4 switches off and transistor Q.sub.3 switches on to cause the pulses from i.c. to control Q.sub.1 instead of Q.sub.2 and the forward current passed increases as the output of amplifier A.sub.3 becomes increasingly positive. The inductors L.sub.1 and L.sub.2 serve to limit the current rise should transistors Q.sub.1 and Q.sub.2 become conductive simultaneously for a period during the zero crossing switch-over. Furthermore the windings L.sub.3, L.sub.4, which make up 1:1 transformers (at the switching frequency) with the respective inductors L.sub.1 and L.sub.2, the diodes D.sub.1, D.sub.2 and the capacitor K.sub.3 ensure that motor current is maintained whilst either Q.sub.1 or Q.sub.2 is switched off, and is "dumped" in the appropriate supply rail 12 or 13 even at low stepping frequency.

Similarly the current in windings 10 and 11 is controlled during the period when the output of amplifier A.sub.3 is ramping downwardly.

The stepper motor also includes a second pair of windings (not shown) which are to be driven with a current waveform 90.degree. out of phase with that in the windings 10 and 11 and either lagging or leading depending on the required direction of drive. To this end a second integrating amplifier A.sub.6, and further comparators A.sub.7 and A.sub.8 are provided. Amplifier A.sub.6 and comparator A.sub.8 provide signals to terminals D and C which are inputs to another circuit exactly the same as that shown in the upper half of the drawing as described above, which circuit drives the other two motor windings. Amplifier A.sub.6 integrates the signal at the output of f.e.t.s. Q.sub.6 and Q.sub.7 which are controlled by the signal at the output of comparator A.sub.7.

To ensure that the two waveform generator circuits run in the required phase relationship, an f.e.t. Q.sub.10 is connected between the output and inverting input terminals of amplifier A.sub.6 to ensure that the integrating capacitor associated therewith is held discharged at certain times.

The gate of transistor Q.sub.10 is connected by a resistor R.sub.40 to a -12 V rail 17. The gate of transistor Q.sub.10 is also connected via a capacitor K.sub.4 and a resistor R.sub.50 in series to the collector of a pnp transistor Q.sub.11 which has its emitter connected to the rail 14 and its collector connected by a resistor R.sub.41 to the rail 17. The base of transistor Q.sub.11 is connected by a resistor R.sub.42 to the output terminal of amplifier A.sub.4 and also connected to the junction of two resistors R.sub.43, R.sub.44 in series between the rail 14 and the collector of an npn transistor Q.sub.12. The emitter of the transistor Q.sub.12 is grounded and its base is connected to the junction of two resistors R.sub.45, R.sub.46 in series between a FWD/RVSE terminal and ground. Further the gate of transistor Q.sub.10 is connected by a resistor R.sub.51 and a capacitor K.sub.5 in series with the drain-source of a field effect transistor Q.sub.13 to ground. The gate of transistor Q.sub.13 is connected directly to the FWD/RVSE terminal. A resistor R.sub.52 connects the drain of transistor Q.sub.13 to the output of amplifier A.sub.4.

For forward running the start of the triangular wave at D should coincide with the leading edge of the square wave from amplifier A.sub.4, whereas for reverse running it should coincide with the trailing edge of that square wave. The square wave at the output of amplifier A.sub.4 is inverted by transistor Q.sub.11. When the signal at the FWD/RVSE terminal is high Q.sub.11 is turned on continuously, thereby preventing it from operating as an inverter, but the f.e.t. Q.sub.13 is off. Thus the positive going edges of the square wave output of amplifier A.sub.4 are differentiated by capacitor K.sub.5 and provide positive going spikes coincident with these edges to turn on f.e.t. Q.sub.10. The negative spikes created at the trailing edges of the square waves have no effect. When the signal at the FWD/RVSE terminal is low f.e.t. Q.sub.13 is on and therefore grounds the pulses arriving via resistor R.sub.52. Transistor Q.sub.11 acts as an inverter and the rising edges of the inverted waveform create positive spikes to turn on f.e.t. Q.sub.10 in synchronism with the trailing edges of the square waves from amplifier A.sub.4.

The circuit described ensures that no shock loads are applied to the rotor of the motor. The use of a chopper current control to control current with a triangular waveform or other stepless symmetrical waveform overcomes the problem in known stepper motor controls of shock excited "pendulum" type vibration of the rotor. Thus no pull-in and drop-out problems arise from rotor resonances being excited. The rotor can be stopped in any desired position between the normal stable positions by maintaining the current in the motor windings at the level at which it was flowing at the moment of coming to rest. Controlled acceleration and deceleration of the motor under the control of the computer are easily achieved.

The use of transformer coupling to the output transistors Q.sub.1 and Q.sub.2 and the employment of a three rail supply enable the output stage to be made very simple. Only two high power transistors are needed and two diodes D.sub.1 and D.sub.2 provide the necessary current recirculation paths, with winding current continuing to flow through the sensing resistor R.sub.1.

The integrated circuit 15 employed, although intended for use in power supply regulation, proves to be entirely suitable for stepper motor control and makes the chopper drive circuit extremely simple.

The circuit may also include a current/voltage phase monitor 18 arranged to provide additional current into the summing junction of amplifier A.sub.1 when the current in resistor R.sub.1 is out of phase with the voltage at the junction of inductors L.sub.1 and L.sub.2. In this way the stepping frequency is reduced if the motor is overloaded.

Turning now to FIG. 2, the phase windings 210, 211 of the motor are represented by a single inductor. The current in these windings is controlled by a pair of complementary FETs, Q.sub.201 and Q.sub.202 connected in series between the power supply rails 212, 213. The windings 210, 211 are connected at one end to the common point of these FETs and at the other end via a current sensing resistor R.sub.201 to ground. The FETs Q.sub.201 and Q.sub.202 are again driven by a PWM modulator i.c. 215 (TL494), but in this case no isolating transformers are needed. The C.sub.1 and C.sub.2 outputs of this i.c. are connected together (these are open collector outputs), and are connected by a constant current diode D.sub.201 as load to a zener regulated supply (resistor R.sub.202, zener diodes D.sub.202 and D.sub.203 in series, and a smoothing capacitor K.sub.201 connected across the zener diodes) which is connected between the rail 212 and ground, and also connected by a resistor R.sub.203 to the bases of two complementary transistors Q.sub.203 and Q.sub.204. These transistors have their emitters connected together and have their collectors connected by respective resistors R.sub.204, R.sub.205 to the rails 212, 213. A pair of complementary FETs Q.sub.205, Q.sub.206 which act as drivers for the FETs Q.sub.201, Q.sub.202 respectively have their gates connected to respective ones of the collector of transistors Q.sub.203 , Q.sub.204 and have their drain-source paths connecting the gates of respective ones of the FETs Q.sub.201, Q.sub.202 to the rails 212, 213. The output of FETs Q.sub.201, Q.sub.202 are protected by two zener diodes D.sub.204, D.sub.205 and each FET Q.sub.205, Q.sub.206 is protected by two zener diodes D.sub.206, D.sub.207 and D.sub.208, D.sub.209 connected as shown in FIG. 2. A capacitor K.sub.202 connects the emitter of transistors Q.sub.203, Q.sub.204 to the bases thereof to provide a deadband rapid in the switching on and off of these transistors and, therefore, of the driver FETs Q.sub.205, Q.sub.206 and the output FETs Q.sub.201, Q.sub.202. The emitter of the transistors Q.sub.203, Q.sub.204 are connected by a resistor R.sub.206 in parallel with the series combination of a resistor R.sub.207 and a capacitor K.sub.203 to the common point of zener diodes D.sub.202, D.sub.203, so as to provide fast turn-on following the deadband delay introduced by capacitor K.sub.202.

The V REF output of the i.c. 215 is connected by a resistor R.sub.208 to the error amplifier inverting input thereof, such input being also connected by a capacitor K.sub.204 to the error amplifier output of the i.c. 215 and by a resistor R.sub.209 to the non-earthy side of the current sensing resistor R.sub.201 so as to provide motor current feedback to the i.c. 215. The G.sub.T and R.sub.T terminals of the i.c. 215 are grounded by a capacitor K.sub.205 and a resistor R.sub.210 respectively. The DEAD TIME CONTROL input of i.c. 215 is connected to a terminal A and is also connected by a resistor R.sub.211 to rail 213 and by a diode D.sub.210 to ground.

The non-inverting input of the error amplifier of the i.c. 215 is connected by a resistor R.sub.212 to the output of an operational amplifier A.sub.201 and by resistor R.sub.213 to ground. Amplifier A.sub.201 operates in conjunction with an operational amplifier A.sub.202 an integrated 8-bit counter and digital-to-analog converter 216 (Ferranti ZN425), and a CMOS anolog bilateral switch 217 (1/4 DG211) as a triangular waveform generator which provides a current demand signal to the pulse width modulator based on i.c. 215.

The ANALOG OUTPUT terminal of the converter 216 is connected directly to the non-inverting input of amplifier A.sub.202 which has its output connected by a resistor R.sub.214 to its inverting inpout so that it operates as a unity gain voltage follower. The MSB output of converter 216 is connected to the control terminal of switch 217 and the signal path of the switch 217 is connected between the output of amplifier A.sub.202 and the non-inverting input of amplifier A.sub.201. The V-REF output of converter 216 is connected directly to the V-REF input thereof and is also connected by a resistor R.sub.215 to the non-inverting input of amplifier A.sub.201. The output of amplifier A.sub.202 is connected by a resistor R.sub.216 to the inverting input of amplifier A.sub.201 and a resistor R.sub.217 of the same value as resistor R.sub.216 is connected between the output of amplifier A201 and its inverting input.

The CLOCK input of the converter 216 is connected to a suitable source of clock pulses, the frequency of which determines the stepping rate of the motor. The amplifier A.sub.201 operates as a unity gain inverting amplifier or as a voltage follower according to whether the switch 217 is conductive or not. When the switch 217 is conductive, which occurs when the MSB output of converter 216 is high, ie. when the analog output is at a voltage higher than half the reference voltage output by V REF terminal of converter 216, the output of amplifier A.sub.202 is connected directly to the non-inverting input of amplifier A.sub.201 which acts as a unit gain voltage follower. Thus, as the count in the counter of converter 216 increases, the output of amplifier A.sub.201 increases linearly. When the switch 217 is non-conductive, the amplifier A.sub.201 as a unity gain inverting amplifier so that the output of amplifier A.sub.201 decreases linearly as the count in the counter of converter 216 increases. Since this counter operates cyclically, the ANALOG output of converter 216, provides repeated saw tooth waveform, which the amplifier A.sub. 201 and switch 217 connect into a symmetrical substantially stepless triangular waveform.

A second counter converter 218, a second switch 219 and two amplifiers A.sub.203 and A.sub.204 are connected similarly to provide a second triangular waveform, which is used for providing a current demand signal to another pulse width modulation circuit (not shown) controlling the other two phase windings of the motor.

The relative phase of the two waveform generators is controlled by a logic, circuit which receives inputs from the second and third MSB outputs of the converter 218 and provides reset pulses periodically to the converter 216. This logic circuit comprises seven NAND gates G.sub.1, to G.sub.7. Gate G.sub.1 has its inputs connected respectively to the second and third MSB outputs of converter 218, whereas gate G.sub.3 has one input connected to the third MSB output and the other connected to the output of gate G.sub.2 both inputs of which are connected to the second M5B output to operate as a logic inverter. Gate G.sub.4 has one input from the output of gate G.sub.3 and one from a forward/reverse input terminal 220. Gate G.sub.6 has one input connected to the output of gate G.sub.1 and the other connected to the output of gate G.sub.5, both inputs of which are connected to the terminal 220. Gate G.sub.7 has one input from gate G.sub.4 and one from gate G.sub.6 and drives a differentiator comprising a capacitor K.sub.206 a resistor R.sub.218 and a diode D.sub.211, the output of this differentiator is connected to RESET terminal of converter 216. When the signal on terminal 220 is low (FORWARD) the output of gate G.sub.7 is in phase with the output of gate G.sub.1, i.e. it is high except when both of the second and third M5B outputs are high. A reset pulse is delivered by the differentiator when the output of gate G.sub.7 goes low, thereby resetting the converter 216 halfway through the ascending flank of the waveform of generators 218, 219, A.sub.203, A.sub.204. Similarly, when the signal at terminal 220 is high converter 216 is reset half way through the descending flank of the other waveform.

A simple digital waveform generator as described above has the advantage that the "integrators" constituted by the counter/converter 216 218 have no tendency to drift.

At very low stepping speed the clock can be directly connected to a control computer, thereby enabling very accurate control of the motor shaft position (which is fed back to the computer) down to one sixty-fourth of a step. At high stepping rates a digit-to-frequency converter or a rate multiplier can be used to give accurate speed and displacement control.

Turning now to FIG. 3 a different pulse width modulation circuit (but still using a TL494 integrated circuit) and a different digital waveform generator are employed. The waveform generator simply uses a counter 301 the output of which is fed to two ROMs 302 and 303 which store waveform data and provide outputs to two digital-to-analog converters 304, 305. One ROM 303 can store two sets of waveform data to enable the necessary advance/retard of the waveforms to be obtained for forward and reverse operation. Converter 304 provides a current demand waveform for the pulse width modulator which drives the phase windings 310, 311 of the motor, whereas converter 305 provides a waveform for the other modulator (not shown) for the other phase windings (not shown).

The pulse width modulator shown in FIG. 3 utilizes two HEXFETs Q.sub.301 and Q.sub.302 connected in series with two series inductors L.sub.301, L.sub.302 between the supply rails 312 and 313. Protective diodes D.sub.301 and D.sub.302 are connected across the two HEXFETs Q.sub.301 and Q.sub.302 respectively. The junction of the two inductors L.sub.301, L.sub.302 (which are provided to limit the rate of current rise should the conduction periods of the two HEXFETs overlap) is connected by another inductor L.sub.303 to one end of windings 310, 311, the other end being grounded via a current sensing resistor R.sub.301. The gate of HEXFET 301 is connected to the emitters of a first pair of complementary driver transistors Q.sub.303, Q.sub.304, the gate of HEXFET Q.sub.302 being connected to the emitter of another pair of complementary driver transistors Q.sub.305, Q.sub.306. The collectors of the transistors Q.sub.303, Q.sub.304, are connected respectively to the +28 V rail 312 and to a -12 V rail 314. Similarly the collectors of transistor Q.sub.305, Q.sub.306 are connected to a -12 V rail 315 and the -28 V rail 313. The +12 V, -12 V rails 314, 315 are zener stabilised, utilizing zener diodes D.sub.303, D.sub.304, feed resistors R.sub.302, R.sub.303 and smoothing capacitors K.sub.302, K.sub.303 respectively. The transistors Q.sub.303, Q.sub.304, Q.sub.305 and Q.sub.306 are controlled by another complementary transistor pair Q.sub.307, Q.sub.308 which have their emitters connected to a 4.7 V supply provided by a zener diode D.sub.305 in series with a resistor R.sub.304 between rail 312 and ground. The collector of transistor Q.sub.307 (npn) is connected by two resistors R.sub.305, R.sub.306 in series to rail 312 with the junction of these resistors connected to the bases of the transistors Q.sub.303, Q.sub.304. Similarly two resistors R.sub.307, R.sub.308 in series connect the collector of transistor Q.sub.308 to the rail 313 and the junction of these resistors is connected to the bases of transistors Q.sub.305, Q.sub.306. The bases of the transistors Q.sub.307, Q.sub.308 are connected together and a capacitor K.sub.304 connects them to the emitters of these transistors. Furthermore a resistor R.sub.309 connects the base of transistor Q.sub.307, Q.sub.308 to the C.sub.2 output terminal of i.c. 316, which terminal is connected to rail 312 by a resistor R.sub.310. The "current limit" amplifier of i.c. 316 is not used, its non-inverting input being grounded and its inverting input being connected to the VREF output thereof. The OUTPUT CONTROL and E.sub.2 terminals are also grounded. The chopping frequency of the i.c. 316 is set at about 100 KHz by a resistor R.sub.311 and a capacitor K.sub.305 connecting the R.sub.T and C.sub.T terminals of i.c. 316 to ground. The error amplifier non-inverting input is grounded by a resistor R.sub.312.

In the example shown in FIG. 3 the error amplifier of the i.c. 316 is used merely as an inverting amplifier stage operating on the signal from a operational amplifier A.sub.301. This receives at its inverting input the current demand signal from the converter 304 via a resistor R.sub.313 and a current feedback signal from an operational amplifier A.sub.302 via a resistor R.sub.314. The non-inverting input of amplifier A.sub.301 is grounded via a resistor R.sub.315. Local feedback around amplifier A.sub.301 is provided by a capacitor K.sub.306 and a resistor R.sub.315A in series between the output and inverting input terminals of amplifier A.sub.301, the voltage at the output of amplifier A.sub.301 being limited by two relatively reversed zener diodes D.sub.306, D.sub.307 in series between the inverting input and the output terminals thereof.

Amplifier A.sub.302 acts as a low pass filter operating to filter out the 100 KHz component of the voltage signal generated in the resistor R.sub.301 when current is flowing in windings 310, 311. Resistor R.sub.301 is connected by resistors R.sub.316, R.sub.317 in series to the inverting input of amplifier A.sub.302, the junction of these resistors being connected to ground by a capacitor K.sub.307. The non-inverting input of amplifier A.sub.302 is grounded via a resistor R.sub.318 and feedback around amplifier A.sub.302 is provided by a capacitor K.sub.308 connected between the output terminal and the inverting input terminal, and by a resistor R.sub.319 connected between the output terminal and the junction of resistors R.sub.316 and R.sub.317.

The DEAD TIME CONTROL terminal of i.c. 316 is biassed to a small negative voltage and the inverting input of the error amplifier of i.c. 316 is biased to an adjustable negative voltage by means of a resistor chain R.sub.320, R.sub.321, R.sub.322, R.sub.323 connected between the anode of a 4.7 V zener diode D.sub.308 and ground. The cathode of zener diode D.sub.308 is grounded and its anode is connected by a resistor R.sub.324 to the -12 V rail 315. The resistor R321 is, in fact, a preset potentiometer and has its slider connected by a resistor R.sub.325 to the inverting input of the error amplifier of i.c. 316. The junction of resistors R.sub.322 and R.sub.323 is connected to the DEAD TIME CONTROL input. The error amplifier receives its input from operational amplifier A.sub.301 via a resistor R.sub.326 and local negative feedback is provided by a resistor R.sub.327.

In the example shown in FIG. 3, the pulse width modulator operates by switching HEXFETs Q.sub.301 and Q.sub.302 on alternately at the chopping frequency referred to and thereby maintaining the actual average current in the windings 310, 311 at the desired level set by converter 304.

The arrangement shown in FIG. 4 includes an analog waveform generator driving a simple bipolar transistor pulse width modulator.

The windings 410, 411 are connected at one end to the junction of two inductors L.sub.401, L.sub.402 connected in series between the collectors of two output transistors Q.sub.401, Q.sub.402, and at the other end via a current sensing resistor R.sub.401 to ground. Transistor Q.sub.401 is a pnp transistor and has its emitter connected to the positive supply rail 412, whereas transistor Q.sub.402 is an npn transistor which has its emitter connected to the negative supply rail 413. The transistor Q.sub.401, Q.sub.402 are controlled by an operational amplifier A.sub.401 which has a load resistor R.sub.402 connected between its output terminal and ground. The positive and negative supply terminals of amplifier A.sub.401 are connected to rails 412 and 413 by respective resistors R.sub.403, R.sub.404 and are also connected respectively to the bases of the transistors Q.sub.401 and Q.sub.402. Positive feedback is provided via a resistor R.sub.405, in parallel with the series combination of a resistor R.sub.406 and a capacitor K.sub.401, between the collector of the transistor Q.sub.401 and the non-inverting input of amplifier A.sub.401. Negative feedback is provided by a resistor R.sub.407 connected between resistor R.sub.401 and the inverting input of amplifier A.sub.401, a capacitor K.sub.402 connecting this inverting input to the rail 413. A current demand signal is applied to the non-inverting input of amplifier A.sub.401 via a resistor R.sub.408 and a resistor R.sub.409 connects this non-inverting input to the rail 413.

The amplifier A.sub.401 in fact operates as a voltage comparator. When the voltage at the inverting input is higher than that at the non-inverting input current flows from ground through the resistor R.sub.402 into the output of amplifier A.sub.401 and hence through resistor R.sub.404 to rail 413. Thus transistor Q.sub.402 turns on. When the voltage at the inverting input is lower than that at the non-inverting input, transistor Q.sub.403 turns on. Resistor R.sub.405 changes the voltage at non-inverting input of amplifier A.sub.401 to cause the amplifier A.sub.401 to operate in the manner of a Schmidt trigger with upper and lower thresholds linearly dependent on the current demand signal. Current feedback via resistor R.sub.407 shunted by capacitor K.sub.402 makes the whole circuit into an oscillator and the inductive load ensures that the charge/discharge rates of capacitor K.sub.402 depend on the average current flow in the load so that pulse-width modulation is obtained to regulate the current to the demand value.

The current demand signal is derived from an operational amplifier A.sub.402 connected as an active integrator having an input resistor R.sub.410 and a feedback capacitor K.sub.403. A resistor R.sub.411 connects the non-inverting input of amplifier A.sub.402 to ground. The output of amplifier A.sub.402 is connected by a resistor R.sub.412 to the non-inverting input of an amplifier A.sub.403 the inverting input of which is connected by a resistor R.sub.413 to ground. Two resistors R.sub.414 and R.sub.415 in series connect the output of amplifier A.sub.403 to its non-inverting input, the junction of such resistors being connected to ground by two back-to-back zener diodes D.sub.401 and D.sub.402 in series. The junction of resistors R.sub.414 and R.sub.415 is also connected by a resistor R.sub.416 to the integrator input resistor R.sub.410. Two capacitors K.sub.404 and K.sub.405 are connected by oppositely poled diodes D.sub.403, D.sub.404 to the junction of resistors R.sub.416 and R.sub.410, such capacitors being grounded. An amplifier A.sub.404 is connected to ensure that the voltages on capacitors K.sub.404 and K.sub.405 are equal and opposite. To this end amplifier A.sub.404 has its inverting input connected by a resistor R.sub.417 to the capacitor K.sub.404 and its non-inverting input grounded via a resistor R.sub.418. A feedback resistor R.sub.419 connects the output of amplifier A.sub.404 to its inverting input to provide unity gain inverting operation and the output of amplifier A.sub.404 is connected to capacitor K.sub.405.

It will be appreciated that the amplifier A.sub.403 operates as a Schmidt trigger circuit determining the polarity of the signal applied via the resistor R.sub.410 to the integrating amplifier A.sub.402. However, the magnitude of the signal applied to the integrator depends on the voltage for the time being stored on the capacitors K.sub.404, K.sub.405. The circuit thus far described therefore operates as a triangular waveform generator, the amplitude of the waveform being fixed, but the frequency being inversely proportional to the voltage stored on the capacitors K.sub.404, K.sub.405.

A second integrator is provided for generating an identical waveform, suitably phase-shifted, for the other phase windings (not shown) of the motor. This second integrator includes an operational amplifier A.sub.405 which has its non-inverting input grounded via a resistor R.sub.420. An input resistor R.sub.421 for this integrator is connected at one end to the anode of a diode D.sub.405, which has its cathode connected to the capacitor K.sub.404, and to the cathode of a diode D.sub.406 which has its anode connected to the capacitor K.sub.405, and at the other end to the inverting input of amplifier A.sub.405. A feedback capacitor K.sub.406 is connected between the output and non-inverting input terminals of the amplifier A.sub.405. Two zener diodes D.sub.407, D.sub.408 connected back-to-back are also connected between the output and inverting input terminals of amplifier A.sub.405.

The polarity of the signal integrated by amplifier A.sub.405 is determined by an operational amplifier A.sub.406 which is connected to operate as a zero-crossing detector. The two inputs of this amplifier A.sub.406 are connected by respective resistors R.sub.422 and R.sub.423 to the output of amplifier A.sub.402. The inverting input is connected by a resistor R.sub.424 to ground whereas the non-inverting input is connected by the drain-source path of an FET Q.sub.403 to ground. The amplifier A.sub.406 operates in inverting or non-inverting mode according to whether the FET Q.sub.403 is conductive or not. The gate of FET Q.sub.403 is connected to a FWD/RVS input terminal 415, the signal at which thus determines the mode of operation of the amplifier A.sub.406. The output terminal of amplifier A.sub.406 is connected by a resistor R.sub.425 to the anode of a zener diode D.sub.409, which is connected back-to-back with a zener diode D.sub.410 having its anode connected to ground. A resistor R.sub.426 connects the anode of zener diode D.sub.409 to the anodes of the diode D.sub.405.

Each of the integrators A.sub.402 and A.sub.405 has a resetting circuit associated with it and these are controlled by a single npn transistor Q.sub.404 with its emitter grounded and its collector connected to the +15 V rail by a resistor R.sub.427. The base of this transistor is connected by a resistor R.sub.428 to the +15 V rail, by a resistor R.sub.429 to the capacitor K.sub.405 and by a diode D.sub.411 to ground, the cathode of diode D.sub.411 being connected to the base of transistor Q.sub.404. The resetting circuit associated with amplifier A.sub.402 includes an FET Q.sub.405 with its gate connected to the collector of transistor Q.sub.404, its drain connected by a resistor R.sub.430 to the collector of transistor Q.sub.404 and by two back-to-back zener diodes D.sub.412 and D.sub.413 to the output of amplifier A.sub.402 and its source connected to the inverting input of amplifier A.sub.402. FET Q.sub.405 is conductive whenever the transistor Q.sub.404 is of which occurs when the voltage on capacitor K.sub.405 is sufficient to hold the base of transistor Q.sub.404 low. In these circumstances the zener diodes D.sub.412 and D.sub.413 determine the peak positive and negative excursion of the output of amplifier A.sub.402 instead of the amplifier A.sub.403. The resetting circuit of amplifier A.sub.405 is identical and comprises a similarly connected FET Q.sub.406, two zener diodes D.sub.414 and D.sub.415 and a resistor R.sub.431.

The voltage on capacitor K.sub.404 is limited by an external limiting circuit (not shown) to which it is connected via a PRO MAX terminal 414. A STOP/RUN terminal 416 is connected to the anode of a diode D.sub.416 the cathode of which is connected to the cathode of a diode D.sub.417 having its anode connected to the capacitor K.sub.404. A constant current diode D.sub.418 connects the cathodes of diodes D.sub.416, D.sub.417 to the -15 V rail. When the signal at the RUN/STOP terminal is low the capacitor K.sub.404 discharges linearly to zero through the constant current diode D.sub.418. When the signal at the RUN/STOP diode is high, capacitor K.sub.404 can charge up vai resistor R.sub.416 and diode D.sub.403 from the output of amplifier A.sub.403.

The circuit also includes a stall or slip detector which operates by comparing the phases of the voltage across and current in the winding 410, 411. This detector circuit includes an operational amplifier A.sub.407 which has its non-inverting input grounded by resistor R.sub.432 and its inverting input connected to the capacitor K.sub.402 of the pulse width modulator circuit. This amplifier A.sub.407 detects zero crossings in the average current waveform. Another operational amplifier A.sub.408 has its inverting input grounded by a resistor R.sub.433 and its non-inverting input connected by a resistor R.sub.434 to the collector of the transistor Q.sub.402, and by a capacitor K.sub.407 in parallel with a variable resistor R.sub.435 to ground. Resistor R.sub.434 and capacitor K.sub.407 operate as an averaging filter and variable resistor R.sub.435 acts to provide phase compensation at higher speeds. Amplifier A.sub.408 thus detects zero crossings in the averaged and phase compensated voltage waveform. The amplifier A.sub.407 and A.sub.408 have open-collector outputs and the output terminals of these are connected together and are connected by a resistor R.sub.436 to the rail 412. These output terminals are also connected by a resistor R.sub.437 to the inverting input of an amplifier A.sub.409, such input also being connected by a capacitor K.sub.408 to ground. The non-inverting input of amplifier A.sub.409 is connected to a reference voltage source which supply a reference voltage representing the maximum torque which the motor can provide without slipping. The amplifiers A.sub.407 and A.sub.408 form a phase detector such that, when the voltage and current waveforms are exactly in phase one or other of the outputs is zero, thereby holding capacitor K.sub.408 discharged. As the torque on the motor is increased the phase difference between the current and voltage waveforms increases so that for an increasing proportion of the time the outputs of both amplifiers A.sub.407, A.sub.408 are high simultaneously. Thus capacitor K.sub.408 charges to a voltage which is dependent on the phase-shift, i.e. on the torque load on the motor.

The output of amplifier A.sub.409 is connected to a SLIP/STALL terminal which may be connected to a stall indicator. In addition the output of amplifier A.sub.409 is connected to the cathode of a diode D.sub.419, the anode of which is connected by a resistor R.sub.438 to the capacitor K.sub.404.

In use, as mentioned above, when the signal at the STOP/RUN terminal 416 is low capacitor K.sub.404 is held discharged so that neither integrator A.sub.402 nor A.sub.405 operates. When the signal at terminal 416 goes high the capacitor K.sub.404 will start to change from which ever one of the amplifiers A.sub.403, A.sub.406 is producing a high output. The signal which is integrated by the amplifiers A.sub.402, A.sub.405 is thus initially small in magnitude and the rate at which the motor is stepped is therefore correspondingly slow. As the capacitor K.sub.404 charges, however, the stepping rate increases until the maximum step rate determined by the transistor Q.sub.404 is reached (or determined by the PRO MAX input if this is less.) The two waveforms produced are maintained 90.degree. out of phase by the operation of amplifier A.sub.406 and any drift which occurs in the integrating amplifier A.sub.402, A.sub.405 is dealt with automatically by clipping of the waveforms by operation of the resetting circuits referred to (unless the PRO MAX control is in use).

Should slip or stall occur the output of amplifier A.sub.409 goes low, causing the capacitor K.sub.404 to discharge through the resistor R.sub.438 and the diode D.sub.419, thereby reducing the stepping rate to aa level at which the output of amplifier A.sub.409 goes high, whereupon capacitor K.sub.404 will start to recharge.

In all of the circuits described above a waveform generator circuit provides a pair of substantially stepless symmetrical periodic waveforms which provide current demand signals to two pulse width modulation circuits driving the phase windings of the motor. Provision is made in every case for maintaining a phase difference of 90.degree. (either a lead or a lag for forward/reverse control) between the waveforms and the step rate is variable either by changing the frequency of a clock signal in the case of the digital waveform generators, or by varying an analog signal which is integrated in the case of the analog waveform generators. In every case these measures provide smooth and accurate control of the stepper motor and permit running of the motor at higher stepping rates than are usually possible.