Demodulated Chrominance Signal Filter With Peaked Response

This invention concerns a composite electrical signal filter suitable for processing a signal such as the demodulated "I" chrominance information signal component of a color television signal.

In accordance with the NTSC television signal broadcast standards adopted in the United States, the chrominance component of a composite broadcast color television signal is formed by encoding a pair of color difference signals as the modulation of a pair of chrominance subcarrier signals, where the subcarrier signals exhibit the same frequency but a mutually quadrature (90.degree.) phase relationship. At the color television signal receiver, the pair of color difference signals must be separated or demodulated from the composite color signal. Thereafter, the separated color components are matrixed with the separated luminance component of the composite color television signal to form the drive signals for the kinescope of the television receiver.

One of the color difference signal components, commonly referred to as the "Q" signal, is transmitted as a double sideband, relatively narrow band signal of approximately 0-0.5 MHz bandwidth. The other color difference signal component, commonly referred to as the "I" signal, is transmitted as a relatively wide band signal of approximately 0-1.5 MHz bandwidth. In order to fit the I,Q encoded chrominance signal within the allocated 6 MHz modulated video signal bandwidth, the upper sideband of the I signal is eliminated prior to transmission. The transmitted I signal component thus contains double sideband information over a 0-0.5 MHz bandwidth, and single sideband information over a 0.5-1.5 MHz bandwidth.

The method of transmitting the I signal using both single and double sideband information is an efficient way of utilizing the frequency spectrum allocated for broadcast color television signals. However, the frequency spectrum energy associated with the single sideband information of the I signal is only one-half of what it would be if the I signal was fully double sidebanded. Therefore in the process of recovering the I signal at the receiver, it is necessary to compensate for the spectrum amplitude loss associated with the absence of one of the I signal sidebands. This can be accomplished by performing frequency selective signal peaking in the course of the I signal recovery process.

Circuits utilized to separate or demodulate the I and Q signals from the composite color signal can include I and Q synchronous demodulators, and associated I and Q signal filters. Delay compensation should be provided to assure that the filtered demodulated I and Q signals exhibit proper phase and timing synchronism when these signals are combined to develop plural red, green and blue color difference signals which are eventually matrixed with the luminance component of the composite color television signal to form the kinescope drive signals.

In accordance with the principles of the present invention, there is disclosed herein a filter network suitable for processing a signal such as a demodulated I chrominance component by providing a given signal delay, wideband signal filtering, and a peaked amplitude response over a given frequency range.

The disclosed filter comprises first and second cascaded resonant sections each comprising an inductance and a capacitance, and each being terminated by resistors which differ substantially in value. An intermediate resonant section comprising an inductance and capacitance is coupled between and in cascade with the first and second sections. The first and second resonant sections respectively exhibit impedances, at a frequency within the peaking frequency range, which approximate the value of the associated terminating resistor. In addition, the filter DC impedance is primarily determined by the values of the terminating resistors.

In accordance with a feature of the invention, the intermediate resonant section comprises plural cascaded inductance-capacitance resonant sections which respectively exhibit, at frequencies within the peaking frequency range, progressively increasing impedance values between the impedance values exhibited by the first and second resonant sections within the peaking frequency range.

In accordance with a further feature of the invention, the filter network comprises a high order, three-terminal composite filter coupled via a single interface terminal to a signal path conveying signals to be filtered.

In the drawing:

FIG. 1 shows a portion of a color television receiver including a color signal demodulator network and an associated filter network according to the present invention; and

FIG. 2 depicts the amplitude-versus-frequency response of the filter network according to the present invention.

In FIG. 1, chrominance information signals from a source 10 are applied to a chrominance signal processor 12, which includes chrominance gain control and phase control networks and various other signal processing circuits of a conventional nature, including a controlled local oscillator for regenerating a color subcarrier reference signal at a frequency of approximately 3.58 MHz. Processed chrominance information signals from processor 12 are applied to information signal inputs of an "I" phase chrominance signal demodulator 14 and a "Q" phase chrominance signal demodulator 18. A regenerated 3.58 MHz chrominance subcarrier reference signal from processor 12 is applied to a reference signal input of demodulator 14, and a mutually quadrature phased subcarrier reference signal is applied to a reference signal input of demodulator 18 via a 90.degree. phase shifting network 20. After appropriate filtering as will be discussed, demodulated I and Q chrominance signal components from the outputs of demodulators 14 and 18 are combined in a matrix amplifier 22 for developing output R-Y, G-Y and B-Y color difference signals. These signals are ultimately combined with the luminance component of the composite color television signal to produce red, green and blue color image representative signals for application to a color image display device.

The signal path within I demodulator 14 includes a phase detector (a synchronous demodulator) and an output circuit for receiving demodulated signals. The output circuit is shown as including an amplifier transistor 15 with an associated collector output load resistor 16, and an emitter follower transistor 17. Functionally corresponding transistors 25, 27 and a load resistor 26 are included in the signal path of Q demodulator 18 for receiving demodulated Q signals. The emitter electrodes of transistors 25 and 15 are respectively coupled via interfacing terminals A and B to Q and I signal filter networks 30 and 40. When I and Q demodulators 14 and 18 are constructed in an integrated circuit device, terminals B and A correspond to external connecting terminals of the integrated circuit device.

in a color television signal processing system according to NTSC broadcast standards such as employed in the United States, the Q signal modulation component of the chrominance signal from processor 12 occupies an approximately 0.5 MHz bandwidth on both upper and lower sidebands with respect to the 3.58 MHz frequency of the chrominance subcarrier signal. Thus in the modulated chrominance frequency spectrum the Q chrominance information to be demodulated includes signal frequencies from 3.08 HMz to 4.08 MHz between the -3 db points. The quadrature phased I signal modulation component of the chrominance signal from processor 12 occupies an approximately 1.5 MHz bandwidth on a lower sideband relative to the chrominance subcarrier frequency, and occupies a 0.5 MHz bandwidth on an upper sideband relative to the chrominance subcarrier signal frequency. Thus in the modulated chrominance frequency spectrum the I chrominance information to be demodulated includes signal frequencies from 2.08 MHz to 4.08 MHz between the -3 db points.

Filtering of the demodulated I and Q signals is respectively accomplished by filter networks 40 and 30 to provide appropriate I and Q signal information to the inputs of matrix 22, and to eliminate unwanted signal frequencies such as harmonics of the 3.58 MHz chrominance subcarrier signal. Q signal filter 30 can comprise, for example, a high order low pass filter with a 0-0.5 MHz amplitude vs. frequency response characteristic between the -3 db points. Filter 30 can be of the type shown in a copending U.S. patent application of W. E. Sepp, Ser. No. 470,618 filed Feb. 28, 1983, titled "High Order Electrical Signal Filters", incorporated herein by reference.

I signal filter network 40 according to the present invention comprises an eleventh order composite LC filter including series connected inductors L.sub.1 -L.sub.6 and associated shunt connected capacitors C.sub.1 -C.sub.5. The configuration of the composite filter is that of a filter having three terminals (a,b,c) and two signal ports (a,b) between which the filter transfer function is developed.

The composite LC filter, which may be provided in encapsulated form, comprises a peaked high order low pass filter with an amplitude vs. frequency response characteristic (transfer function) as shown in FIG. 2. The composite filter exhibits significant attenuation above approximately 2 MHz for attenuating the 3.58 MHz subcarrier frequency and harmonics thereof. The composite filter also exhibits a relatively constant delay of about 500 nanoseconds to 1.5 MHz. In the latter regard it is noted that both filter networks 30 and 40 exhibit substantially equal amounts of signal delay so that the signal transit delay times of the I and Q signal processing paths are substantially equal. The frequency bandwidth of I filter network 40 is significantly greater (i.e., three times greater) than that of Q filter network 30. Thus I filter network 40 exhibits a higher order filter configuration relative to that of Q filter 30, in order to provide enough signal delay in the I signal path so that the signal processing delays of the I and Q signal paths are substantially equal.

The amplitude response of the composite filter is relatively "flat" over the low frequency portion of the I signal bandwidth from 0-0.5 MHz, and is peaked over the higher frequency portion of the I signal bandwidth from 0.5 MHz to 1.5 MHz. Such signal peaking suitably compensates for the amplitude loss, in the demodulated I signal, associated with the absence of one of the sidebands in the high frequency portion of the transmitted I signal.

The response indicated by FIG. 2, and in particular the peaked response thereof, results from the manner in which the composite filter is unequally terminated, and from the manner in which the plurality of cascaded inductance-capacitance (LC) resonant sections are arranged, as follows.

Composite filter 40 is unequally terminated at opposite ends (terminals a and b) by means of terminating resistors 41 and 43 of substantially different values. Composite filter network 40 exhibits a DC impedance primarily determined by the sum of the values of terminating resistors 41 and 43. With the values illustrated for these resistors, the filter terminating impedances exhibit a mismatch ratio of 9:1.

The composite filter comprises a plurality of mutually interactive cascaded resonant LC sections L.sub.1 C.sub.1, L.sub.2 C.sub.2, L.sub.3 C.sub.3, L.sub.4 C.sub.4 and L.sub.5 C.sub.5. The first section L.sub.1 C.sub.1 as connected to 200 ohm terminating resistor 41 via terminal a, exhibits an impedance of approximately 207 ohms at a frequency of 1 MHz which corresponds to the midband frequency of the high frequency portion of the I signal bandwidth over which peaking is developed. Thus the impedance of section L.sub.1 C.sub.1 at 1 MHz substantially matches the impedance of terminating resistor 41. The other end of the composite filter is terminated by an LC section comprising an inductor L.sub.6 and capacitor C.sub.5, connected via terminal b to higher value terminating resistor 43. This section exhibits an impedance of approximately 1.1 kilohms at a frequency of 1 MHz, which acceptably approximates the 1.8 kilohm value of terminating resistor 43 at this frequency. Acceptable impedance matching between terminal LC sections L.sub.1 C.sub.1 and L.sub.6 C.sub.5 and associated terminating resistors 41 and 43 is produced if the impedance exhibited by the respective terminal LC section, at 1 MHz, is not more than twice or less than one-half the value of the associated terminating resistor. Intermediate sections L.sub.2 C.sub.2, L.sub.3 C.sub.3, L.sub.4 C.sub.4 and L.sub.5 C.sub.5 exhibit progressively increasing impedance values at 1 MHz, i.e., from somewhat above 200 ohms for section L.sub.2 C.sub.2 to somewhat less than 1.1 kilohms for section L.sub.5 C.sub.5. The illustrated values for the inductors and capacitors of network 40 were chosen from among normally commercially available standard component values.

In addition, the component values of sections L.sub.1 C.sub.1, L.sub.2 C.sub.2, L.sub.3 C.sub.3, L.sub.4 C.sub.4 and L.sub.5 C.sub.5 were chosen to exhibit resonances at 1.5 HMz, 3.1 MHz, 2.1 MHz, 1.7 MHz and 1.0 MHz, respectively. These frequencies correspond to the frequency "poles" of the composite filter, with the higher frequency poles having a progressively higher "Q" (figure of merit) since progressively increasing amplitude peaking is wanted at progressively increasing frequencies, compared to the lower frequency poles at which less peaking is wanted. The lower frequency poles are associated with larger delay and less amplitude peaking, compared with the higher frequency poles which are associated with significantly less delay but a greater amount of amplitude peaking.

The terminating resistors can exhibit a mismatch ratio other than 9:1, e.g., 4:1. However, smaller mismatch ratios result in a smaller amount of amplitude peaking, while excessively large mismatch ratios can result in unwieldy values of inductors and capacitors being required, particularly for the terminal LC sections. Specifically, in the latter case the values of the inductors and capacitors can become sufficiently small and large, respectively, such that the effects of component value tolerances can introduce additional design difficulties.

The DC resistance associated with the series path including inductors L.sub.1 -L.sub.6, between terminals "a" and "b", should be negligible. For example, a resistor of significant value disposed in such series path can undesirably upset the desired peaked low pass response of the filter, such as by damping the amount of amplitude peaking and by introducing other unwanted effects. Inductors L.sub.1 -L.sub.6 should be high "Q" elements with negligibly small DC resistance values of less than 5 ohms, preferably on the order of 1 or 2 ohms. Also, inductors L.sub.1 -L.sub.6 should each exhibit a self-resonant frequency (i.e., with associated parasitic capacitances) significantly above 1.5 MHz, and preferably should not exhibit mutual inductance coupling.

In this embodiment, amplifier transistor 15 in I demodulator network 14 is intended to exhibit a substantially unity DC gain, and an AC gain in accordance with the transfer function of filter network 40 for developing a demodulated I signal voltage across output resistor 16 in accordance with the transfer function of filter network 40. The substantially unity DC gain requirement dictates that the DC impedance presented to the emitter of transistor 15 via terminal B exhibits substantially the same value (2 kilohms) as the DC collector impedance represented by collector resistor 16. This requirement is satisfied by composite filter network 40, which exhibits a DC impedance primarily determined by the sum of the values of terminating resistors 41 and 43. The composite filter additionally provides a DC bias current path from the emitter of transistor 15 to ground via terminal B, resistor 43, inductors L.sub.1 -L.sub.6, and resistor 41.

As will be described in greater detail subsequently, network 40 synthesizes, at interface terminal B, an impedance related to the transfer function of the composite LC filter. Signal currents conducted by the emitter-collector path of amplifier transistor 15 in I demodulator network 14 accordingly exhibit the transfer function of the composite filter, and a corresponding signal voltage developed across output load resistor 16 also exhibits the composite filter transfer function. Thus the demodulated I signal, as applied to the base input of transistor 15, is provided from the output of demodulator network 14 in accordance with the transfer function of the composite I filter.

The recovered I signal provided by demodulator 14 in accordance with the peaked transfer function shown by FIG. 2 suitably compensates for the amplitude loss associated with the absence of one of the sidebands in the high frequency portion of the transmitted I signal. Thus greater use is made of the available information content of the wideband I signal, particularly in the high frequency portion thereof containing information relating to orange and cyan color information. Wide bandwidth (0-1.5 MHz) processing of the I signal in accordance with the disclosed arrangement is particularly advantageous in high definition color television signal processing systems, such as color receivers employing comb filtering techniques for extracting the luminance and chrominance components from the composite color television signal prior to luminance and chrominance processing. With wide bandwidth I demodulation, greater use can be made of the available color information content of the television signal, resulting in improved color picture definition and enhanced subjective color sharpness.

The arrangement of filter network 40 is particularly advantageous since it presents a means by which a high order, two-port, three terminal filter can be connected to associated circuits in the signal path by means of only a single interfacing terminal, e.g., terminal B. This result is desirable when there is only a single terminal (e.g., terminal B) of an integrated circuit (such as comprising circuit 14) available to provide a transfer function of a filter normally requiring both input and output terminals coupled in the signal path. Additional aspects of the operation of filter network 40 will now be described in detail.

Network 40 also includes a substantially unity voltage gain PNP emitter follower transistor 47 with a high impedance base signal input electrode coupled to terminal B via a small (100 ohm) resistor 48, and a low impedance emitter output electrode coupled to filter terminal c. The arrangement of the composite LC filter with transistor 47 causes the collector current of amplifier transistor 15 to exhibit a transfer function H(s) corresponding to the transfer function H(s) of the composite filter. Accordingly, an output signal voltage developed across collector load resistor 16 exhibits the transfer function of the composite filter.

The demodulated I signal voltage developed at the low impedance emitter of transistor 15 appears at terminal B and at the base input of voltage follower transistor 47. Transistor 47 exhibits a substantially unity signal voltage gain (e.g., approximately 0.98), and applies the I signal voltage via the low impedance emitter of transistor 47 to intermediate terminal c of the composite filter. In essence, transistor 47 acts as a second source of signal voltage for application to the composite filter via filter terminal c, the first source of signal voltage corresponding to the emitter of amplifier transistor 15 which is coupled to filter terminal "b" via terminal B and resistor 43.

It is noted that with this arrangement, signal voltages of similar phase and substantially equal magnitude are respectively applied to intermediate filter terminal c, and to resistor 43 which is connected to filter terminal "b" (i.e., a substantially zero signal voltage differential exists between filter terminal c and terminal B). Also, resistor 41 connects filter terminal "a" to a point of fixed potential (ground). Thus filter terminals b and c are excited with input signal voltage whereas filter terminal "a" is not. As a result, the current conducted by resistor 41 varies in accordance with the input signal voltage applied to terminal B, but does not exhibit the transfer function of the composite filter. However, the current conducted by resistor 43 and terminal B exhibits the transfer function of the composite filter. This current corresponds to the collector-emitter current of amplifier transistor 15, and causes load resistor 16 to exhibit a signal voltage having the transfer function of the composite filter.

To better understand the operation of network 40, assume for the moment that the three terminal (a,b,c) composite filter is connected in a conventional configuration. In such case intermediate terminal "c" would be connected to point of fixed reference potential (e.g., ground), and terminal "b" would be connected to ground via resistor 43. Input signals would be applied to terminal "a" via "input" resistor 41, and filtered signals would appear at terminal "b" across "output" resistor 43. In such a conventional filter configuration, terminal "a" alone would be excited with input signals, while terminal c and the end of "output" resistor 43 which is remote from terminal b would both be at a fixed potential relative to input terminal a. The current conducted by "output" resistor 43 would exhibit the transfer function of the three terminal filter. This same result is achieved by the arrangement of network 40, wherein both filter terminals b and c are similarly excited with signals, but filter terminal a is coupled to fixed ground reference potential. Thus the disclosed arrangement of network 40, wherein two of the filter terminals are excited with signals relative to the third terminal which is coupled to a fixed potential, produces a result equivalent to that produced by a conventional three terminal filter configuration wherein only one of the filter terminals is excited with signal relative to the other two terminals which are coupled to fixed potentials.

Network 40 synthesizes, at terminal B, an impedance which is related to (i.e., the reciprocal of) the transfer function of the three-terminal composite filter. The signal current conducted by resistor 43, terminal B and the collector-emitter path of transistor 15 corresponds to the product of the I signal voltage at the emitter of transistor 15 and the transfer function of the composite filter. In this regard it is noted that the emitter signal voltage of transistor 15 substantially corresponds to the base signal voltage of transistor 15, but the emitter and collector signal currents of transistor 15 exhibit the composite filter transfer function.