Phase Coded Pulse Expander-Compressor

The invention relates to pulse-compression techniques, and more particularly to a pulse-compression technique using a fast Fourier transform circuit in a matched filter.

It is known to raise the average transmitted power of a given radar by expanding the pulse length and simultaneously keeping a constant bandwidth so that the range resolution capability of the radar is not reduced. To achieve this, a long pulse containing some sort of a linear frequency modulation or stepwise approximation to a liner frequency modulation is transmitted. Upon reception, the pulse is compressed to permit separation of adjacent range resolution cells.

U.S. Pat. No. 4,237,461 issued to Ben H. Cantrell on Dec. 2, 1980, discloses a digital pulse-expander-compressor which uses a stepwise approximation to a linear frequency modulation.

This device has poor Doppler tolerance. That is, the autocorrelation function or compressed-pulse waveform that is obtained with the device exhibits large range-time grating lobes in the presence of large Doppler shifts on the echo pulses. These large grating lobes are undesirable as they may give rise to false detections and range, if time discrimination is used to calculate range.

It is therefore an object of this invention to provide a digital pulse expander-compressor that does not develop large range-time-sidelobes with Doppler frequency shifts of the echo from a target.

Another object is to realize such a compressor using a minimum amount of hardware.

These and other objects of the invention are achieved by a digital pulse-expander-compressor which includes a pulse coder means that provides, from a single input pulse, a series of weighted pulses having a time interval greater than the input pulse; and a pulse compressor means, of the matched filter type, that receives and processes echo signals corresponding to the output signals of the coder means. The pulse compression means includes weighting means which is the same as that employed in the coder means. The weighting means includes a discrete Fourier transform circuit; a plurality of phase-shifters of which a respective phase-shifter is connected to every input of the discrete Fourier transform circuit except the first input; and a plurality of inverters of which a respective inverter is connected to every other output of the discrete Fourier transform circuit.

This invention permits a minimum amount of hardware to be used to generate and compress a highly Doppler-tolerant polyphase code.

Additional advantages and features will become apparent as the subject invention becomes better understood by reference to the following detailed description when taken in conjunction with the accompanying drawings wherein:

FIG. 1 is a block diagram of an embodiment of the invention.

FIG. 2 is the autocorrelation function obtained with the pulse-expander-compressor shown in FIG. 1 when subject to no Doppler shift and no bandwidth limitation.

FIG. 3 is the autocorrelation function obtained with the prior art pulse-expander-compressor when subject to no Doppler shift and no bandwidth limitation.

FIG. 4 is the autocorrelation function obtained with the pulse-expander-compressor shown in FIG. 1 with a Doppler shift equal to 5% of the bandwidth and no bandwidth limitation.

FIG. 5 is the autocorrelation function obtained with the prior art pulse expander-compressor when subject to a Doppler shift equal to 5% of the bandwidth and no bandwidth limitation.

FIG. 1 shows a block diagram of the digital phase-coded pulse-expander-compressor disclosed in the above-referenced U.S. Pat. No. 4,237,461 with the addition of phase-shifters 69-71 and inverters 72-73 to the weighting means 20 comprising the chief improvement of the preferred embodiment. The embodiment of the invention is shown as it might be used as part of a pulse-compressor radar. The components of the invention are enclosed in the dashed lines. In brief, the system operates in the following manner. The input signal is a pulse produced by a pulse generator 12, for example. The pulse is clocked into an input-signal expansion means (ISEM) 13 providing three equal intervals of delay. The ISEM may comprise three stages 14, 16, 18 of a shift register or a delay line tapped at equal delay intervals, for example. Outputs are taken from the input to the first delay stage, the output of the first delay stage, the output of the second delay stage, and the output of the third delay stage, viz, y(0), y(1), y(2), y(3), respectively. During the first clock interval the pulse to be coded is clocked out of the pulse generator 12 and put on the input line to the first delay stage 14; during the second clock interval the pulse at the input of the first delay stage 14 passes through the stage and appears at its output with the proper delay; and so on. The clocking circuit is shown in simplified form in block 66. The ISEM 13 esentially provides for time expansion of the input pulse. It provides four input signals to the weighting means 20 viz., 1000,0100, 0010 and 0001, in a time which is four times as long as that of the input pulse. (Presence of a pulse is denoted by a 1, absence of a pulse by a 0.)

The signals y(0), y(1), y(2) and y(3) are fed into a weighting means 20 which provides four output signals, z(0), z(1), z(2), and z(3). The output signals from the weighting means 20 are individually fed to Fourier signal time-dispersion means 21 (FSTDM). This comprises a plurality of adders 22-25, each coupled to a respective output of the weighting means 20 and a plurality of delay stages 26-28 interconnecting the adders. Each of the delay stages 26-28 supplies a delay equal to four clock intervals. Thus, signal z(0) has no delay, signal z(1) has a delay of four intervals, signal z(2) a delay of 8 intervals, and z(3) a delay of 12 intervals. The particular arrangement of the delays and the number of the delays is not fixed by any rule but is a matter of choice by the designer, the more delays inserted, the more the outgoing signal is time-expanded. As a matter of fact, the device would operate with no delays at all in the z(0), z(1), z(2) and z(3) lines but this would be undesirable since the object is to time-stretch the outgoing signal so that the increased detection capability of a long-pulse radar system can be achieved.

It should also be noted that the number of delay stages in the ISEM 13, although they may be as few as one, can be greatly increased in order to time-expand the outgoing signal.

The outputs of the FSTDM 21 are a series of complex signals carried on composite leads comprising separate I (inphase) and Q (quadrature) lines. A single signal with a real and an imaginary part is formed for each clocking interval. These signals are fed to a phase modulator 52 which phase-modulates the carrier signal of the transmitter, and the modulated carrier is propagated into space by the antenna.

The return signals are received, processed by the receiver 60, and fed to a sampler-and-A/D converter 62 which returns them to sample digital form.

These digital echo signals are in complex form and are fed to a conjugator circuit 64 which processes them to form their complex conjugates. These conjugate signals are fed back through the ISEM 13 in inverted order relative to the manner in which the original input pulse passed through the ISEM 13. The delayed outputs are again fed to the weighting means 20 whose output signals are fed to the Fourier signal time-dispersion means (FSTDM) 21 to provide delays for the signals z(0), z(1), z(2) and z(3) which are inverted in order relative to the delays provided by the FSTDM 21 for the outgoing weighting means signals. The output of the FSTDM 21 is a a cross-correlated signal which is then detected by an envelope detector 36.

The output of the FSTDM 21 is the impulse response of a matched filter which comprises complex conjugator 64, input-signal expansion means 13, weighting means 20, and time-dispersion means 21. A matched filter provides an impulse response which is the complex-conjugate, time-inverted, equivalent of the input pulse. The conjugator 64 provides the complex conjugation of the echo signals and the expansion means 13 and time-dispersion means 21 provide the time inversion of the echo signal. The matched filter also effects convolution of the received signals with the input signal. The replica of the input signal in this device is actually a product of the operation of the weighting means 20.

According to the present invention, the weighting means 20 comprises a fast Fourier (FFT) circuit 68, which forms discrete Fourier transform (DFT) signals; a plurality of phase-shifters 69-71 of which a respective phase shifter is connected to every input of the DFT 68 except the first input; and a plurality of inverters 72-73 of which a respective inverter is connected to every other output of the DFT 68. The phase-shifters 69-71 and the input to the first delay stage 14 provide outputs x(n) in accordance with the formula

In the formula, the y(n) are the output signals taken from the input to delay stage 14 and from the output of delay stages 14, 16, and 18; n is the number of the output signal being taken from the delay stages (n=0 for the output signal taken from the input to delay stage 14, and n=1,2,3 for the output signals taken from the outputs of delay stages 14,16, and 18, respectively); and N, an even number, is the number of delay stages plus one, since a pulse is being taken off just prior to the first stage 14. The output signals x(n) are individually fed to the DFT 68 which provides outputs s(k) in accordance with the formula ##EQU1## In the formula, k is the number of the output signal being taken from the DFT 68. Every other output of the DFT 68 is inverted by the inverters 72-73, so that the outputs z(k) of the weighting means 20 are provided to the FSTDM in accordance with the formula

In the example shown n=0, . . . 3, k=0, . . . 3,N=4, although many more delay stages and outputs from the weighting means could be used, as well as fewer. In fact, as little as one delay stage could be used. The phase shifts associated with phase shifters 69-71 are .pi./16, 4.pi./16, and 9.pi./16 respectively. A detailed discussion of the discrete Fourier transform can be read in "Digital Signal Processing", Oppenheim and Schafer, published in 1975 by Prentice-Hall, Inc., Englewood Cliffs, N.J. Various ways of implementing the DFT are shown; in particular, see chap. 6, FIG. 6.10.

The operation of the present invention may be explained as follows:

These outputs are fed to the phase modulator 52 through the FSTDM delays. For the first four clock pulses, the only inputs to the modulator are the signals on the z(0) line; for the next four clock pulses, the signals on the z(1) line; for the next four, the signals on the z(2) line; and for the next four, the signals on the z(3) line. This signal modulates the phase of the transmitter carrier wave. Note that the output signal extends over 16 clock pulse periods. The phase-modulated carrier is propagated through space and, if the signal strikes a target, an echo signal is returned. This echo signal is processed through the receiver 60 and sent through a sampler-and-A/D converter circuit 62 which converts the phase-modulated carrier into a digitized signal again. The complex conjugate of the digitized signal is taken and fed to the ISEM 13, but the ISEM delays are now in time-inverted order, that is, y(3) now has no delay, y(2) is delayed one clock interval, y(1) two clock intervals and y(0) three clock intervals. The time-inverted, delayed echo signals are fed through the weighting means 20 to the FSTDM 21 to provide delays which are inverted in order relative to the delays provided by the FSTDM 21 for the outgoing weighting means signals. Thus, no delay is inserted in signal z(3), four units of delay in z(2), eight units of delay in z(1), and twelve units of delay in z(0). The output of FSTDM 21 has a peak pulse in the sixteenth clock pulse interval, which can be seen in FIG. 2, the output of the envelope detector 36.

FIG. 2 illustrates the autocorrelation function or compressed pulse waveform that is obtained with the digital pulse-expander-compressor shown in FIG. 1 with no Doppler shift and no precompression bandwidth limitation. The sampler number corresponds to a range cell in time in a radar. In this case, the pulse compression ratio p=100 and the highest range-time sidelobe is 4p below the peak response. For comparison purposes, FIG. 3 illustrates the autocorrelation function that is obtained with the Cantrell digital pulse-expander-compressor (shown in the above-referenced U.S. Pat. No. 4,237,461) with no Doppler and no bandwidth limitation--it has 4-db lower peak range-time sidelobes.

FIG. 4 illustrates the effect of a Doppler shift equal to 5% of the bandwidth on the autocorrelation function obtained with the compressor shown in FIG. 1. FIG. 5 illustrates, for comparison purposes, the compressed pulse that results using the Cantrell compressor with this Doppler shift. Note that the undesirable large grating lobes that appear with Doppler in the autocorrelation function obtained with the Cantrell compressor are absent from the autocorrelation function obtained with the compressor of the present invention.

Thus, there has been provided a new and improved digital pulse-expander-compressor having increased Doppler tolerance.

Obviously many modifications and variations of the present invention are possible in light of the above teachings. It is therefore to be understood that within the scope of the appended claims the invention may be practiced otherwise than as specifically described.