Electronic Tracking for Tuners

The present invention relates to tuning systems employing a radio frequency stage and a local oscillator with respective tuned circuits which desirably track each other over a given tuning range.

Television (TV) tuning systems heterodyne a radio frequency (RF) signal corresponding to a desired channel received from broadcast or cable TV signal sources with a local oscillator signal having a frequency set in accordance with the desired channel in order to produce an intermediate frequency (IF) signal from which display and sound information is derived. The frequency of a local oscillator signal is commonly controlled by a tuning voltage. The tuning voltage is developed in response to the selection of a channel by a viewer in any one of a variety of well known ways which may be generally characterized as either voltage synthesis or frequency synthesis. In voltage synthesis the tuning voltages for tuning each channel is stored in either analog form by means of a potentiometer or in digital form in an addressable memory. In frequency synthesis the frequency of the local oscillator or the IF signal is compared to a reference frequency and the tuning voltage is changed until the two frequencies are equal. Typically frequency synthesis is accomplished by a phase-locked loop.

In addition to tuning the local oscillator, the tuning voltage for the local oscillator (LO) can be employed to adjust the frequency selectivity of tunable RF filter circuits which precede the mixer to pass only the RF signal of the selected channel. When the passband of the RF filter circuits does not properly track the frequency of the local oscillator signal (such that the frequency difference is precisely the IF frequency), the desired RF signal may be attenuated and the possibility of interference from adjacent TV channels is increased. While increasing the bandwidth of the RF filter circuits might help reduce attenuation of the desired RF signal in a case of mistracking, it could also worsen the interference problem.

Since the voltage variable capacitance ("varactor") diodes in the the RF filter and local oscillators circuits receive a common tuning voltage but do not have precisely the same tuning characteristics over the tuning range of their respective frequency ranges, mechanically variable inductors and capacitors have been used as trimming elements in the RF filter and LO circuits to cause the frequency characteristics of these circuits to track one another in response to the tuning voltage. The adjustment of these trimming elements is difficult and time consuming, often requiring an iterative procedure. Additionally, they increase the wear and tear on the mechanical portions of the adjustable elements which increases the probability of component failure. Trimming and padding components may also reduce the overall range of frequency variation of the local oscillator to a point where a separate local oscillator is required for each TV band, i.e., low VHF, mid band cable, high VHF, super band cable and UHF bands.

Additionally, in a double-conversion or double-heterodyning tuning system, wherein two frequency conversions are performed to translate the frequency of the received RF signal to a first IF signal and then to a second and final IF signal, RF/LO frequency tracking is even more difficult to achieve. In these double-conversion tracking systems the frequency range of the local oscillator required to generate the first IF signal is generally much higher than that of the received RF signals which must be selected by the RF filter circuits. In a voltage controlled tuner this disparity of frequency ranges makes it difficult to adjust the LO and RF circuits to track one another. When mechanical tracking adjustments are employed, at high frequencies encountered in double-conversion tuning systems, the nearness of an adjusting tool or the hand of the person making the adjustment significantly interferes with the adjustment process.

The present invention concerns a tuning system in which the initial or factory tracking adjustment of the RF and LO circuits is relatively simple. It also allows for electronic tracking of the tuning voltages for the RF and LO tuned circuits, even when a single local oscillator is used for tuning both the low VHF and high VHF television signal bands.

In the present invention, a tuning system for a receiver for providing an IF signal from a selected one of RF signals residing in a signal frequency band, comprises filter means for passing RF signals corresponding to a selected channel in response to a filter tuning control signal, mixing means responsive to the selected RF signals and a local oscillator signal for frequency translating the selected RF signals to develop a predetermined IF signal and a local oscillator for generating the local oscillator signal in response to an oscillator tuning control signal. A DC signal translating means includes an amplifier having an input responsive to said oscillator tuning control signal and having first and second predetermined gain transfer characteristics for modifying the oscillator tuning control signal and for providing at an output the filter tuning control signal. A switch means responsive to one of the tuning control signals changes the gain characteristic of the amplifier from the first characteristic to the second, so as to derive a piecewise-linear approximation of the filter tuning control signal from the oscillator tuning control signal.

In the accompanying drawing:

FIG. 1 illustrates, in block diagram form, a television receiver including a tuning system constructed in accordance with principles of the invention;

FIGS. 2a and 2b illustrate graphic representations of the desired tracking characteristic between the tuning voltages of the RF and local oscillator circuits; and

FIG. 3 is a schematic diagram of a non-linear signal processor constructed in accordance with principles of the present invention to accomplish tracking between the RF and LO circuits of the tuning system of FIG. 1.

In the double conversion tuning system of FIG. 1, RF television signals for broadcast and cable channels are received at UHF antenna input 10, at VHF antenna input 30 and at CATV input 31. In the United States, each channel is allocated about 6 MHz of bandwidth in the frequency spectrum and has a picture carrier at a frequency 1.25 MHz higher than the frequency at the lower limit of its pre-assigned channel bandwidth. The channel numbers and frequency bands of the received RF signals are indicated in the following table:

As indicated in the table, the RF signals in the VHF and CATV bands (54 to 402 MHz) span more than a seven-to-one range of frequencies. Since tuning over a range greater than three-to-one is impractical due to the limited range of voltage-variable capacitance diodes, the tuning apparatus of FIG. 1 for the VHF and CATV bands is partitioned to tune in a low and a high tuning band separated at a frequency within the MB-CATV band, i.e., at about 150 MHz. As a result each of the high and low tuning bands includes signals having less than a three-to-one range of frequencies. Since the UHF tuning band includes RF signals from 470 to 890 MHz, which is less than a three-to-one tuning range, partitioning is not required.

As shown in FIG. 1, when the TV channel selected is within the UHF band (470-890 MHz), a band switching voltage VB3 is present, enabling a UHF band frequency selective tunable filter 14 to pass RF signals from UHF antenna 10 to an RF combiner or diplexer 18. A fixed frequency IF trap 12 precedes tunable filter 14 to attenuate RF signals at the first IF frequency, i.e., about 416 MHz, to prevent the generation of unwanted beat signals. A UHF amplifier 16 is also enabled by band switching voltage VB3 for amplifying the RF signals selected by tunable filter 14 and applying them to diplexer 18.

When the selected channel is in the L-VHF or the lower portion of the MB-CATV bands (Low band, 54-150 MHz), a band switching voltage VB1 is present, enabling a low-band frequency selective tunable filter 20 to pass RF signals from VHF antenna 30 or CATV input terminal 31 to a VHF amplifier 22. When the selected channel is in the upper portion of the MB-CATV, H-VHF or SB-CATV bands (High band 150-402 MHz), a band switching voltage VB2 is present, enabling a high-band frequency selective tunable filter 24 to pass RF signals from VHF antenna 30 or CATV input 31. A fixed frequency high pass filter 26 precedes tunable filters 20 and 24 to attenuate low frequency undesired signals not in the television bands that may otherwise cause signal interference. VHF amplifier 22 is enabled by a band switching voltage VB12 which is present whenever tunable filters 20 or 24 are enabled, for amplifying the selected RF signals and applying them to a second input of diplexer 18. Tunable filters 20, 24 and 14 are responsive to tuning voltages VT1, VT2 and VT3, respectively, for passing substantially only the RF signal corresponding to the selected channel.

A first mixer 40 receives the RF signals selected by either of tunable filters 14, 20 or 24 via diplexer 18 and frequency translates these signals such that the picture carrier of the selected channel is located at a first IF frequency of 415.75 MHz, which is between the SB-CATV and UHF-TV bands. A tunable voltage controlled local oscillator (VCO) 42 is enabled by band switching voltage VB12 and is responsive to a tuning voltage VT for generating local oscillator signals within a frequency range sufficient for mixer 40 to heterodyne RF signals received in either of the Low or High bands to the first IF frequency. A tunable VCO 44 is enabled by the presence of band switching voltage VB3 and is responsive to the tuning voltage VT for generating a local oscillator signal having a frequency range sufficient for mixer 40 to heterodyne RF signals in the UHF band to the first IF frequency.

An amplifier 46 amplifies the local oscillator signals from VCO's 42 and 44 in order that they are of sufficient relative strength to operate mixer 40 at an acceptable distortion level.

The first IF signal from mixer 40 is then amplified by tuned IF amplifier 50 and applied to a second mixer 60 for frequency translating the first IF signal to a second IF signal having a picture carrier frequency of 45.75 MHz, the standard IF frequency used in the United States. A fixed frequency local oscillator 62 supplies a 370 MHz local oscillator signal to mixer 60. An automatic fine tuning control AFT loop can be employed to more precisely regulate the second IF frequency.

A conventional IF filter stage 64 for processing IF signals with a picture carrier at 45.75 MHz, filters the IF signal and applies the result to television signal processing circuitry 66 for demodulating the video and sound information of the IF signal. A kinescope 68 displays a picture in accordance with the video information and a loudspeaker 70 produces an audio response in accordance with the audio information.

Channel selection and determination of the frequency of the LO signal developed by VCO's 42 and 44 are accomplished as follows. A tuning control 72 responds to selection of a channel to develop the tuning voltage VT and the band switching signals VB1, VB2 and VB3. An "OR" circuit 73 is responsive to the VB1 and VB2 band switching signals for developing the VB12 band switching signal. The voltage range of tuning voltage VT for the various bands is indicated in FIGS. 2a and 2b and increases as the channel number increases.

Band switching signals VB1, VB2 and VB3 are at a high level of about 18 volts when a channel in the band to which they correspond has been selected, and are at zero volts when a channel outside that particular band is selected. U.S. patent application Ser. No. 363,567, entitled TUNING SYSTEM FOR A MULTI-BAND TELEVISION RECEIVER, in the name of M. W. Muterspaugh filed on Mar. 30, 1982, and U.S. patent application Ser. No. 271,724, entitled A PHASE-LOCKED LOOP TUNING SYSTEM INCLUDING A PRESCALER CONDITIONED TO OSCILLATE AT AN OUT OF BAND FREQUENCY, filed in the name of D. J. Carlson, et al., on June 5, 1981, both assigned to the same assignee as the present invention, are incorporated herein by reference for the purpose of providing a more detailed description of the double conversion tuning system of FIG. 1 and for describing a phase-locked loop tuning control suitable for developing tuning and band switching signals of the sort developed by control 72, respectively.

Tuning voltages VT1, VT2 and VT3 for tunable filters 20, 24 and 14, respectively, are derived from tuning voltage VT for local oscillators 42 and 44 by a non-linear signal processing circuit 80 such that for a selected channel, for which the proper local oscillator frequency has been generated by means of tuning control unit 72, the proper RF signal is also selected so that IF signal produced by the heterodyning process has the desired IF frequency. In other words, the frequency selectivities of the tunable filters are made to track their frequency of the respective local oscillator signal by non-linear signal processing ciruit 80 for proper heterodyning.

In the prior art where the same VT is applied to a local oscillator and a respective RF stage, trimming and padding capacitors and inductors are adjusted in the tuned circuits of the local oscillator and RF stage in order to force their tuned circuit responses to achieve the required frequency tracking relationship. As previously noted, the use of these adjustable reactances is undesirable since they require difficult and time-consuming factory alignment procedures, introduce structural integrity problems and, due to the additional reactances, tends to reduce the range of frequency variations obtainable by the tuned circuits. As to the latter point, prior art tuning system for tuning RF signals in the five bands noted in Table 1, typically required separate VCO's for tuning the Low and High bands, due to the addition of reactive components which were necessary to achieve satisfactory RF/LO tracking. Non-linear signal processing circuit 80 eliminates the need for additional adjustable reactive components to achieve tracking in the tunable filters and VCO's and therefore eliminates their disadvantages. Due to the elimination of these reactive trimming elements, which, as earlier noted, tends to limit the frequency range of the associated tunable circuit, a single VCO can be used for generating the local oscillator signal, e.g., 471-813 MHz for the Low and High bands. A preferred embodiment for processor 80 will be described with reference to FIGS. 2a, 2b and 3.

FIGS. 2a and 2b graphically illustrate the tracking relationship required in order that the RF signals corresponding to the selected channel are properly heterodyned to the first IF frequency. The tuning voltage VT1 required for low band tunable filter 20 includes the range of voltages between curves 202 and 204. Since the passband of the tunable filter is greater than the bandwidth of the RF signals for the various channels, the tuning voltage VT1 may be within the range of voltages shown between curves 202 and 204 and still pass the selected channel. Note that as the frequency is increased towards the higher end of the MB-CATV band, the tolerance requirement for the RF/LO tuning voltage tracking is relaxed, due to the increased bandwidth of the filter towards the high end of the tuning range. A curve centered between curves 202 and 204 (identified with "X's") provides ideal RF/LO frequency tracking. Due to the above-noted relaxed tolerance, only two intersecting straight lines 206 and 208 are required to provide a piecewise-linear approximation of the ideal tracking curve.

Curves 210 and 212 of FIG. 2a show the voltage range allowed for proper RF/LO tuning voltage tracking in the High band. Again, note that only two intersecting straight line segments 214 and 216 are required to provide piecewise-linear approximation of the ideal tracking curve centered between curves 214 and 216. FIG. 2b graphically illustrates piecewise-linear approximation using straight lines 218 and 220 for providing an acceptable piecewise-linear approximation of the ideal RF/LO tuning voltage tracking between curves 222 and 224 for the UHF band.

As previously noted, when individual tuning voltages are generated for the oscillators and tunable filters, adjustable reactive components for providing tracking of the tuned circuit characteristics are substantially eliminated and the frequency range of the oscillators is increased. As is indicated schematically in FIG. 1 and graphically in FIG. 2a, VCO 42 is able to supply LO signals for both the Low and High bands using a single tuning voltage VT from about 2 to about 22 volts. Thus, VCO 42 does not require any switched reactive components (typically a second inductor in the VCO tuned circuit which is coupled or decoupled in response to band switching signals) for changing the tuned circuit response so as to provide proper RF/LO signal frequency tracking. However, if it is desired to generate the filter tuning voltages from the VCO tuning voltages, it is clear from FIG. 2a that processor 80, for the Low band must provide a voltage increase in order that a 2 to about 23 volt filter tuning voltage range (VT1) is generated by the corresponding oscillator tuning voltage range(VT) of about 2 to 7 volts.

A preferred embodiment for processor 80 is shown in FIG. 3 and comprises three operational amplifiers (OP AMPS) 310, 340 and 360. Each OP AMP is responsive to the tuning voltage VT for generating tuning voltages VT1, VT2 and VT3 for tunable filters 20, 24 and 14, respectively. Each of the OP AMP circuits includes a voltage establishing circuit for providing to the OP AMP a DC offset voltage corresponding to the difference of the LO and RF tuning voltages at the beginning of each band and a break point voltage corresponding to the intersection point of the straight line segments of FIGS. 2a and 2b. Additionally, the feedback path of each OP AMP includes a voltage sensitive threshold conduction device, e.g., a diode, responsive to the break point voltage for selectively controlling the amount of feedback in the OP AMP circuit. The amount of feedback is changed by the conduction of the device to provide the OP AMP's with a first signal transfer characteristic having a slope corresponding to the lower straight line segments illustrated in FIGS. 2a and 2b, and a second signal transfer characteristic having a slope corresponding to the upper straight line segments.

For OP AMP 310, a resistor 312 couples the LO tuning voltage VT to its non-inverting (+) input. A feedback network coupled between the output of OP AMP 310 and its inverting (-) input includes resistors 314, 316, 318, diode 320 and a voltage establishing circuit 322 including resistors 326 and 330 and potentiometers 324 and 328. For the resistor values illustrated in FIG. 3, potentiometer 328, which receives operating voltage via resistor 330, can be adjusted to provide at its wiper a break point voltage such that when the LO tuning voltage is less than 5.9 volts, diode 320 is reversed-biased and the slope of the transfer characteristic of OP AMP 310 is substantially determined by resistors 314 and 316. Potentiometer 324 receives operating voltage via resistor 326 and can be adjusted to provide at its wiper a DC offset voltage to the inverting input of OP AMP 310 which corresponds to the offset voltage required between the LO tuning voltage VT and the filter tuning voltage VT1 at the lower end of the Low band. This establishes for OP AMP 310 a first signal transfer characteristic corresponding to straight line 206 of FIG. 2a.

For LO tuning voltages greater than 5.9 volts diode 320 becomes forward-biased. When diode 320 is forward-biased, resistor 318 is substantially connected in parallel with resistor 316. This reduces the level of feedback for OP AMP 310, establishing a second signal characteristic having increased slope and corresponding substantially to that of straight line segment 208 of FIG. 2a. Note that since the anode of diode 320 is coupled to the inverting input of OP AMP 310, (which, in an ideal operational amplifier, is caused to have the same voltage as that applied to the non-inverting input) diode 320 becomes forward-biased when the LO tuning voltage exceeds the voltage at the wiper of potentiometer 328 by an amount equal to the junction potential of diode 320. Additionally, since the resistance value of potentiometer 328 is relatively small with respect to the value of resistor 318, adjustment of potentiometer 328 does not substantially affect the impedance of the feedback path for OP AMP 310 when diode 320 is forward-biased. Furthermore, readjustment of either potentiometers 324 or 328 in an iterative manner is not necessary since the wiper of potentiometer 328 is isolated from the remainder of the circuit when diode 320 is reverse-biased.

The generation of tuning voltage VT2 by OP AMP 340 for the High band is similar in principle to the above-described generation of the tuning voltage for the Low band. However, as indicated in FIG. 2a the slope of line 216, required for the upper portion of the piecewise-linear approximation needed to achieve satisfactory performance, is slightly less than unity. Accordingly, to produce a gain less than unity, a voltage divider including resistors 342 and 344 is provided for coupling a reduced version of LO tuning voltage to the non-inverting input of OP AMP 340. Additionally, the slope of the transfer characteristic must decrease after reaching a predetermined LO tuning voltage. Thus, for OP AMP 340, resistor 346 couples operating voltage to the wiper of potentiometer 350 for supplying via resistor 348 a DC offset voltage to its inverting input for establishing the proper starting point of straight line segment 216 for the High band. The slope for the transfer characteristic is determined by resistor 358. Potentiometer 352 is adjusted to provide a break point voltage at its wiper corresponding to the intersection point of lines 214 and 216. When the RF tuning voltage VT2 increases to the point that diode 354 becomes forward-biased, resistor 356 is substantially coupled in parallel with resistor 358, thus reducing the gain of OP AMP 340 to match the slope of straight line segment 214, thus completing a piecewise-linear approximation of satisfactory RF/LO tracking for the High band. It should be noted that the feedback arrangement for OP AMP 340 could be the same as for OP AMP 310 except that diode 320 could be poled in the opposite direction. This, however, would result in a confusing alignment procedure because both potentiometer adjustments would affect the low frequency end of the high band, and different alignment procedures would be needed for the low and high bands.

For the UHF tuning band, operational amplifier 360 is biased similarly to the arrangement described above with respect to OP AMP 340. However, instead of having a transfer characteristic which must decrease after a specific point, the transfer characteristic for OP AMP 360 must decrease after the intersection of straight line segments 218 and 220 of FIG. 2b in order to provide piecewise-linear approximation of satisfactory tracking in the UHF-band. Consequently, diode 362 is poled oppositely from diode 354 and potentiometer 368 is adjusted so that resistors 370 and 372 are substantially coupled in parallel so as to establish the slope of straight line segment 218. Potentiometer 364 receives operating potential from resistor 366 and provides a DC offset voltage at its wiper which is applied by resistor 367 to the inverting input of OP AMP 360 for establishing the beginning point of straight line segment 218. Potentiometer 368 is adjusted so that when the UHF filter tuning voltage VT3 is approximately 7.5 volts, diode 362 becomes reverse-biased, removing resistor 370 from the feedback path of OP AMP 360 and changing the slope of its transfer characteristic from straight line segment 218 to straight line segment 220. This completes the RF/LO tracking required for satisfactory operation of the FIG. 1 tuning system.

Modifications of the described tracking arrangement are considered to be within the scope of the appendant claims. For example, some resistors of the feedback network of the OP AMPs of FIG. 3 can be made variable in order to allow adjustment of the slope of each straight line segment of the piecewise-linear approximations. Thus, for OP AMP 310, fixed resistors 316 and 318 can be replaced by potentiometers. Furthermore, additional diodes can be coupled to the inverting input of the OP AMP's so as to providing multiple break points in the piecewise-linear approximations for providing a more accurate tracking.